JP2003318610A - High frequency plane circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To compensate a high-frequency plane circuit for deterioration of its transmission characteristics which is provided on a dielectric board such as glass sheets for cars, etc., and has a transmission line having a width of 0.5 mm or larger between ground conductors and an air bridge crossing the transmission line. <P>SOLUTION: The high-frequency plane circuit 10 comprises a coplanar transmission line 15, having a necked portion 14 longer than the traverse width of an air bridge 17, a dielectric board 11 has a relative dielectric constant of 6.0-7.5, the distance between ground conductors 13 is 0.8-3.8 mm, and a central conductor 12 has a width of 0.6-2.4 mm, thus forming the transmission line 15. It has a spacer 18, having a relative dielectric constant of 4-17 and a thickness of 21-40 μm. <P>COPYRIGHT: (C)2004,JPO

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a high-frequency plane circuit having a transmission line for transmitting microwaves, for example, a coplanar transmission line for transmitting a high-frequency signal of 0.8 to 10 GHz and an air bridge crossing the transmission line. Regarding

[0002]

2. Description of the Related Art In recent years, in the field of communication using microwaves, an active element such as an FET (field effect transistor) is mounted on a plane circuit provided with a plane antenna, and a supplied power signal is amplified to be transmitted from the plane antenna. Active integrated antennas that emit as radio waves have been proposed. Then, it is desired to practically use this active integrated antenna.

For example, in an automatic toll collection system (ETC system), an active integrated antenna in which a transmission line and a plane antenna are provided on an automobile glass plate made of glass having a relative dielectric constant of 6.0 to 7.5 is manufactured. It may be used as a transmitter. In this case, for the signal transmission in the active integrated antenna, a coplanar transmission line is preferably used in order to simplify the configuration of the circuit. For example, the distance between the ground conductors is set to about 0.5 to several mm and the coplanar transmission is performed. The track is constructed.

On the other hand, communication technology using microwaves has advanced today, and in particular, semiconductor process technologies such as sputtering, dry etching and wet etching have been used.
Various techniques for finely fabricating a monolithic integrated circuit (MMIC) in which active elements and passive elements are provided on a dielectric substrate such as GaAs (gallium arsenide) or alumina have been proposed. For example, Japanese Patent Laid-Open No. 11-346105 discloses a microwave planar circuit in which an air bridge is provided in a coplanar transmission line having a center conductor on a dielectric substrate and a ground conductor provided so as to sandwich the center conductor from both sides. Is disclosed. That is, the publication discloses a structure of an air bridge and a coplanar transmission line traversed by the air bridge so as to compensate for a decrease in characteristic impedance caused by an increase in capacitance between the center conductor of the coplanar transmission line and the air bridge. The configuration of the corresponding portion of is disclosed with specific dimensions.

The air bridge crosses over the central conductor of the coplanar transmission line above the coplanar transmission line provided on the circuit board, and separates both sides of the central conductor to equalize the potentials of the adjacent ground conductors. It is a bridge-shaped conductor that electrically connects the ground conductors on both sides. In particular, since higher-order modes occur in discontinuous portions such as bent portions, T-shaped branch portions, and cross branches of the coplanar transmission line, it is necessary to provide an air bridge to make the ground conductors on both sides have the same potential.

In addition, 'Theoretical and Experimental S
tudy of Various Types of Compensated Dielectric Br
idges for Millimeter-Wave Coplanar Application ''
(Eric Rius et al., IEEE Trans. Microwave Theory T
ech., vol.48, No.1, pp.152-156, Jan.2000) has a thickness of 20 μm between the air bridge and the coplanar transmission line.
Discloses a high frequency circuit in which a spacer made of a dielectric layer is provided and the height of the air bridge is 20 μm. Also, 'Three-Dimensional High-Frequency Distributio
n Networks-Part 1: Optimization of CPW Discontinui
ties' (Thomas M. Weller et al., IEEE Trans. Microw
ave Theory Tech., vol.48, No.10, pp.1635-1642, Oc
t.2000) discloses a high frequency circuit in which the height of the air bridge with respect to the coplanar transmission line is 3 μm.
In order to compensate for the decrease in characteristic impedance caused by the proximity of the air bridge and the center conductor of the coplanar transmission line and the resulting deterioration of the transmission characteristics of the coplanar transmission line, the coplanar transmission line is The bridge has a constricted portion in which the width of the center conductor is narrowed at a corresponding portion of the center conductor which the bridge crosses.

On the other hand, Japanese Unexamined Patent Publication No. 2001-144177 discloses a circuit provided with the above-mentioned air bridge and coplanar transmission line by using the process technology of the semiconductor integrated circuit of the dry etching process and the wet etching process using a photosensitive resist. Discloses a method of manufacturing.

[0008]

The above-described coplanar transmission line in the MMIC is manufactured by using the semiconductor manufacturing process technology, and has a relative dielectric constant of Ga of 12.9.
It was manufactured by finely processing a dielectric substrate made of As or a dielectric substrate made of alumina with a relative permittivity of 9.6, and the height of the air bridge was 20 μm in each case.
m or less, and the distance between the ground conductors is 260 μm
(0.26 mm) or less. On the other hand, in the case of the above-mentioned active integrated antenna, for example, the dielectric constant is 6.0.
A coplanar transmission line having a long distance between the ground conductors, such as a distance between the ground conductors of approximately 0.5 to several mm, is produced on a glass plate of ˜7.5. On the other hand, the characteristic impedance of the coplanar transmission line changes depending on the distance between the ground conductors.
The above-mentioned coplanar transmission line having a distance between the ground conductors of about 0.5 to several mm is a coplanar transmission line having a short distance between the ground conductors produced by using the above-described MMIC technology for high frequency signals in the GHz band. Can be said to be completely different.

Therefore, in the coplanar transmission line having a large distance between the ground conductors, if the values disclosed in the above-mentioned prior art for the height of the air bridge are used as they are,
The characteristic impedance cannot be made appropriate and the deterioration of the transmission characteristic cannot be compensated. Moreover, in order to obtain an active integrated antenna with high radiation efficiency, the numerical value of the characteristic impedance must be adjusted accurately. In particular, it is important to provide a high-frequency planar circuit that compensates for the transmission characteristics of a high-frequency signal of 0.8-10 GHz corresponding to the frequency band of radio waves of 0.8-10 GHz that is increasingly used in the field of communication technology. .

Further, the combination of the dimensions of the coplanar transmission line and the dimensions of the air bridge disclosed in the above-mentioned prior art shows only one embodiment or several embodiments, and the combination of the dimensions of the coplanar transmission line is not shown. It does not disclose a method of adjusting the size of the air bridge with respect to the size. Therefore, it is impossible to design a high-frequency plane circuit so that the relative permittivity is 6.0 to 7.5 and the distance between the ground conductors is longer than that of the conventional coplanar transmission line so as to have stable transmission characteristics.

On the other hand, it is conceivable to find a combination of the configuration of the coplanar transmission line and the configuration of the air bridge, which have good transmission characteristics, while variously changing the configuration of the coplanar transmission line and the configuration of the air bridge. However, when setting the configuration of the coplanar transmission line, for example, it is necessary to determine the distance between the ground conductors, the width of the central conductor, the width of the constricted portion of the central conductor crossed by the air bridge, and the length of the constricted portion. When the conductor width of the air bridge, the height of the air bridge, and the spacer made of a dielectric material holding the air bridge are provided, it is necessary to determine the relative permittivity of the spacer and the width of the spacer. As described above, since there are an unlimited number of combinations of the configuration of the coplanar transmission line with good transmission efficiency and the configuration of the air bridge, among these, the coplanar transmission line having a good and stable transmission characteristic for a high frequency signal of a predetermined frequency. And it is extremely difficult to find the configuration of the air bridge.

Further, when such an active integrated antenna is constructed by using a glass plate for an automobile as a substrate, it is necessary to manufacture it by a method suitable for mass production, but it is much more difficult than a substrate used for a semiconductor device. Since a large-area automobile glass plate is used as the substrate, the above-mentioned publication (Japanese Patent Laid-Open No. 2001-2001
It is impossible to practically manufacture a small-area high-frequency plane circuit including a coplanar transmission line and the like on a large-area glass plate for automobiles by using a semiconductor process technique such as Japanese Patent No. 144177).

Therefore, in order to solve the above-mentioned problems, the present invention solves the above problems by using a dielectric substrate having a relative permittivity of 6.0 to 7.5 such as a glass plate for automobiles and both sides of the center conductor on the dielectric substrate. A ground line is provided with a ground conductor, and the width between the ground conductors is relatively large compared to that made by MMIC, and a conductor that crosses the center conductor of this transmission line and connects the ground conductors on both sides. In a high-frequency plane circuit including an air bridge, the high-frequency plane circuit can compensate for deterioration of transmission characteristics and can be easily manufactured without using a semiconductor process technology, particularly 0.8 to 10 GHz and 0.8 to 6 GH.
It is an object of the present invention to provide a high frequency plane circuit that efficiently transmits a high frequency signal in the z frequency band.

[0014]

In order to achieve the above-mentioned object, the inventors of the present invention have conducted various studies and as a result, have found that the transmission line in a high-frequency plane circuit has a certain preferable range for the transmission line with good transmission characteristics. The present invention has been made by finding that there is such a problem. That is, according to the present invention, a dielectric substrate having a relative permittivity of 6.0 to 7.5, a center conductor provided on the dielectric substrate, and both sides of the center conductor so as to sandwich the center conductor with a space therebetween. A transmission line having a constricted part in which the width of the center conductor is partially narrowed, and a dielectric provided on the constricted part. A high-frequency plane circuit comprising a spacer and an air bridge that is held by the spacer and that crosses the central conductor above the constricted portion and that connects the ground conductors on both sides of the central conductor. The length of the constricted portion is longer than the transverse width of the air bridge, the distance between the ground conductors of the transmission line is 0.8 to 3.8 mm, and the width of the central conductor other than the constricted portion is 0.6-2.4 M, and further the dielectric constant of the spacer 21 to a thickness in the 4-17
A high-frequency planar circuit having a thickness of 40 μm is provided.

At this time, the capacitance characteristic parameter relating to the capacitance characteristic of the transmission line due to the provision of the air bridge is α, and the induction characteristic parameter relating to the inductivity of the transmission line due to the provision of the constricted portion is β,
When the α-β coordinate plane having the capacitance characteristic parameter α on the horizontal axis and the inductive parameter β on the vertical axis is determined, the capacitive parameter α and the inductive parameter β are
It is preferable to configure the transmission line and the bridge so that the transmission line and the bridge exist within a range surrounded by a straight line represented by the following formulas (1) to (4). α = 4 (1) α = 5.5 / L ₇ ^1/2 +15.77 (2) β = 0.32 · α + 0.71 · L ₇ − 2.05 (3) β = 0.32 · α + 0.29 · L ₇ + 1.25 (4) However, the capacitance characteristic parameter α is α = ε _r · L ₃ ·
L ₄ / L ₂ , the induction characteristic parameter β is β = L ₅ ·
log _e (L ₇ / L ₄ ), L ₇ is the distance (mm) between the ground conductors, ε _r is the relative permittivity of the spacer, L ₂
Is the height (mm) of the air bridge, L ₃ is the width (mm) of the air bridge, L ₄ is the width (mm) of the center conductor of the constricted portion, and L ₅ is the length of the constricted portion ( mm). The setting of the capacitance characteristic parameter α and the induction characteristic parameter β performed based on the above equations (1) to (4) is preferably used when transmitting a high frequency signal of 0.8 to 6 GHz on the transmission line. .

Further, in the α-β coordinate plane, the capacitive parameter α is set in a range of a region surrounded by a straight line represented by the above equations (1) and (4) and the following equations (5) and (6). And the transmission line and the bridge are preferably arranged such that the inductive parameter β is present. α = 10.6 / L ₇ ^1/2 +0.92 (5) β = 0.32 · α + 0.58 · L ₇ −1.2 (6) The above formulas (1), (4) and (5) The capacitance characteristic parameter α and the inductive characteristic parameter β are set on the basis of (6) and 0.8-10 GH for the transmission line.
It is preferably used when transmitting a high frequency signal of z.

The spacer has a thickness of 25 to 35 μm.
More preferably m. For example, a planar antenna is provided at the end of the transmission line on the dielectric substrate, and an active element is further provided on the dielectric substrate. The dielectric substrate is, for example, transparent glass. Such a high-frequency plane circuit is formed, for example, by laminating each constituent member to be the ground conductor, the center conductor, the spacer, and the air bridge by screen printing on a transfer film to form a predetermined pattern, It is produced by transferring to a dielectric substrate.

[0018]

BEST MODE FOR CARRYING OUT THE INVENTION The high-frequency planar circuit of the present invention will be described below in detail with reference to the preferred embodiments shown in the accompanying drawings.

FIG. 1 is a perspective view showing a high frequency plane circuit 10 which is an example of the high frequency plane circuit of the present invention. 2 is a cross-sectional view of the high frequency plane circuit 10 taken along the line AB shown in FIG. 1, and FIG. 3 is a high frequency plane circuit 10 taken along the line XY shown in FIG.
FIG. FIG. 4 is a high-frequency plane circuit 1 shown in FIG.
It is a top view of 0.

The high frequency plane circuit 10 includes a dielectric substrate 11
And the ground conductors 13 on both sides of the center conductor 12 so as to sandwich and sandwich the center conductor 12 provided on the dielectric substrate 11.
(13a, 13b) are adjacent to each other, and the coplanar transmission line 15 having the constricted portion 14 in which the width of the central conductor 12 is partially narrowed, and the central conductor 16 of the constricted portion 14 are crossed above the dielectric substrate 11. , The ground conductor 1 on both sides of the center conductor 12
The air bridge 17 formed of a conductor electrically connecting the three and the constricted portion 14 is provided.
And a spacer 18 for holding 7. Spacer 1
Reference numeral 8 is a member made of a dielectric material for holding the air bridge 17, and is provided in contact with the upper portion of the constricted portion 16 as shown in FIG. In FIG. 1, in order to hide the waist 16,
The spacer 18 is shown by a dotted line.

The dielectric substrate 11 has a relative dielectric constant of 6.0 to 6.0.
The plate-shaped dielectric has a thickness of 7.5 and a thickness of, for example, 1 to 10 mm, more preferably 2 to 6 mm, and a coplanar transmission line is provided on the substrate surface of the dielectric substrate 11. Examples of the dielectric substrate 11 include a glass plate for automobiles. It may also be laminated glass or double glazing. In this case, either the glass surface of the laminated glass plate or the double-layer glass plate may be the substrate surface of the dielectric substrate 11.

The coplanar transmission line 15 is the dielectric substrate 1
1 and the ground conductors 13a, 13b that are adjacent to each other on both sides of the center conductor 12 with a constant gap.
It is a known transmission line through which a high-frequency signal is transmitted along the slot portion 19 formed between and. The distance between the ground conductors 13a and 13b provided on both sides of the center conductor 12 is 0.
It is 8 to 3.8 mm. On the other hand, the width of the central conductor 12 is 0.
6 to 2.4 mm, the width of the slot portion 19 is 0.2 to 0.6
mm, and the slot portions 19 having the same width are provided on both sides of the center conductor 12. In addition, coplanar transmission line 1
One end of 5 is connected to, for example, the drain terminal of an FET (field effect transistor) mounted on the dielectric substrate 11, and the other end is, for example, a planar antenna provided on the dielectric substrate 11. And is configured to emit a high frequency signal amplified by the FET as a radio wave.

The coplanar transmission line 15 includes the ground conductor 13
The central conductor 12 has a narrowed portion 14 corresponding to a crossing portion of the air bridge 17 connecting between the a and 13b. The provision of the air bridge 17 increases the capacitance of the transmission line, and the increase reduces the characteristic impedance of the coplanar transmission line 15. However, the constricted portion 14 compensates for the decrease in the characteristic impedance of the central conductor 12. Has a function of increasing the characteristic impedance by making the width narrower than the width. The length of the constricted portion 14 is the same as that of the air bridge 1 described later.
7, the center conductor 16 of the constricted portion 14 is arranged so that the air bridge 17 crosses a part of the constricted portion 14.
Has been set. The constricted portion 14 has, for example, a width of the central conductor 16 of 0.2 to 0.8 mm, and
Is longer than the width of the air bridge 17 and is 6.0 mm or less. The ground conductors 13a, 1
A conductive material such as gold or silver is used for 3b and the central conductors 12 and 16, and the conductive material is not particularly limited.

The air bridge 17 includes a ground conductor 13a,
A conductor connecting 13b, which is held by a spacer 18 and crosses the central conductor 16 of the constricted portion 14. A conductive material such as gold or silver is used for the air bridge 17, and the conductive material is not particularly limited. The spacer 18 is a dielectric having a relative permittivity ε _r of 4 to 17 and a height of 21 to 40 μm.
The dielectric material of the spacer 18 is not particularly limited, but for example, B ₂ O ₃ —SiO ₂ —PbO based glass is used.

Such a high-frequency plane circuit 10 is manufactured by using, for example, the transfer type print pattern forming method described in Japanese Patent Laid-Open No. 2000-151080. That is, on a film such as a transfer paper having a base paper and an easily peelable layer formed on the base paper, a paste of a conductive material to be the air bridge 17 and a dielectric material to be the spacer 18 are formed. After laminating the paste at a predetermined position, the ground conductor 13 (13a, 13b) and the center conductor 1 as shown in FIG.
The circuit patterns are formed by laminating the pastes of the conductor materials to be the conductors 2 and 16 to prepare the circuit patterns to be transferred. Lamination of various pastes is performed by a known screen printing method on the easily peelable layer on the film. On the other hand, an adhesive is laminated on the surface of the dielectric substrate 11 by known screen printing. After that, the circuit pattern laminated on the film is pressed on the substrate to be the dielectric substrate 11 with a predetermined pressure using a press plate, further heated, and then only the base paper is peeled off to form the circuit pattern on the dielectric substrate. Transfer to 11. Furthermore, the dielectric substrate 11 on which the circuit pattern is transferred
Is fired at a predetermined temperature. Thus, the high frequency plane circuit 10
Can be produced.

When the spacers 18 are made of glass, the dielectric material paste of the spacers 18 contains, for example, B ₂ O ₃ --SiO ₂ --having a softening point of 520 to 580 ° C.
A PbO-based glass frit is used. When the dielectric substrate 11 is a glass plate, it is preferable to use a glass frit that is sufficiently softened and fluidized under the conditions for firing the glass plate. Also,
A filler material such as Al ₂ O ₃ or ZrO ₂ may be added within a range that does not hinder the above-mentioned softening flow in order to suppress the thermal expansion coefficient of the spacer 18 and improve the strength.

As described above, in the high frequency plane circuit 10, the dielectric substrate 11 has a relative permittivity in the range of 6.0 to 7.5, and the distance between the ground conductors 13a and 13b is 0.8 to 3.8 mm.
In addition, the coplanar transmission line 15 is formed in the range where the width of the central conductor 12 is 0.6 to 2.4 mm, and the relative permittivity ε _r is 4 to.
The spacer 18 is composed of 17 and a thickness of 21 to 40 μm. In such a configuration, the waist portion 1
Since the length of 4 can be set longer than the cross width of the air bridge 17 and the length of the constricted portion 14 (L ₅ in FIG. 4) can be set, the characteristic impedance can be efficiently set as shown below. It can be adjusted, and the transmission characteristics in the high frequency plane circuit are improved. Moreover, stable transmission characteristics can be obtained in a predetermined frequency band.
In the structure of the high-frequency planar circuit 10 shown in FIGS. 1 and 3, the height direction is enlarged to facilitate understanding of the air bridge 17 or the spacer 18.

By the way, a transmission line such as a coplanar transmission line for transmitting a high frequency signal is constructed so that the characteristic impedance becomes a constant value, for example, 50Ω. As is well known, this characteristic impedance is characterized by using the capacitance characteristic (capacitance) and the inductive characteristic (inductance) of the transmission line when the loss due to resistance is not considered. In the high-frequency plane circuit 10, the capacitance characteristic parameter α and the inductive characteristic parameter β are determined so that the transmission characteristics of the coplanar transmission line 15 are not deteriorated even if the air bridge 17 is provided, and the coplanar transmission line 15 and the air characteristic are set. The bridge 17 and the spacer 18 are configured. That is, when the present inventors use the capacitance characteristic parameter α and the inductive characteristic parameter β shown below,
0.8-6 GHz frequency band, 0.8-10 GH
The present invention has been completed by finding the regions of the capacitance characteristic parameter α and the inductive characteristic parameter β in which the transmission characteristic becomes good in the z frequency band.

Here, the capacitance characteristic parameter α and the inductive characteristic parameter β are defined as α = ε _r · L ₃ · L ₄ / L ₂ and β = L ₅ · log _e (L ₇ / L ₄ ). As shown in FIGS. 3 and 4, L ₇ is a ground conductor 13a, 1
Distance (mm) between 3b, ε _r is the relative permittivity of the spacer 18, L ₂ is the height (mm) of the air bridge 17, L ₃ is the width (mm) of the air bridge 17, and L ₄ is the constricted portion 14.
The width (mm) of the central conductor 16 of L and the constricted portion 1 is L _5.
The length is 4 (mm).

Here, regarding the capacitive parameter α and the inductive characteristic parameter β in the high frequency plane circuit 10, when an α-β coordinate plane having the capacitive characteristic parameter α on the horizontal axis and the inductive parameter β on the vertical axis is determined, It is preferable for the high frequency signal of 0.8 to 6 GHz to exist in the range surrounded by the straight lines represented by the following formulas (1) to (4). α = 4 (1) α = 5.5 / L ₇ ^1/2 +15.77 (2) β = 0.32 · α + 0.71 · L ₇ − 2.05 (3) β = 0.32 · α + 0.29 · L ₇ + 1.25 (4)

By constructing the coplanar transmission line 15, the air bridge 17, and the spacer 18 having such a capacitive parameter α and inductive characteristic parameter β,
The return loss can be -20 dB or less for a high frequency signal of 0.8 to 6 GHz. By providing the air bridge 17 as described above, the characteristic impedance of the coplanar transmission line 15 changes and a part of the high frequency signal is reflected. The return loss is the reflection signal generated by the change of the characteristic impedance. The ratio of power to high frequency signals. Therefore, -2
0 dB or less means that the power of the reflected signal with respect to the power of the input high frequency signal is 1% or less, which means that the transmission characteristics are good. In particular, the value of -20 dB, which is used to judge the quality of the transmission characteristic by using the ratio of the power of the reflected signal to the power of the incident high frequency signal, is a reference value when it is determined that the transmission characteristic is good. This value is commonly used by traders.

For example, ε _r = 12.5, L ₂ = 0.03
(Mm), L ₃ = 0.2 (mm), L ₄ = 0.2 (m
m), L ₅ = 2.8 (mm) and L ₇ = 1.6 (m
m), the capacitance characteristic parameter α = 16.67 (m
m) and the induction characteristic parameter β = 5.822 (mm), which exists in the range surrounded by the straight lines represented by the formulas (1) to (4) on the α-β coordinate plane. in this way,
The high-frequency plane circuit 10 is located in a range surrounded by the straight lines represented by the above equations (1) to (4) for the high-frequency signal of 0.8 to 6 GHz and the capacitance characteristic parameter α and the induction characteristic parameter β. Thus, the configuration of the coplanar transmission line 15, the configuration of the air bridge 17, and the configuration of the spacer 18 are determined. This makes it possible to reduce the return loss to −20 dB or less for a high frequency signal of 0.8 to 6 GHz, and to obtain stable transmission characteristics. The point that the return loss becomes -20 dB or less will be described later.

Further, the capacitive parameter α and the inductive characteristic parameter β in the high frequency plane circuit 10 are 0.8.
Exists in a range surrounded by straight lines represented by the above formula (1), the above formula (4), the following formula (5) and the following formula (6) on the α-β coordinate plane for a high frequency signal of 10 GHz. Preferably. α = 10.6 / L ₇ ^1/2 +0.92 (5) β = 0.32 · α + 0.58 · L ₇ − 1.2 (6)

For example, ε _r = 4, L ₂ = 0.03 (m
m), L ₃ = 0.2 (mm), L ₄ = 0.2 (mm),
At L ₅ = 1.1 (mm) and L ₇ = 1.6 (mm), the capacitance characteristic parameter α was 5.33 (mm), the induction characteristic parameter β was 2.29 (mm), and the α-β coordinate was obtained. Equations (1), (4), (5) and (6) in the plane
It exists in the range surrounded by the straight line represented by. Thus, in the case of a high frequency signal of 0.8 to 10 GHz, the capacitance characteristic parameter α and the inductive characteristic parameter β are α−
The configuration of the coplanar transmission line 15 and the configuration of the air bridge 17 are located in the range surrounded by the straight lines represented by the above formulas (1), (4), (5) and (6) on the β coordinate plane. And the configuration of the spacer 18 is defined.
As a result, the return loss in a high frequency signal of 0.8 to 10 GHz can be set to -20 dB or less, and stable transmission characteristics can be obtained. Return loss is -2
The point at which it becomes 0 dB or less will be described later.

As described above, the relative permittivity of the dielectric substrate 11 is in the range of 6.0 to 7.5, and the ground conductors 13a and 13b.
In the range of 0.8 to 3.8 mm and the width of the center conductor 12 in the range of 0.6 to 2.4 mm.
And the relative permittivity ε _r is 4 to 17 and the thickness is 21 to
When the spacer 18 is formed in the range of 40 μm,
The well-known FDTD (Finite Difference Time Dom) that quantitatively and accurately simulates the actual behavior that the condition that the return loss is -20 dB or less is determined using the capacitive parameter α and the inductive characteristic parameter β
ain) method.

The FDTD method is one of the methods for analyzing the transient behavior of electromagnetic waves by numerical simulation using an electronic computer. This method, for example, divides the circuit to be analyzed and the space surrounding this circuit by the desired cells, and uses Maxwell's equation, which is the governing equation, or the equation expressing Maxwell's equations in an integral form for each cell. , A method of creating a difference scheme for the electric field and the magnetic field in each cell and each time step, and calculating the transient behavior of electromagnetic waves using this difference scheme. Details are described in, for example, “Electromagnetic field and antenna analysis by FDTD method” (Kou Uno, published by Corona Publishing Co., Ltd.).

FIG. 5 shows a circuit model 20 of a high frequency plane circuit 10 having a substrate surface having a width of 20 mm and a length of 10 mm.
It is a perspective view which arranged the analysis model in the analysis space 22 of a rectangular parallelepiped of 0 mm, 10 mm in length, and 20 mm in height. This analytical model is then mesh-divided into a rectangular parallelepiped shape and divided into 800,000 cells.

FIG. 6A is a cross-sectional view showing a part of a cut surface of the analysis model divided into meshes, which is cut along the substrate surface of the circuit model 20. This cross section is
It shows a state of an analytical model of a partial region on the substrate surface having a width of 8 mm and a length of 10 mm. FIG. 6A shows the ground conductor 13 of the analytical model corresponding to the ground conductors 13a and 13b.
a ′ and 13 b ′ correspond to the center conductor 12, the center conductor 12 ′ of the analytical model corresponds to the constricted portion 14, and the constricted portion 14 ′ of the analytical model corresponds to the constricted transmission line 15. Coplanar transmission line 15 'is shown.

FIG. 6 (b) is a sectional view of the analytical model taken along the line UV in FIG. 6 (a), which is enlarged with the constricted portion 14 'as the center. Figure 6
In (b), the dielectric substrate 11 ′ of the analytical model corresponding to the dielectric substrate 11, the central conductor 16 ′ of the constricted portion 14 ′ of the analytical model corresponding to the central conductor 16 of the constricted portion 14,
An analysis model air bridge 17 ′ is shown corresponding to the air bridge 17, and an analysis model spacer 18 ′ is shown corresponding to the spacer 18.

[0040]

Example 1 With respect to such an analytical model, as shown in Table 1 ((1) to (5)) below, the width L ₆ of the center conductor 12 is 1, the ground conductor 13a ′, 1
The distance between 3b ′ is L ₇ = 1.6 mm, the slit portion 19
With the width of 0.3 mm fixed and the dimensions L ₂ , L ₃ , L ₄ , L ₅ , and L ₈ shown in FIGS. 3 and 4 changed variously to obtain 132 analysis models (M _{111 to} M ₃₀₁ ). Was created and the return loss S ₁₁ in the analytical model was calculated. The calculated return loss S ₁₁ is 0.8 to 2 GHz, 0.8 to 4 GH
z, 0.8-6 GHz, 0.8-8 GHz, 0.8-1
Return loss S ₁₁ of each frequency band of 0 GHz is -20d
Whether or not it is B or less is discriminated and classified into 6 types. At the same time, the capacity characteristic parameter α in each analysis model
And the induction characteristic parameter β was calculated.

[0041]

[Table 1]

[Table 2]

[Table 3]

[Table 4]

[Table 5]

Here, the return loss S ₁₁ is the analysis space 22.
It is the ratio of the reflected power to the power input to (see FIG. 5). The most dominant first-order mode among the transmission modes of the coplanar transmission line 15 ′ is calculated on one end surface of the analysis space 22 that is in contact with one end of the coplanar transmission line 15 ′, and the electromagnetic wave of this first-order mode is input to the source signal. Is input to the analysis model and the electromagnetic field in the analysis space 22 is analyzed by the FDTD method. From this analysis, the coplanar transmission line 1
The return loss S ₁₁ is calculated by calculating the reflected signal that has returned to the same one end face of the analysis space 22 to which the 5 ′ input wave source signal has been input. Return loss S ₁₁
Is 0 dB means that the input high frequency signal is completely reflected and is not transmitted, and the smaller the return loss S _{11, the} more the power of the input high frequency signal is transmitted without loss.

FIG. 7 is a scatter diagram in which the results of the return loss S ₁₁ in a total of 132 analytical models of M _{111 to} M ₃₀₁ are divided and plotted for each predetermined classification. That is, FIG.
Is based on the classification result of the return loss S ₁₁ of the analytical model of M _{111 to} M ₃₀₁ on the α-β coordinate plane having the capacitance characteristic parameter α on the horizontal axis and the induction characteristic parameter β on the vertical axis.
"▲", "■", "●", "△", "○" and "×"
It is plotted in.

In FIG. 7, “▲” indicates that the return loss S ₁₁ is −20 dB or less in the frequency band of 0.8 to 10 GHz. "■" is, return loss S ₁₁ in a frequency band of 0.8~10GHz does not meet the following conditions -20dB, return loss S ₁₁ in a frequency band of 0.8~8GHz
Is -20 dB or less. "●" is 0.8
Return loss S ₁₁ is -20d in the frequency band of ~ 8 GHz.
It shows that the condition of B or less is not satisfied, and the return loss S ₁₁ is −20 dB or less in the frequency band of 0.8 to 6 GHz. “Δ” indicates that the return loss S ₁₁ does not satisfy the condition of −20 dB or less in the frequency band of 0.8 to 6 GHz and is 0.8.
Return loss S ₁₁ is -20d in the frequency band of ~ 4 GHz.
B or less is shown. "○" means 0.8 to 4 GHz
The return loss S ₁₁ does not satisfy the condition that the return loss S ₁₁ is −20 dB or less in the frequency band of, and the return loss S ₁₁ is −20 dB or less in the frequency band of 0.8 to 2 GHz. "X"
Is a return loss S even in the frequency band of 0.8 to 2 GHz.
₁₁ indicates that the condition of -20 dB or less is not satisfied.

FIG. 8 shows changes in the return loss S ₁₁ with respect to the frequencies of the analytical models M ₁₃₃ , M ₁₅₄ and M ₂₅₂ plotted in FIG. The analytical model M ₁₃₃ has a return loss S ₁₁ of −20 dB or less at 0.8 to 10 GHz. Therefore, the analysis model M ₁₃₃ is shown in FIG.
Are plotted with “▲”. On the other hand, the analytical model M ₁₅₄ has a return loss S ₁₁ of −20 dB or less at 0.8 to 6 GHz, but has a return loss S ₁₁ of greater than −20 dB at 7 GHz to 10 GHz. Therefore, the analysis model M ₁₅₄ is represented by “●” in FIG.
Is plotted in. Further, the analytical model M ₂₅₂ is 0.
It has a return loss S ₁₁ of more than −20 dB at 8 to 2 GHz. Therefore, the analytical model M ₂₅₂ is plotted with “x” in FIG. 7.

Further, T ₁₁ , T ₁₃ , T ₁₄ and T ₁₆ in FIG. 7 are straight lines represented by the following equations, respectively, and the above equation (1) when L ₇ = 1.6 mm is set. ~
This corresponds to the straight line represented by (4). Further, T ₁₂ and T ₁₅ in FIG. 7 are also straight lines represented by the formulas shown below, and the straight lines represented by the above formulas (5) and (6) when L ₇ = 1.6 mm. Corresponding to. T ₁₁ : α = 4 T ₁₃ : α = 20.12 T ₁₄ : β = 0.32 · α − 0.91 T ₁₆ : β = 0.32 · α + 1.71 T ₁₂ : α = 8.30 T ₁₅ : β = 0.32 · α-0.27

From this, the return loss S ₁₁ is 0.8 to 6
The range of −20 dB or less in the GHz frequency band is a range surrounded by a quadrangle having points P ₁₁ , P ₁₃ , P ₁₄ , and P ₁₅ as vertices, and this range is defined by the above formulas (1) to (4). ) Is a range surrounded by a straight line (L ₇ = 1.6 mm).
Similarly, the range in which the return loss S ₁₁ is −20 dB or less in the frequency band of 0.8 to 10 GHz is points P ₁₁ , P ₁₂ , P.
₁₆ and P ₁₇ are in a range surrounded by a quadrangle having vertices and are surrounded by a straight line (L ₇ = 1.6 mm) represented by the above formulas (1), (4), (5) and (6). The range is

Example 2 Furthermore, the following Table 2 ((
As shown in 1) and (2)),
Width L₆= 0.6 mm, between ground conductors 13a 'and 13b'
The distance of L₇= 1 mm, the width of the slit portion 19 is 0.2 m
Fixed to m, the dimension L shown in FIGS. 3 and 4 ₂, L₃,
L_Four, L_FiveAnd L₈And change the analysis model to 5
3 (M₅₁₁~ M₆₁₉), And in the analysis model
Return loss S₁₁Was calculated.

[0049]

[Table 6]

[Table 7]

FIG. 9 shows the results of the return loss S ₁₁ in a total of 53 analysis models of M _{511 to} M ₆₁₉ , which are divided into 6 types as in the case of FIG. 7, and are represented by "▲", "■", " ● ”,
It is a scatter diagram plotted by "△", "○", and "x". That is, 0.8 to 2 GHz, 0.8 to 4 GHz,
0.8-6GHz, 0.8-8GHz, 0.8-10G
While determining whether the return loss S ₁₁ of each frequency band of Hz is -20 dB or less and classifying into 6 types,
The capacitance characteristic parameter α and the inductive characteristic parameter β in each analysis model are calculated, and M ₅₁₁ ~ on the α-β coordinate plane.
The return loss S ₁₁ of the analytical model of M ₆₁₉ is divided into 6 categories and plotted.

T ₅₁ , T ₅₃ , T ₅₄ and T ₅₆ in FIG.
Are straight lines represented by the formulas shown below and L ₇
It corresponds to the straight lines represented by the above formulas (1) to (4) when = 1 mm. In addition, T ₅₂ and T ₅₅ in FIG.
Are also straight lines represented by the formulas shown below, and L ₇
It corresponds to the straight lines represented by the above equations (5) and (6) when = 1 mm. T ₅₁ : α = 4 T ₅₃ : α = 21.27 T ₅₄ : β = 0.32 · α-1.34 T ₅₆ : β = 0.32 · α + 1.54 T ₅₂ : α = 11.52 T ₅₅ : β = 0.32 · α-0.62

From this, the return loss S ₁₁ is 0.8 to 6
The range below -20 dB in the frequency band of GHz is point P.
₅₁ , P ₅₃ , P ₅₄ , and P ₅₅ are in a range surrounded by a rectangle having vertices as vertices, and the straight line (L
₇ = 1 mm). Similarly, the return loss S ₁₁ is −2 in the frequency band of 0.8 to 10 GHz.
The range of 0 dB or less is a range surrounded by a quadrangle having points P ₅₁ , P ₅₂ , P ₅₆ , and P ₅₇ as vertices, and this range is defined by the above formulas (1), (4), (5), and ( It corresponds to the range surrounded by the straight line (L ₇ = 1 mm) represented by 6).

(Example 3) Furthermore, the following Table 3 ((
As shown in 1) and (2)),
Width L₆= 2.4 mm, between the ground conductors 13a 'and 13b'
The distance of L₇= 3.6 mm, the width of the slit portion 19 is 0.
Fixed to 6 mm, the dimension L shown in FIGS. 3 and 4₂, L
₃, L_Four, L_FiveAnd L₈Analysis model
34 (M ₇₁₁~ M₈₂₁) Is created and added to the analysis model.
Return loss S₁₁Was calculated.

[0054]

[Table 8]

[Table 9]

FIG. 10 shows the results of the return loss S ₁₁ in a total of 34 analytical models of M _{711 to} M ₈₂₁ , which are divided into 6 types as in the case of FIG. ● ”,
It is a scatter diagram plotted by "△", "○", and "x". That is, 0.8 to 2 GHz, 0.8 to 4 GHz,
0.8-6GHz, 0.8-8GHz, 0.8-10G
While determining whether the return loss S ₁₁ of each frequency band of Hz is -20 dB or less and classifying into 6 types,
The capacitance characteristic parameter α and the inductive characteristic parameter β in each analysis model are calculated, and M ₅₁₁ ~ on the α-β coordinate plane.
The return loss S ₁₁ of the analytical model of M ₆₁₉ is divided into 6 categories and plotted.

T ₇₁ , T ₇₃ , T ₇₄ and T in FIG.
₇₆ are straight lines represented by the formulas shown below, respectively.
It corresponds to the straight lines represented by the above equations (1) to (4) when ₇ = 3.6 mm. Further, T ₇₂ and T ₇₅ in FIG. 10 are also the straight lines represented by the formulas shown below, and the straight lines represented by the above formulas (5) and (6) when L ₇ = 3.6 mm. Corresponding to. T ₇₁ : α = 4 T ₇₃ : α = 18.67 T ₇₄ : β = 0.32 · α + 0.51 T ₇₆ : β = 0.32 · α + 2.29 T ₇₂ : α = 6.51 T ₇₅ : β = 0.32 · α + 0.89

From this, the return loss S ₁₁ is 0.8 to 6
The range of −20 dB or less in the frequency band of GHz is substantially within a range surrounded by a quadrangle having points P ₇₁ , P ₇₃ , P ₇₄ , and P ₇₅ as vertices, and this range is defined by the above formula (1) to (4)
Corresponds to the range surrounded by the straight line (L ₇ = 3.6 mm). Similarly, return loss S ₁₁ is 0.8-10G
The range below -20 dB in the frequency band of Hz is the point P.
₇₁ , P ₇₂ , P ₇₆ , and P ₇₇ are in a range surrounded by a quadrangle having vertices, and the straight lines (L ₇ = L ₇ = (4), (5), and (6)) (3.6 mm).

(Embodiment 4) Further, in each of the analytical models M ₁₃₃ , M ₁₅₄ and M ₂₅₂ shown in Table 1, the return loss S ₁₁ when the relative permittivity of the dielectric substrate 11 is changed.
I examined the change of. The dielectric constant of the dielectric substrate 11 is 6.0,
The return loss S ₁₁ was calculated by swinging to four levels of 6.5, 7.0 and 7.5. 11 (a) to 11 (c) show the results. In any case, it was found that there is no change in the classification result of the return loss S ₁₁ being −20 dB or less at 0.8 to 6 GHz and the classification result of being −20 dB or less at 0.8 to 10 GHz. Even if the thickness of the dielectric substrate 11 is changed within the range of 2 to 6 mm,
Classification result of -20 dB or less at 6 GHz, and 0.
It was found that there was no change in the classification result of -20 dB or less at 8 to 10 GHz.

As described above, in the high frequency flat circuit 10, the constricted portion 14 is provided longer than the cross width of the air bridge 17, and the dielectric constant of the dielectric substrate 11 is 6.0 to 7.5. The ground conductors 13a, 13b
In the range of 0.8 to 3.8 mm and the width of the center conductor 12 in the range of 0.6 to 2.4 mm.
And the relative permittivity ε _r is 4 to 17 and the thickness is 21 to
Since the spacer 18 is formed in the range of 40 μm,
It is possible to efficiently adjust the characteristic impedance and provide a high frequency plane circuit having stable transmission characteristics.

Although the high-frequency planar circuit of the present invention has been described in detail above, the present invention is not limited to the above-described examples and embodiments, and various improvements and changes are made without departing from the gist of the present invention. Of course it is okay.

[0061]

As described above in detail, in the high-frequency flat circuit, the constricted portion is provided longer than the cross width of the air bridge, and the dielectric constant of the dielectric substrate is 6.0. ˜7.5, the distance between the ground conductors is 0.8 to 3.8 mm, and the width of the center conductor 12 is 0.6 to
The transmission line is constructed in the range of 2.4 mm and the relative permittivity is 4
Since the spacer is formed in a range of -17 to 21 and a thickness of 21 to 40 μm, it is possible to efficiently adjust the characteristic impedance and provide a high-frequency planar circuit having stable transmission characteristics. In particular, it can be suitably used for a glass plate for automobiles, and can be easily manufactured in a large amount by using a known transfer-type printing pattern forming method.

[Brief description of drawings]

FIG. 1 is a perspective view showing an example of a high-frequency plane circuit of the present invention.

FIG. 2 is a cross-sectional view of the high-frequency planar circuit taken along the line A-B shown in FIG.

FIG. 3 is a cross-sectional view of the high-frequency planar circuit taken along the line XY shown in FIG.

FIG. 4 is a plan view of the high-frequency planar circuit shown in FIG.

5 is a diagram illustrating an analytical model of the high-frequency planar circuit shown in FIG.

6 (a) and 6 (b) are cross-sectional views of the mesh-divided analysis model shown in FIG.

7 is a scatter diagram showing an example of a result of return loss calculated using the analytical model shown in FIG.

8 is a diagram showing an example of frequency characteristics of return loss calculated using the analytical model shown in FIG.

9 is a scatter diagram showing another example of the result of the return loss calculated using the analytical model shown in FIG.

10 is a scatter diagram showing another example of the result of the return loss calculated using the analytical model shown in FIG.

11A to 11C are diagrams showing another example of frequency characteristics of return loss calculated using the analytical model shown in FIG.

[Explanation of symbols]

10 High frequency plane circuit 11 Dielectric substrate 12, 16 center conductor 13 Ground conductor 14 Constriction 15 Coplanar transmission line 17 Air Bridge 18 Spacer 19 slots 20 circuit model 22 analysis space

   ─────────────────────────────────────────────────── ─── Continued front page    (72) Inventor Koichiro Takahashi             1150 Hazawa-machi, Kanagawa-ku, Yokohama-shi, Kanagawa             Asahi Glass Co., Ltd. (72) Inventor Katsutoshi Nakayama             1150 Hazawa-machi, Kanagawa-ku, Yokohama-shi, Kanagawa             Asahi Glass Co., Ltd. F-term (reference) 5J014 CA02 CA08

Claims (3)

[Claims] 1. A dielectric substrate having a relative dielectric constant of 6.0 to 7.5, a center conductor provided on the dielectric substrate, and both sides of the center conductor so as to sandwich the center conductor with a space therebetween. A transmission line having a ground conductor provided on the adjacent dielectric substrate, and a transmission line having a constricted portion in which the width of the central conductor is partially narrowed, and a spacer made of a dielectric material provided on the constricted portion. A high-frequency planar circuit comprising: an air bridge that is held by the spacer and that crosses the central conductor above the constricted portion and that connects the ground conductors on both sides of the central conductor. The length of the constricted portion is longer than the transverse width of the air bridge, and the distance between the ground conductors of the transmission line is 0.8 to 3.8.
mm and the width of the central conductor other than the constricted portion is 0.6
Is about 2.4 mm, and the spacer has a relative dielectric constant of 4 to 17 and a thickness of 21.
A high-frequency planar circuit having a thickness of -40 μm. 2. A capacitance characteristic parameter relating to the capacitance characteristic of the transmission line due to the provision of the air bridge is α
And an induction characteristic parameter relating to the inductive property of the transmission line due to the provision of the constricted portion is β, an a-β coordinate plane having the capacitance characteristic parameter α on the horizontal axis and the inductive parameter β on the vertical axis. When determined, the transmission line and the bridge are configured such that the capacitive parameter α and the inductive parameter β are in a range surrounded by a straight line represented by the following formulas (1) to (4). The high frequency plane circuit according to claim 1. α = 4 (1) α = 5.5 / L ₇ ^1/2 +15.77 (2) β = 0.32 · α + 0.71 · L ₇ − 2.05 (3) β = 0.32 · α + 0.29 · L ₇ + 1.25 (4) However, the capacitance characteristic parameter α is α = ε _r · L ₃ ·
L ₄ / L _2, the inductive characteristic parameter beta _{is, β = L 5 · log e} (L
₇ / L ₄ ), L ₇ is the distance between the ground conductors (mm), ε _r is the relative permittivity of the spacer, L ₂ is the height of the air bridge (mm), and L ₃ is the air bridge. (Mm), L ₄ is the width (mm) of the central conductor of the constricted portion, and L ₅ is the length (mm) of the constricted portion. 3. On the α-β coordinate plane, the capacitive parameter α is within a range of a region surrounded by a straight line represented by the above formulas (1) and (4) and the following formulas (5) and (6). And so that the inductive parameter β exists,
The high frequency plane circuit according to claim 2, wherein the transmission line and the bridge are configured. α = 10.6 / L ₇ ^1/2 +0.92 (5) β = 0.32 · α + 0.58 · L ₇ − 1.2 (6)