JP2003284398A - Method for measuring constant of alternating-current motor and controller - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a method for constant measurement wherein it is unnecessary to correct an on-voltage drop every control period and set on-voltage drop characteristics of a power element with accuracy. <P>SOLUTION: The revolution of an alternating-current motor 2 is stopped by an inverter controller 8 for the alternating-current motor. In this stage, a voltage phase is fixed at a preset arbitrary value by a sine wave generator so that alternating magnetic flux is produced. A voltage command for a sine wave with a frequency of fh is given, and the voltage command value is adjusted so that the magnitude of an output current becomes a predetermined value. When a predetermined time has passed thereafter, the average value Vave of the absolute value of the voltage command and the average value Iave of the absolute value of the output current are computed and the phase difference θdif between the voltage command and the output current is measured by an average value-phase difference computing portion 27. Combined resistance (R1+R2) is computed from the cosine component of the ratio of Vave to Iave and measured. Then leakage inductance l=l1+l2 is computed from the sin component of the ratio of Vave to Iave and measured. <P>COPYRIGHT: (C)2004,JPO

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a combined primary and secondary resistance (R1 + R2) and a leakage inductance (l = l) of an AC motor used as a control constant in vector control or the like.
l1 + l2) is measured in detail, using an inverter device that variably controls the AC motor, the combined resistance (R1 + R2) of the AC motor and the leakage inductance (l
= L1 + l2) and an AC motor control device.

[0002]

2. Description of the Related Art As a conventional technique, a method of measuring a combined resistance (R1 + R2) and a leakage inductance (l = l1 + l2) of an AC motor with the AC motor stopped is disclosed in, for example, Japanese Patent Laid-Open No. 06-98595. The disclosed "constant measuring method and control device for AC electric motor" is known. FIG. 11 is a block diagram of a conventional constant current measuring device for an AC motor, and during normal operation, the inverter input voltage V
By performing PWM control of dc by the inverter 104, an AC voltage is generated and the induction motor 105 is controlled at a variable speed. Further, a control circuit 106 using a one-chip microcomputer or the like
Thus, during normal operation, the speed sensorless vector control processing 107 is performed so as to follow the speed command ωr, and the PWM signal is generated in the gate circuit 108. In this case, speed and torque control is performed based on the primary resistance measurement value r1, the secondary resistance value r2, the leakage inductance measurement value (l1 + l2), other motor constant setting values, and the output of the motor current detector 109. There is.

Therefore, the constants such as the combined resistance value (r1 + r2) and the leakage inductance (l1 + l2) are measured before the operation. First, a sine wave modulation signal for measurement is created by the single-phase AC excitation processing 110, and the gate circuit 108
In order to perform the measurement by operating the inverter 104 via the AC current and flowing an alternating current to the electric motor 105 by the alternating excitation voltage, the effective power current Iq reactive power current Id arithmetic processing circuit 111 is Letting θ be the rotational phase of the vector of the AC excitation voltage obtained by integrating the primary frequency command ω1 for measurement, Iq and Id are calculated based on sin θ, −cos θ and the U-phase motor current iu.
Next, in the resistance / inductance calculation processing circuit 112, I
From q and Id calculation values and the magnitude Vc1 of the excitation voltage command,
(R1 + r2) and (l1 + l2) are calculated.

[0004]

However, in the above-mentioned conventional example, by correcting the inverter output voltage error due to the ON voltage drop of the power element, the combined resistance is determined only from the magnitude of the excitation voltage command and the instantaneous current detection value of the motor. (R1 +
r2) and the leakage inductance (l1 + l2) are calculated and measured with high accuracy. Among them, the output voltage error due to the ON voltage drop of the power element changes depending on the magnitude of the output current. It was necessary to correct the ON voltage drop for the instantaneous value of. For this reason, there are problems that the software for measuring the constant becomes complicated and that the on-voltage drop of the power element needs to be set accurately in advance.

Therefore, in the present invention, the amplitude error and the phase error due to the ON voltage drop of the power semiconductor element are corrected to the average value V_ave of the absolute value of the voltage command after measurement and the phase difference θ_dif between the voltage command and the output current. Therefore, the combined resistance (R1 + R2) and the leakage inductance (l = l = 2) that do not require the correction of the ON voltage drop for each control cycle with respect to the instantaneous value and the setting of the accurate ON voltage drop characteristic of the power element are required.
It is an object of the present invention to provide a highly accurate calculation method of (11 + 12)) and a control device for an AC motor using the measured value.

[0006]

In order to achieve the above object, the invention according to claim 1 provides a power converter comprising a power semiconductor element capable of controlling the magnitude, frequency and phase of an output voltage. The primary current supplied via the AC motor is separated into a component (exciting current) parallel to the magnetic flux of the AC motor and a component (torque current) orthogonal to the magnetic flux, and each is independently adjusted to each phase of the AC motor. In the constant measurement method of an AC motor having a current detector that detects a flowing output current, the inverter device is in a state in which rotation of the AC motor is stopped, and a voltage phase so that an alternating magnetic flux is generated by a sine wave generator. Is fixed to a preset arbitrary value, a voltage command of a sine wave having a frequency fh is given, and the voltage command value is adjusted so that the magnitude of the output current becomes a predetermined value. The phase difference θ_dif average value I_ave and the voltage command and the output current of the absolute value of the average value V_ave and an output current of the absolute value is measured, the V_a
From the cos component of the ratio of ve and I_ave, the primary and secondary combined resistances (R1 + R2) are calculated and measured, and V_a
Calculate or calculate from the sin component of the ratio of ve and I_ave, or

[Equation 3] It is characterized in that more leakage inductance l = l1 + l2 is calculated and measured. According to a second aspect of the present invention, in the constant voltage measuring method for an alternating current motor according to the first aspect, a voltage command for measuring at least an amplitude and a phase difference error due to an ON voltage drop of a bower semiconductor element among output voltage errors. After being corrected to the average value V_ave of the absolute values of and the phase difference θ_dif between the voltage command and the output current, the combined resistance (R1 + R2) of the primary and secondary and the leakage inductance l
It is characterized in that the calculation and measurement of = l1 + l2 are performed. In the invention according to claim 3, the primary current supplied through a power converter composed of a power semiconductor element capable of controlling the magnitude, frequency and phase of the output voltage is supplied to the magnetic flux of the AC motor. Of an AC motor having a current detector for detecting an output current flowing in each phase of the AC motor by separating a component (exciting current) parallel to and a component (torque current) orthogonal thereto and adjusting each independently. In the control device, the inverter device fixes the voltage phase to a preset arbitrary value so that an alternating magnetic flux is generated by a sine wave generator in a state where the rotation of the AC motor is stopped, and the frequency fh
And a mean value V_ave of the absolute value of the voltage command after a predetermined time has elapsed, after adjusting the voltage command value so that the magnitude of the output current becomes a predetermined value.
A means for measuring the average value I_ave of the absolute value of the output current and the phase difference θ_dif between the voltage command and the output current is provided, and the primary and secondary combined resistances (R1 + R2) are calculated from the cos component of the ratio of V_ave and I_ave. Computation and measurement, and calculation or measurement from the sin component of the ratio of V_ave and I_ave, or

[Equation 4] It is characterized in that more leakage inductance l = l1 + l2 is calculated and measured. The invention described in claim 4 is the control device for an AC electric motor according to claim 3,
Out of the output voltage error, at least an amplitude and phase difference error due to ON voltage drop of the semiconductor device is provided with a means for correcting the average value V_ave of the absolute values of the measured voltage command and the phase difference θ_dif between the voltage command and the output current. The corrected V_ave and θ_dif are used to calculate and measure the primary and secondary combined resistance (R1 + R2) and the leakage inductance l = l1 + l2, and the AC motor is controlled based on these measured values.

[0007]

BEST MODE FOR CARRYING OUT THE INVENTION A first embodiment of the present invention will be described below with reference to the drawings. FIG. 1 is a circuit configuration diagram of a voltage type inverter according to a first embodiment of the present invention. FIG. 2 is a detailed circuit diagram of the inverter control device shown in FIG. FIG. 3 is a detailed circuit diagram of the average value / phase difference calculator shown in FIG. FIG. 4 is an equivalent circuit diagram of the electric motor shown in FIG. FIG. 5 is a flowchart of the operation of the voltage source inverter shown in FIG. FIG. 6 is a vector diagram of impedance of the equivalent circuit shown in FIG. FIG. 7 is a time chart of the voltage command value / current detection value of the average / phase difference calculator shown in FIG. In FIG. 1, 1 is a voltage source inverter, 2 is an AC motor, 3 is a current detector, 4 is a comparator, 5
Is an oscillator, 6 is an adder, 7 is a gate circuit, 8 is an inverter controller, 9 is a dead band compensator, 10 is a DC power supply, 11 is a speed command circuit, 14 is a speed detector, and 27 is an average value / phase calculation. It is a vessel. In the figure, the voltage source inverter 1 PWMs the DC voltage applied from the DC power supply 10.
It is converted into alternating current of arbitrary frequency and voltage according to the control method.
The voltage source inverter 1 is a switching element TUP or TV which is made of a power semiconductor element such as a transistor or IGBT.
.. TWN and feedback diodes DUP, DVP ,.

An AC motor 2 is connected to the AC output terminals of each phase U, V, W of the voltage source inverter 1. The U-phase, V-phase, and W-phase primary currents Iu, Iv, and Iw of the AC motor 2 are detected by the current detectors 3u, 3v, and 3w. Also,
The speed detector 14 detects the rotation speed of the AC motor 2. In the inverter control device 8, the speed command value ωr * created by the speed command circuit 11 and the current detectors 3u, 3v,
The U-phase, V-phase, and W-phase primary currents Iu, Iv, and Iw of the AC motor 2 detected by 3w and the speed detection value ωr from the speed detector 14 are added, and U, V having a 120 ° phase difference, W
Voltage command pattern signals (Vu ^* , Vv ^* , V
w ^* ) and the dead band compensators 9u, 9v, 9w for compensating the voltage error due to the dead band.
The dead band compensators 9u, 9v, 9W output the voltage error compensation value to the adders 6u, 6v, 6w using the phase γ as an argument.

The adders 6u, 6v, 6w add the voltage pattern signals Vu ^* , Vv ^* , Vw ^* and the output values of the dead band compensators 9u, 9v, 9w, and add the added values to the comparator 4u, Send to 4v, 4w. The output signal of the oscillator 5 that generates a carrier wave signal for PWM control is also compared with the comparators 4u and 4u.
v4w, the comparators 4u, 4v, and 4w compare the output signals of the adders 6u, 6v, and 6w with the carrier signal, and the switching element TUP that constitutes the voltage source inverter 1,
A PWM pulse for turning on / off the TVP and TWN is generated. The gate circuit 7 switches the switching element TUP in response to the PWM pulse output from the comparators 4u, 4v, 4w.
A gate signal is given to TVP, ... TWN.

In FIG. 2, 8 is an inverter control device,
12 is an exciting current command circuit, 15 is a 3-phase / 2-phase converter, 1
6 is a 2-phase / 3-phase converter, 17 is a primary angular frequency calculation circuit,
18 is a speed control circuit, 19 is a torque current control circuit, 20
Is an excitation current control circuit, 21 is a voltage command compensation circuit, 22 is an adder, 23 is an integrator, 24 is a phase calculator, 25 is a tuning processing unit, and 26 is a current amplitude calculator. In the figure, the inverter control device 8 includes a torque current feedback value Iqfb obtained by detecting the primary currents Iu, Iv, Iw to the AC motor 2 and performing coordinate conversion, and an exciting current feedback value Idfb.
A three-phase / two-phase converter 15 for transmitting the signal is provided. The output value of the speed control circuit ASR18 provided so that the speed command value ωr ^* from the speed command circuit 11 and the speed detection value ωr from the speed detector 14 are equal to each other is the torque current command value I.
qref, and this Iqref and the 3-phase / 2-phase converter 15
A torque current control circuit ACRq19 for controlling the torque current feedback value Iqfb output by
Excitation current command value Idref from the excitation current command circuit 12
And the exciting current feedback value Idf from the 3-phase / 2-phase converter 15
An exciting current control circuit ACRd20 for controlling the exciting current direction voltage is provided so as to match with b.

Further, the induced voltage generated by the magnetic flux of the AC motor 2 and the voltage of the component in the torque current direction of the counter electromotive force by the primary resistance R1 and the induced voltage by the induced electromotive force, and the leakage inductance (l = l1 + l2) of the AC motor 2 and It has a voltage command compensating circuit 21 that outputs a voltage of an exciting current direction component of the counter electromotive force by the primary resistance R1. The voltage of the torque current direction component of the output of the voltage command compensation circuit 21 is added to the output of the torque current control circuit 19 by the adder 22A to generate the torque current direction voltage command value Vqref, and the voltage of the excitation current direction component is , The output of the exciting current control circuit 20 and the adder 22
B is added to generate an exciting current direction voltage command value Vdref. Further, the torque current direction voltage command value Vqref,
Based on the excitation current direction voltage command value Vdref, a voltage command pattern signal (Vu of each phase of U, V, and W having a phase difference of 120 °).
A 2-phase / 3-phase converter 16 for generating and outputting ^{( *} , Vv ^* , Vw ^* ) is provided.

The three-phase / two-phase converter 15 and the two-phase / 3-phase converter 16 are calculated by the following equations (1) and (2), respectively.

[Equation 5] In addition, the inverter control device 8 uses Idref, Iqre
A primary angular frequency command calculation circuit 17 that obtains a slip frequency command value ωs ^* from the secondary resistance R2 set as f, calculates the primary angular frequency ω1 ^* from the detected speed value ωr from the speed detector 14, and outputs the calculated primary angular frequency ω1 ^*. The primary angular frequency ω1 ^* from the primary angular frequency command calculation circuit 17 is integrated by the integrator 23, and the 3-phase / 2-phase converter 15 and the 2-phase / 3-phase converter 16 are included.
To the phase θ. Further, the phase θ is tan ⁻¹ (Iqfb / Idfb) which is the output of the phase calculation circuit 24.
Is added by the adder 22c and output as the phase γ to the dead band compensators 9u, 9v, 9w.

In the inverter control device of FIG. 2, the tuning process forming the core part of the present invention is performed by the tuning process unit 25 for controlling the combined resistance and leakage inductance measuring operation before actual operation, and switching from the tuning process unit 25. Switch circuits 13A, 13B, and 13C that can switch between the actual operation and the constant measurement operation by the signal Csw are provided, and when the constant is measured before the actual operation, the switch circuit is switched to the b terminal side. In the switch circuit 13A, the torque current direction voltage command value Vqre
f is switched to 0, and in the switch circuit 13B, the excitation current direction voltage command value Vdref is set to the tuning voltage command value V_ref set for constant measurement in the tuning processing unit 25.
And the switch circuit 13C performs processing for switching the phase θ to a set value (fixed value). Therefore, as the output of the tuning processing unit 25 during the constant measurement, the switch switching signal Csw, the tuning voltage command value V ref set as the constant measurement signal, and the same phase θh are output, and the switch 13C arbitrarily sets the phase θ. Perform the process to set the fixed value of. Further, the current amplitude calculator 26 inputs the torque current feedback value Iqfb and the exciting current feedback value Idfb, and calculates and outputs the amplitude value I_fb of the output current.

In FIG. 3, 27 is an average value / phase calculator, 28 is an LPF, 29 is an HPF, 30 is an absolute value circuit,
Reference numeral 31 is a multiplier, 32 is a subtractor, and 33 is a sine wave generator. In the figure, the average value / phase difference calculator 27 indicates the signal V_ref from the inverter controller 8 during constant measurement,
I_fb and phase θh are input, and the phase difference θ_d of both signals
if and average values V_ave and I_av of absolute values of each signal
e for calculating high-pass filters HPF29A and 29B for removing DC components, a sine wave generator 33 for outputting sin and cos, an output of the HPF 29 and sin
Multiplying circuits 31A to 31D for multiplying θh and cosθh, low-pass filters LPF28A to F for performing average value processing,
It has absolute value calculation circuits (ABS) 30A and 30B and phase calculation circuits 24B and 24C.

The voltage source inverter 1 shown in FIG. 1 above.
2 and the average value / phase difference calculator 27 shown in FIG. 3, the outline of the constant measurement process is as follows. First, in the inverter control device 8 shown in FIG. Then, the tuning processing unit 25 switches the switches 13A, 13B, and 13C to the terminal b side for measurement by the switching signal Csw, and Vqref is set to 0 and Vdref is set.
Is set to the voltage command value V_ref for constant measurement and the phase θ is set to a fixed value, and the voltage command pattern signals Vu ^{* to} Vw ^* in the motor stopped state at the time of constant measurement are output, and the voltage type shown in FIG. The AC motor 2 is driven by the inverter to detect the primary currents Iu, Iv, Iw at the time of constant measurement. The same phase θh as the constant measurement voltage command V_ref and the current amplitude Ifb are output to the average value / phase difference calculator 27 in FIG. 3. Average value·
In the phase difference calculator 27, the phase difference θ_dif and the average values V_ave, I_av are calculated from the V_ref, θh, Ifb signals.
e is calculated, and a constant resistance calculator (not shown) calculates a combined resistance (R1 + R2) and a leakage inductance (l1 + l).
2) is sought and stored. When the constant measurement is completed, the switches 13A, 13B, and 13C in FIG. 2 are switched to return to the a terminal side for actual operation, and the obtained motor constant is used to control the inverter 1 shown in FIG. 1 to perform actual operation. Is.

Next, the constant measuring method of the present invention will be described with reference to FIG. At the time of constant measurement (at the time of tuning), the tuning processing unit 25 of the inverter control device 8 determines the magnitude and frequency fh of the constant current AC current flowing at the time of tuning (S100). The magnitude of the alternating current in this case is, for example, about 50% to 100% of the smaller one of the two rated current values based on the rated current values of the voltage source inverter 1 and the AC motor 2, and the frequency fh
Is a frequency at which 2π · fh · M >> R2 holds. FIG. 4A is a T-1 type equivalent circuit of an AC motor,
1, R2, l = l1 + l2, M is a primary resistance, 2
Regarding the next resistance, leakage inductance, and mutual inductance, when the operating frequency to be applied is high as in the constant measurement fh, ωM >> R2, so that almost no current flows in M and the equivalent circuit is shown in FIG. 4 (b). become that way. Further, the phase difference θ_dif between the voltage and the current at this time and (R1 + R2), (l
The relationship of (1 + l2) × 2πfh, Z is as shown in FIG.

Next, the tuning processing unit 25 sets Vqref to 0 in the switch circuit 13A, the tuning voltage command value V_ref in 13B, and the phase θ to a fixed value in 13C by the switching signal Csw during constant measurement (S101). ). Where V_ref = Vamp · sin
(2πfh · t), and the initial value of Vamp is 0, and the measurement operation is started. After that, Vamp (amplitude of V_ref) is increased so that the average value I_ave of the absolute values of the detected current values becomes the predetermined current value determined in S100. Then, when the value of I_ave reaches the predetermined value, a predetermined time elapses until the outputs of the LPFs 28A to 28F in the average value / phase difference calculator 27 are stabilized (S102).

The average current value I_ave is calculated by the absolute value circuit 30B and the average value processing circuit LPF28F from the input I_fb in the average value / phase difference calculator 27 shown in FIG. V_ave is the absolute value circuit 30A and LP
Required by F28E. Further, the phase differences θv ′ and θi ′ between the basic signal and each signal are obtained as the outputs of the phase calculation circuits 24B and 24C, and the difference θv′−θi ′ is obtained as the subtractor 32 output θ_dif. Next, the obtained voltage average value V_ave, current average value I_ave, and phase difference θ_di
The value of f is stored in each memory and the operation is stopped (S103). FIG. 7 shows how the voltage command V_ref and the detected current value I_fb change at this time. FIG. 7A shows how the output of the LPF stabilizes with the passage of time t, and FIG. 7B shows a relationship diagram of V_ref, I_fb and the phase difference θ_dif at that time.

When V_ave, I_ave, and θ_dif are obtained, the following equations (3), (4), and (5) are obtained.

[Equation 6] (R1 + R2) and (l1 + l2) are calculated (S10
4). Further, from the equation (3), the equation (4), and the equation (5), the impedance of the circuit is Z = (V_ave / √3).
If the angle between / I_ave, Z and the real axis Re is θ_dif, it can be obtained from the relationship shown in the figure.

According to the first embodiment described above, the measurement was started so far with the initial value of Vamp set to 0, but during actual operation, the flowing current value is changed from the V / f characteristic value of the AC motor. It is also possible to shorten the time by predicting, setting some values in advance, and adjusting from there. Further, in FIG. 3, the average value V_av of each signal is
Although e and I_ave are obtained by the LPFs 28E and 28F, they may be obtained by a moving average or V_av.
e, the phase θV ′ is the tuning voltage command value V_ref
Since the amplitude, frequency, and time constant of the LPF 28E are self-explanatory, they can be calculated. Further, the tuning voltage command value V_ave is added to the offset value V_ref_ofs.
Assuming that the voltage command value is the voltage added with, the offset voltage is output as direct current, and therefore the primary resistance R1 can be simultaneously obtained from this direct current. If R1 is found,
It is easy to obtain R2 from (R1 + R2). Further, in the present embodiment, the combined resistance and the leakage inductance are measured with the AC signal V_ref having one kind of amplitude and frequency, but the measurement is performed under a plurality of conditions, and the average of the measurement results and the influence of the offset amount are measured. The present invention can be implemented by calculating using the difference amount in order to avoid it.

Next, a second embodiment of the present invention will be described with reference to the drawings. FIG. 8 is a flowchart of the operation of the voltage source inverter according to the second embodiment of the present invention. FIG. 9 is a diagram showing an ON voltage drop amount of the voltage source inverter shown in FIG. FIG. 10 is an explanatory diagram of the correction process based on the ON voltage drop amount shown in FIG. In the first embodiment, since the voltage drop of the power semiconductor element is not corrected, an error may occur in the measured V_ave and θ_dif. Therefore, in the second embodiment, V_ave is measured.
And means for correcting the values of θ_dif (not shown) are provided in the average value / phase difference calculator 27, and as shown in FIG.
The correction processing of S105 and S106 is added to the processing of the first embodiment.

In FIG. 8, S100 to S104.
Since the processing of 1 is the same as that of the first embodiment, duplicate description will be omitted. First, V_ave measured by the processing of S100 to S103 is corrected (S105). Specifically, the peak value Ipk of the measured alternating current is calculated as the average value of the current ×
As π / 2, the instantaneous current value when the current phase is θ_dif is obtained as Ipk × sin θ_dif. next,
The average value Von_ave of the on-state voltage drop amount of the power semiconductor element at each current value is calculated by the following equation (6) to obtain V_ave
To Von_ave to obtain a new V_ave.

Von_ave = (A + B) / 2 × (90−θ_dif) / 90 (6) where A = 2/3 × [Von (Ipk) + Von (Ip
k / 2)] B = 2/3 × [Von (Ipk × sin θ_dif) +
Von (Ipk / 2 × sin θ_dif)] Note that Von (I) represents the on-voltage drop amount of the power semiconductor element when the current I is flowing, and as shown in FIG. Calculated as an average value with the amount of drop, and the average value as a function of the current I
It is built in the phase calculator 27.

Next, the phase difference θ_dif is corrected (S
106). Specifically, the phase correction amount θcmp is calculated by the following equation (7), and θcmp is added to θ_dif to obtain a new θ_dif. θ_dif = K × 4/3 × Von (Ipk) / V_ave × θ_dif / 90 (7) In the equation (7), K is the on-voltage drop amount 4 when the voltage command value V_ave and the peak current value Ipk are 4 / 3 x Von
It is a proportional coefficient for correction of the phase angle, which is determined by the ratio with (Ipk).

In the above equations (6) and (7), (90-θ_
dif) / 90 and θ_dif / 90 indicate that the remaining portion that is canceled by averaging the influence of the ON voltage drop of the power semiconductor element is averaged in 90 ° phase units. This relationship is shown in FIG. Figure 10 (a)
It can be seen that the influences in the areas A1 and A2 of FIG. 10B and the areas B1 and B2 of FIG. 10B are canceled out, that the area A1 matches θ_dif, and that the area B1 matches 90-θ_dif. Further, as can be seen from FIG. 10A, since the influence of the voltage error near the current 0 is canceled out, it is not necessary to know the characteristic of the ON voltage drop amount at the minute current, and the influence of the ON voltage component can be obtained. It becomes possible to correct. Finally,
The control constant corrected by the ON voltage amount is calculated in S104, and the measurement is ended (S104).

As described above, according to the present invention, R2 and (l = l1 + l2) are obtained from the measured combined resistance and leak inductance measurement values and the preset value of the primary resistance R1 to calculate the primary angular frequency. The circuit 17 and the voltage command compensating circuit 21 are set, and the switch circuits 13A to 1
By switching 3C back to the a terminal side, the measured resistance,
Vector control can be performed by a control constant such as leakage inductance. Further, although the implementation target has been described so far as the induction motor with the speed detector, the invention can be applied to an induction motor without the speed detector or a synchronous machine. That is, it is possible to create a frequency command from a speed estimator or a speed command value instead of the speed detector, or to compensate for the slip frequency command value ωs *, and to measure the resistance and leakage inductance measured by the speed estimator. The control constant of can be set. Needless to say, the combined resistance and leakage inductance measuring method disclosed by the present invention can be used without any problem even if the control method during actual operation is different.

[0027]

As described above, according to the present invention,
Combined resistance (R1 + R2), leakage inductance (l = 1
1 + l2), etc., the output voltage detector of the inverter is unnecessary, and the combined resistance and leakage inductance can be measured and calculated with the electric motor stopped. By correcting the effect of the on-voltage drop to the average value V_ave of the voltage command and the phase difference θ_dif between the voltage command and the output current, the on-voltage drop of the power element in consideration of the instantaneous correction of the on-voltage drop for each control cycle and the accuracy. There is an effect that it is possible to easily perform highly accurate measurement and calculation of control constants without setting the characteristics.

[Brief description of drawings]

FIG. 1 is a circuit configuration diagram of a voltage type inverter according to a first embodiment of the present invention.

FIG. 2 is a detailed circuit diagram of the inverter control device shown in FIG.

FIG. 3 is a detailed circuit diagram of the average value / phase difference calculator shown in FIG.

4 is an equivalent circuit diagram of the AC motor shown in FIG.

5 is a flowchart of the operation of the voltage source inverter shown in FIG.

FIG. 6 is a vector diagram of impedance of the equivalent circuit shown in FIG.

7 is a time chart of the voltage command value / current detection value of the average value / phase difference calculator shown in FIG.

FIG. 8 is a flowchart of the operation of the voltage source inverter according to the second embodiment of the present invention.

9 is a diagram showing an on-voltage drop amount of the voltage source inverter shown in FIG.

FIG. 10 is an explanatory diagram of a correction process based on the ON voltage drop amount shown in FIG.

FIG. 11 is a configuration diagram of a conventional constant current measuring device for an AC motor.

[Explanation of symbols]

1 voltage source inverter 2 AC motor 3 Current detector 4 comparator 5 oscillators 6 adder 7 gate circuit 8 Inverter control device 9 Dead band compensator 10 DC power supply 11 Speed command circuit 12 Excitation current command circuit 13 switch circuit 14 Speed detector 15 3 phase / 2 phase converter 16 2 phase / 3 phase converter 17 Primary angular frequency calculation circuit 18 Speed control circuit 19 Torque current control circuit 20 Excitation current control circuit 21 Voltage command compensation circuit 22 adder 23 Accumulator 24 Phase calculator 25 Tuning processing section 26 Current amplitude calculator 27 Average value / phase calculator 28 LPF 29 HPF 30 Absolute value circuit 31 Multiplier 32 subtractor 33 Sine wave generator

   ─────────────────────────────────────────────────── ─── Continued front page    F term (reference) 5H576 BB06 BB07 CC01 DD02 DD04                       DD05 DD07 EE01 EE11 FF05                       GG02 GG04 HA02 HB02 JJ03                       JJ04 JJ17 JJ26 KK06 LL01                       LL22 LL29 LL40

Claims (4)

[Claims] 1. A primary current supplied through a power converter composed of a power semiconductor element capable of controlling the magnitude, frequency and phase of an output voltage, a primary current supplied to a component parallel to the magnetic flux of the AC motor (excitation). Current) and a component (torque current) orthogonal to the current) and independently adjusting each of them to detect the output current flowing in each phase of the AC motor. The apparatus fixes the voltage phase at an arbitrary preset value so that an alternating magnetic flux is generated by the sine wave generator while the rotation of the AC motor is stopped, and the frequency fh
Of the sine wave, the voltage command value is adjusted so that the magnitude of the output current becomes a predetermined value, and after a lapse of a predetermined time, the average value V_ave of the absolute values of the voltage command and the absolute value of the output current are The average value I_ave and the phase difference θ_dif between the voltage command and the output current are measured, and V_ave is measured.
And the combined resistance of primary and secondary (R1 + R2) is calculated and calculated from the cos component of the ratio of I_ave to V_ave.
Or calculate from the sin component of the ratio of I_ave and
Or the following equation, According to the method, the leakage inductance l = l1 + l2 is calculated and measured. 2. The constant value measuring method for an AC motor according to claim 1, wherein at least an amplitude and a phase difference error due to an ON voltage drop of a bower semiconductor element among output voltage errors are average values of absolute values of measured voltage commands. After correction to V_ave and the phase difference θ_dif between the voltage command and the output current, the combined primary and secondary resistance (R1 + R2) and the leakage inductance l = 1
A method for measuring constants of an AC motor, characterized by calculating and measuring 1 + 12. 3. A primary current supplied through a power converter composed of a power semiconductor element capable of controlling the magnitude, frequency and phase of an output voltage, a primary current supplied to a component parallel to the magnetic flux of the AC motor (excitation). Current) and a component (torque current) orthogonal to the current) and independently adjusting each of them, and having a current detector for detecting an output current flowing in each phase of the AC motor. Is a voltage phase fixed to an arbitrary value set in advance so that an alternating magnetic flux is generated by a sine wave generator while the rotation of the AC motor is stopped, and the frequency f
A means for giving a voltage command of a sine wave of h, and a voltage command value is adjusted so that the magnitude of the output current becomes a predetermined value, and after a predetermined time elapses, an average value V_ave of absolute values of the voltage command.
And a means for measuring the average value I_ave of the absolute value of the output current and the phase difference θ_dif between the voltage command and the output current, and the cos of the ratio of V_ave and I_ave.
The combined resistance (R1 + R2) of the primary and the secondary is calculated and measured from the component, and the ratio of V_ave and I_ave is calculated as sin.
Compute and measure from the component, or the following equation, A control device for an AC motor, characterized in that the leakage inductance l = l1 + l2 is calculated and measured. 4. The control device for an AC motor according to claim 3, wherein an average value V_ave of absolute values of measured voltage commands of at least an amplitude and a phase difference error due to an ON voltage drop of a power semiconductor element among output voltage errors. And a means for correcting the phase difference θ_dif between the voltage command and the output current, and using these corrected V_ave and θ_ave, the primary and secondary combined resistances (R1 + R2) and the leakage inductance l = l1 + l2 are calculated and measured. A control device for an AC motor, which controls the AC motor based on a measured value.