JP2003033031A - Switching power supply - Google Patents

Abstract

PROBLEM TO BE SOLVED: To return energy stored in an inductance to a power supply through simple circuitry to make applicable switching semiconductor devices of low breakdown voltage. SOLUTION: When a switching semiconductor device 3 is off, energization is performed in the forward direction by a charge-storage diode 4 to store energy stored in an inductance means 2 in a voltage clamp capacitor 5. When the current flowing in the forward direction is zeroed, energization is performed in the reverse direction to return the energy stored in the voltage clamp capacitor 5 to a direct-current power supply 1 through the inductance means 2. Thus, of energy charged in the voltage clamp capacitor 5, energy which was not discharged by reverse-direction energization of the charge-storage diode 4 is discharged by forward-direction energization of the charge-storage diode 4 through a discharge circuit 11.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply provided with a snubber circuit for protecting a switching semiconductor element from a surge voltage generated by a sudden change in voltage or current during switching of the switching semiconductor element.

[0002]

2. Description of the Related Art As a typical conventional switching power supply, there is a circuit configuration as shown in FIG. In this switching power supply, the DC power supply 1, the primary winding of the transformer 2 and the M
A switching semiconductor element 3 such as an OSFET is connected in series. The first diode 4 and the first capacitor 5, which are connected in series with each other, are connected in parallel across the switching semiconductor element 3 in the direction shown. On the secondary winding side of the transformer 2, a rectifying diode 6, a freewheeling diode 7, a smoothing inductor 8,
The smoothing capacitor 9 and the load 10 are connected. In addition, 11 "is a resistor connected between the cathode side of the first diode 4 and the positive electrode of the DC power source 1, and 12 is a parasitic capacitance of the switching semiconductor element 3 or is connected in parallel with the parasitic capacitance. A synthetic capacitance of a capacitor, 13 is a control circuit, 14 is a second diode connected in parallel to the switching semiconductor element 3, or the switching semiconductor element 3 is a MOSFET.
The body drain diode in the case of is shown.

Next, a typical operation of this circuit will be described with reference to FIG.

Period 1 (t1 <t ≦ t2) At time t1, when the switching semiconductor element 3 is turned off,
The current flowing through the switching semiconductor element 3 until then flows into the capacitor 12,
When the voltage V1 of 1 is rapidly increased and becomes equal to the voltage Ei of the DC power supply 1, the voltage applied to the winding of the transformer 2 becomes zero. During this period, the voltage V1 of the switching semiconductor element rises almost linearly. When the winding voltage of the transformer 2 becomes zero, the freewheeling diode 7, which has been reverse-biased by the voltage of the transformer 2 until now, becomes conductive, so that the load current is rectified with the freewheeling diode 7 and rectified. It flows through the diode 6 and puts the secondary winding of the transformer 2 in a short-circuited state. This time is set to t2.

Period 2 (t2 <t ≦ t3) Since the secondary winding of the transformer 2 is short-circuited at time t2, the exciting current of the transformer 2 maintains the value at time t2 and is kept constant. The primary-side converted current of the load current and the exciting current have flowed into the capacitor 12 until then, but the load current is reduced because it shifts to the freewheeling diode 7. However, due to the leakage inductance of the transformer 2 and the inductance of the wiring, it does not become zero immediately. As the charging current continues to flow through the capacitor 12, the voltage V1 of the switching semiconductor element 3 continues to rise,
When the voltage of the second capacitor 5 is reached, the first diode
Do 4 starts conducting. This time is set to t3.

Period 3 (t3 <t ≦ t4) At time t3, the voltage V1 of the switching semiconductor element 3 becomes equal to that of the first capacitor 5 due to the conduction of the first diode 4.
Since the capacitance is selected to be sufficiently larger than the capacitance of the capacitor 12, it is clamped by the voltage of the first capacitor 5. When the charging currents of the second capacitor 12 and the first capacitor 5 decrease to the value of the exciting current of the transformer 2, the rectifying diode 6 becomes non-conducting and the short-circuited state of the secondary winding of the transformer 2 is released. To be done. This time is t4.

Period 4 (t4 <t ≦ t5) At time t4, the short-circuited state of the secondary winding of the transformer 2 is released, and the exciting current kept constant starts to decrease. Further, the voltage of the difference between the first capacitor 5 and the power supply voltage is applied to the transformer 2 so that the exciting current is reset.
The voltage V1 of the switching semiconductor element 3 is clamped by the voltage of the first capacitor 5 following the period 3. The time at which the exciting current of the transformer 2 becomes zero is t5.

Period 5 (t5 <t ≦ t6) At time t5, when the charging currents of the capacitors 12 and 5 become zero, the diode 4 becomes non-conductive,
A discharge current starts flowing from the capacitor 12 through the primary winding of the transformer 2 and the loop of the DC power supply 1. After that, the switching semiconductor element 3 is turned on by a control signal from the control circuit 13 that controls so as to stabilize the output voltage. This time is set to t6.

Period 6 (t6 <t ≦ t8) At time t6, when the switching semiconductor element 3 is turned on, the rectifying diode 6 becomes conductive. Since the rise of the current of the rectifying diode 6 is limited by the leakage inductance of the transformer 2, the freewheeling diode 7 is continuously turned on, and the secondary winding of the transformer 2 is short-circuited. . At this time, since the DC power supply voltage Ei is borne by the leakage inductance of the transformer 2, the current of the switching semiconductor element 3 and the current of the rectifying diode 6 increase linearly. When the current in the rectifying diode 6 becomes equal to the current in the smoothing inductor 8, the freewheeling diode 7 becomes non-conductive. This time is t7
And When the freewheeling diode 7 becomes non-conductive, electric power is supplied from the DC power source 1 to the secondary side of the transformer 2 via the switching semiconductor element 3, the transformer 2, and the rectifying diode 6. This period continues until the switching semiconductor element 3 is turned off. This time is t8.

After that, the operation of the period 1 is returned to, and the above-described operation is repeated.

In this circuit, the switching semiconductor element 3 is turned on after the first diode 4 is turned off in the period 5 as described above. At this time, the first diode 4
If the reverse conduction is not completed, a short-circuit current flows through the loop of the first diode 4, the first capacitor 5 and the switching semiconductor element 3, which causes an increase in loss and noise. Therefore, in this circuit, a diode having a short carrier lifetime is required as the first diode 4.

Further, since the same current as the average value of the currents flowing into the first capacitor 5 is discharged through the resistor 11 ″, the loss of the resistor 11 ″ becomes large. However,
In such a switching circuit, since the charged power is returned to the DC power source through the resistor 11 ″, the loss of the resistor 11 ″ becomes large, which causes a decrease in efficiency.

[0013]

In order to achieve the technical idea of the present invention, the present invention provides a charge storage type diode having a relatively long carrier lifetime and an inductance means during forward conduction of the diode. The energy is stored and the energy is returned to the power supply when the diode is conducting in the reverse direction, and the voltage clamping means for clamping the voltage across the switching element and the portion of the charge storage diode that cannot be discharged when the diode is conducting in the reverse direction. By including a discharge circuit that discharges energy,
An object of the present invention is to provide a switching power supply that can reduce power loss without increasing the number of circuit components and a control method thereof.

[0014]

[Means for Solving the Problems] In order to solve the above problems, in the invention of claim 1, a switching semiconductor element for selectively opening and closing a current path from a DC power supply to a load, and a switching semiconductor element are provided. Inductance means connected in series, a charge storage diode connected in parallel to the switching semiconductor element and connected in series with each other, a voltage clamp capacitor that acts to clamp the voltage of the switching semiconductor element, and the charge storage -Type diode and the voltage clamp capacitor, a discharge circuit connected between the DC power source and a connection point, and a control circuit that controls switching of the switching semiconductor element, the charge storage diode, the switching circuit. When the semiconductor device is off, the semiconductor device first conducts in the forward direction and the The energy stored in the rectifying means is stored in the voltage clamp capacitor, and when the current flowing in the forward direction becomes zero, the energy is stored in the voltage clamp capacitor by conducting in the reverse direction. Returning to the power supply, the discharge circuit discharges energy that has not been released by the reverse conduction of the charge storage diode among the energy charged in the voltage clamp capacitor by the forward conduction of the charge storage diode. A characteristic switching power supply is provided.

In order to solve the above problems, in the invention of claim 2, a switching semiconductor element that selectively opens and closes a current path from a DC power source to a load, and an inductance connected in series to the switching semiconductor element. Means, a charge storage diode and a voltage clamp capacitor connected in parallel to the inductance means and connected in series with each other, between a connection point between the charge storage diode and the voltage clamp means, and the DC power supply. The charge storage diode includes a connected discharge circuit and a control circuit that controls switching of the switching semiconductor element, and the charge storage diode is first conducted in the forward direction to the inductance means while the switching semiconductor element is off. Store the stored energy in the voltage clamp capacitor, then in the forward direction When the flowing current becomes zero, it conducts in the reverse direction and returns the energy stored in the voltage clamp capacitor to the DC power supply through the inductance means, and the discharge circuit forwards the charge storage diode to the voltage clamp. There is provided a switching power supply characterized by discharging, of the energy charged in a capacitor, the energy not released by reverse conduction of the charge storage diode.

In order to solve the above-mentioned problem, in the invention of claim 3, the energy of the voltage clamp capacitor according to claim 1 or 2 is set so that the voltage of the voltage clamp capacitor does not exceed a set voltage. A switching power supply is provided with a discharge circuit whose amount can be adjusted in parallel with the voltage clamp capacitor.

In order to solve the above-mentioned problems, in the invention of claim 4, in the invention of claim 1 or 2, the discharge circuit generates a signal synchronized with a drive signal for driving the switching semiconductor element. A switching power supply having a variable impedance corresponding to a frequency is provided.

In order to solve the above-mentioned problem, in the invention of claim 5, in any one of claims 1 to 4, the switching semiconductor element has completed reverse conduction of the charge storage diode. Provide a switching power supply that is turned on later.

[0019]

BEST MODE FOR CARRYING OUT THE INVENTION A first embodiment according to the present invention will be described with reference to FIGS. 1 to 3. This embodiment has the circuit configuration shown in FIG. 1, and the connections between the circuit components are as shown. In FIG. 1, the same symbols as those referred to in FIG. 11 indicate corresponding circuit components. The first diode 4 is a diode having a relatively long carrier lifetime as compared with a high speed diode (hereinafter referred to as a charge storage diode). A charge storage diode having a relatively long carrier lifetime has the characteristic of essentially holding reverse conduction for a long time as compared with a short one, but if a carrier equal to the storage carrier is injected from the reverse direction. , The reverse blocking ability of the diode is restored. The present invention is based on such new findings.

The voltage clamp capacitor 5 acts to clamp the voltage applied across the switching semiconductor element 3. For the gate of the switching semiconductor element 3,
A control circuit 13 for outputting an ON signal to the switching semiconductor element 3 is connected after the reverse direction of the charge storage diode 4 is substantially restored or when the voltage of the switching semiconductor element 3 becomes substantially zero volt. . A discharge circuit 11 is connected across the connection point between the charge storage diode 4 and the voltage clamp capacitor 5 and one end of the DC power supply 1. This discharge circuit will be described later with reference to FIG.

Next, a typical operation of this embodiment will be described with reference to the waveforms of the respective parts in FIG. Period 1 (t1 <t ≦ t2) At time t1, when the switching semiconductor element 3 is turned off,
The current flowing through the switching semiconductor element 3 until then flows into the capacitor 12,
When the voltage V1 of 1 is rapidly increased and becomes equal to the voltage Ei of the DC power supply 1, the voltage applied to the winding of the transformer 2 becomes zero. During this period, the voltage V1 of the switching semiconductor element rises almost linearly. When the winding voltage of the transformer 2 becomes zero, the freewheeling diode 7, which has been reverse biased by the voltage of the transformer 2 until now, becomes conductive,
The load current flows through the fuwheeling diode 7 and the rectifying diode 6 to short-circuit the secondary winding of the transformer 2.
This time is set to t2.

Period 2 (t2 <t ≦ t3) Since the secondary winding of the transformer 2 is short-circuited at time t2, the exciting current of the transformer 2 holds the current value at time t2 and keeps it constant. Be drunk The primary-side converted current of the load current and the exciting current have flowed into the capacitor 12 until then, but the load current is reduced because it shifts to the freewheeling diode 7. However, due to the leakage inductance of the transformer 2 and the inductance of the wiring, it does not become zero immediately. As the charging current continues to flow through the capacitor 12, the voltage V1 of the switching semiconductor element 3 continues to rise, and when the voltage of the voltage clamp capacitor 5 is reached, the charge storage diode 4 starts conducting. This time is set to t3.

Period 3 (t3 <t ≦ t4) At time t3, the charge storage diode 4 becomes conductive,
The voltage V1 of the switching semiconductor element 3 is clamped by the voltage of the voltage clamp capacitor 5 because the capacity of the voltage clamp capacitor 5 is selected to be sufficiently larger than the capacity of the capacitor 12. When the charging current of the second capacitor 12 and the voltage clamp capacitor 5 decreases to the value of the exciting current of the transformer 2, the rectifying diode 6 becomes non-conductive, and the short-circuited state of the secondary winding of the transformer 2 is released. To be done. This time is t4.

Period 4 (t4 <t ≦ t5) At time t4, the short-circuited state of the secondary winding of the transformer 2 is released, and the exciting current that has been kept constant starts to decrease.
Further, the voltage of the difference between the voltage clamp capacitor 5 and the power supply voltage is applied to the transformer 2, and the exciting current is reset. The voltage V1 of the switching semiconductor element 3 is clamped by the voltage of the voltage clamp capacitor 5 after the period 3. The time when the exciting current of the transformer 2 becomes zero is t
Set to 5.

Period 5 (t5 <t ≦ t6) At time t5, when the charging current of the voltage clamp capacitor 5 becomes zero, the forward conduction of the charge storage diode 4 ends. The charge storage diode 4 becomes reverse conduction due to the electric charge accumulated in the junction portion during forward conduction and remaining without being recombined, so that the discharge current of the voltage clamp capacitor 5 is changed to the charge storage diode 4, Transformer 2, DC power supply 1,
And a loop consisting of a voltage clamp capacitor 5,
The energy stored in the voltage clamp capacitor 5 is collected in the DC power supply 1 when the charge storage diode 4 is turned on in the forward direction. This period ends when the charge remaining at the junction of the charge storage diode 4 becomes zero. This time is t6
And

Here, the ratio of the forward current and the reverse current of the charge storage diode 4 is taken as the power recovery rate, and the characteristics of the charge storage diode 4 necessary for the present invention will be described with reference to FIG. FIG. 3 shows the power recovery rate characteristics of the charge storage diode 4 with respect to carrier lifetime / switching cycle. It can be seen from FIG. 3 that when a diode having a carrier lifetime equal to or longer than the switching period is used, the power recovery rate sharply improves, which is preferable, but even if the carrier lifetime is shorter than that, the power consumption is reduced. It is clear that recovery is possible.

When the power recovery rate becomes poor, the voltage across the voltage clamp capacitor 5 and the switching semiconductor element 3 becomes high, and the discharge circuit 11 discharges the energy of the voltage clamp capacitor 5 in order to avoid it. In some cases, the power loss will increase.

Therefore, in order to efficiently realize the discharge of the electric power from the voltage clamp capacitor 5 due to the reverse conduction of the charge storage diode 4, a charge storage diode having a carrier lifetime longer than the switching period is used. However, from the above knowledge and the basic technical idea of the present invention, the electric power can be recovered even by the charge storage type diode having the carrier lifetime shorter than the switching period.

Further, when the carrier lifetime of the charge storage diode 4 is about the same as the time corresponding to the switching cycle, even if the unrecovered power of the voltage clamp capacitor 5 is discharged by the discharge circuit 11, that power is not recovered. The loss becomes very small with respect to the output power, and the above operation can be obtained without deteriorating the conversion efficiency.

Period 6 (t6.ltoreq.t.ltoreq.t7) When the reverse conduction of the charge storage diode 4 ends at time t6, the resonance circuit is formed by the exciting inductance of the capacitor 12, the transformer 2, the leakage inductance and the wiring inductance. Once formed, the capacitor 12 is further discharged. Although the effect of the present invention is not impaired even if the switching semiconductor element 3 is turned on after the reverse conduction of the charge storage diode 4 is completed, the voltage of the capacitor 12 is set to zero by selectively designing the constant. , The anti-parallel diode 14 (MOS) of the switching semiconductor element 3 can be discharged.
In the case of a FET, when the body diode of the FET is conductive, the switching semiconductor element 3 is made conductive so that zero voltage switching becomes possible. Here, the operation when zero voltage switching is possible will be described. When the voltage V1 of the switching semiconductor element 3 becomes equal to the voltage of the DC power source 1, the voltage of the transformer 2 becomes almost zero, and the rectifying diode is turned on.
The secondary winding of the transformer 2 is short-circuited by the rectifying diode 6 and the freewheeling diode 7. Transformer 2 is reset. This time is t7. In the period from t4 to t6, the current flowing through the primary winding of the transformer 2 is the exciting current of the transformer 2, so that the current does not substantially flow through the secondary winding.

Period 7 (t7 <t ≦ t10) The voltage of the capacitor 12 is further discharged by the resonant circuit of the leakage inductance of the transformer 2, the inductance of the wiring, and the capacitor 12, and when it reaches zero volts, the switching semiconductor element 3 Turns on. This time is t
8 Since the rectifying diode 6 conducts and the winding of the transformer 2 is short-circuited, most of the DC power supply voltage Ei is borne by the leakage inductance of the transformer 2, and the current increases linearly in the forward direction. . Since the leakage inductance of the transformer 2 is small, the currents of the switching semiconductor element 3 and the rectifying diode 6 increase rapidly, and when the current of the rectifying diode 6 becomes equal to the current of the inductor 8 at time t9, a free current is generated. Wheeling diode 7 is reverse biased and becomes non-conductive. Free Wheeling Dio
When the switch 7 becomes non-conductive, a voltage converted from the voltage Ei of the DC power supply 1 in the number of turns appears in the secondary winding of the transformer 2, and the switching semiconductor element 3, the transformer 2, and the rectifying diode 6 are included.
Power is supplied from the DC power supply 1 to the secondary side of the transformer 2 via the. This period continues until the switching semiconductor element 3 is turned off. This time is t 10.

After that, the operation of the period 1 is returned to, and the same operation as described above is repeated. The waveform of each part is as shown in FIG.

As described above and shown in FIG. 3, even when the carrier lifetime of the charge storage diode 4 is shorter than the switching cycle, part of the electric power charged in the voltage clamp capacitor 5 is returned to the power supply. . As the charge storage diode 4, a diode having a large leak current can be used.

Next, referring to FIG. 4, a second embodiment of the present invention will be described. The same symbols as those shown in FIG. 1 indicate corresponding members. FIG. 4 is a circuit diagram of a push-pull type forward converter, which includes a charge storage diode 4 ′, a second capacitor 12 ′, a voltage clamp capacitor 5 ′, and a discharge circuit 11 ′. , The second capacitor 12, the voltage clamp capacitor 5, and the discharge circuit 11 are connected in parallel to the switching semiconductor element 3 ′. The charge storage diode 4 ′ is similar to the charge storage diode 4. Since the main operation is almost the same as that of the embodiment shown in FIG. 1, the description of the operation will be omitted.

Next, a third embodiment of the present invention is shown in FIG. It is a non-insulated switching power supply in which the transformer 2 in the embodiment shown in FIG. 1 is replaced with a resonance inductor 2'that forms a resonance circuit with a second capacitor. The charge storage diode 4 is a diode having a carrier lifetime similar to that described above. Since the main operation is almost the same as that of the embodiment shown in FIG. 1, the description of the operation will be omitted. In FIG. 5, the same symbols as those shown in FIG. 1 indicate corresponding members.

Next, FIGS. 6A and 6B show fourth and fifth embodiments of the present invention, respectively. FIG. 6A shows a series connection circuit of the charge storage diode 4 and the voltage clamp capacitor 5 connected in parallel with the inductance of the transformer 2 in the embodiment shown in FIG. It will be connected in parallel with the discharge circuit 11. Further, FIG. 6B shows a series connection circuit of the charge storage diode 4 and the voltage clamp capacitor 5 connected in parallel with the resonance inductor 2 ′ in the embodiment shown in FIG. The clamp capacitor 5 will be connected in parallel with the discharge circuit 11. The charge storage diode 4 may have a carrier lifetime that is about the same as or shorter than the switching cycle of the switching semiconductor element 3. As for the main operation, the circuit of FIG. 6A is almost the same as the embodiment shown in FIG. 1, and the circuit of FIG. 6B is almost the same as the embodiment shown in FIG. The description of the operation is omitted. Note that FIG.
In FIG. 1, the same symbols as those shown in FIG. 1 or 5 indicate corresponding members. Here, the voltage clamp capacitor 5 directly clamps the primary winding voltage of the transformer 2, but this clamps the voltage across the switching semiconductor element 3.

Next, one embodiment of the discharge circuit 11 is shown in FIG.
It shows in (1)-(4). Terminals 11-a and 1 in FIG.
1-b are terminals 11-a and 11- of the discharge circuit 11 of FIG.
Each corresponds to b.

FIG. 7A shows the charge storage diode 4
In this method, the unrecovered power of the voltage clamp capacitor 5 that was not recovered during the reverse conduction of (1) is returned to the DC power supply 1 while being consumed by the resistor 17.

In FIG. 7B, the terminal 11-a is connected to the collector of the PNP transistor 15, the terminal 11-b is connected to the resistor 18, and the other end of the resistor 18 is connected to the emitter of the transistor 15. A Zener diode 20 is connected between the collector and the base of the transistor 15 in the orientation shown in the drawing, and a resistor 19 is connected between the base and the emitter. In the circuit of FIG. 7 (2), a current flows from the terminal 11-b to the terminal 11-a so that the collector-emitter voltage of the transistor 15 becomes the voltage of the Zener diode 20, and the voltage between these terminals is , Zener diode 2
The voltage value is equal to the sum of the voltage of 0 and the voltage drop of the resistor 18. When applied to the circuit shown in FIG. 1, the voltage clamp capacitor 5 operates so as to have a voltage value substantially equal to the sum of the voltage of the Zener diode 20 and the voltage drop of the resistor 18. Therefore, the peak voltage of the switching semiconductor element 3 is clamped at a voltage value equal to the sum of the voltage of the Zener diode 20, the voltage drop of the resistor 18, and the voltage Ei of the DC power supply 1.

When the circuit of FIG. 7B is used, the voltage of the voltage clamp capacitor 5 is stable even when the load suddenly changes or the input voltage fluctuates. Enables high-speed control.

FIG. 7C shows the zener diode 20 of FIG. 7B changed to a control circuit 16, and the other configuration and operation are the same as those of FIG. 7B.
The control circuit 16 has a structure capable of controlling the voltage of the transistor 15. When applied to the circuit of FIG. 1, it becomes possible to control the peak value of the voltage of the switching semiconductor element 3 in accordance with the current of the load circuit 10 or the change of the voltage Ei of the DC power supply 1, for example, and to control the input voltage and the load current. Even if the range is wide, the peak voltage of the switching semiconductor element 3 can be minimized.

Although any of the above-described embodiments has the great effect as described above, the discharge circuit 11 is not shown in FIG.
In the case of using only the resistor as shown in (1) and using the conventional control circuit having the circuit configuration as shown in FIG. 9,
There arises a problem that the conversion frequency of the switching power supply fluctuates greatly. This problem will be described with reference to FIG. 10 which shows operation waveforms corresponding to respective times in FIG.

At time t7 in FIG. 10, the voltage detection circuit 32 detects that the voltage across the switching semiconductor element 3 has become zero or the minimum voltage, and outputs a detection signal to the drive circuit 35 to drive the drive circuit. Reference numeral 35 turns on the switching semiconductor element 3. Since the error amplifier 37 controls the output voltage of the switching power supply to a constant voltage, it detects the output voltage and outputs an error signal obtained by amplifying the error voltage with respect to the set voltage. The current detection circuit 33 detects the current of the switching semiconductor element 3 or the current of the diode 6, and the error amplifier 37 compares the current detection signal with the error signal. When the current detection signal becomes larger than the error signal, the comparison is made. The device 36 outputs a signal to the drive circuit 35, and the drive circuit 35 turns off the switching semiconductor element 3.

Since the voltage of the switching semiconductor element 3 is equal to the magnitude of the input voltage at startup, the voltage detection circuit 32 does not operate. Therefore, the drive signal of the switching semiconductor element 3 at the time of startup is the startup circuit 3
4 outputs. The starting circuit 34 detects the input voltage,
When the input voltage exceeds the selected voltage value, a signal is output to the drive circuit 35, and the drive circuit 35 turns on the switching semiconductor element 3.

In such a driving method, the conversion frequency is determined by the main circuit constants and the input / output conditions. Therefore, when the input / output conditions change, the conversion frequency greatly changes, and the input / output filter is set at the lowest frequency. There is a problem such as an increase in cost and an increase in the size of the switching power supply because it is necessary or the component is determined at the highest frequency.

Therefore, even if the input / output conditions change next, the change width of the conversion frequency can be made very small.
One embodiment of No. 1 is shown in FIG. The terminals 11-a and 11-b in FIG. 7 (4) correspond to the terminals 11-a and 11-b of the discharge circuit 11 in FIG. 1, respectively. By adopting the discharge circuit 11 in the switching power supply shown in FIG. 8 and controlling the amount of discharge current, the voltage of the voltage clamp capacitor 5 and the voltage applied to the winding of the transformer 2 are changed,
The feature of this embodiment is that the off time of the switching semiconductor element 3 can be changed so that the conversion frequency becomes constant.

First, in this control method, the voltage applied to the winding at the time of resetting the transformer 2 can be controlled by controlling the voltage of the voltage clamp capacitor 5, and accordingly, when the switching semiconductor element 3 is turned off. It is based on the knowledge that the conversion frequency changes as the slope of the reset current changes and the time until completion of resetting changes.

The control circuit 13 shown in FIG. 8 is different from the conventional control circuit shown in FIG. 9 in that the f / v conversion circuit 21, the error amplifier 23,
Reference voltage source 24, light emitting diode 25-a of photocoupler
And a resistor 26 is added. f / v conversion circuit 2
1 receives a drive signal supplied from the drive circuit 35 to the switching semiconductor element 3 and outputs a DC voltage having a magnitude corresponding to the conversion frequency of the switching power supply. Error amplifier 2
3 obtains an error voltage between this DC voltage and the reference voltage of the reference voltage source 24 for setting a desired conversion frequency, amplifies this error voltage to a desired level to generate an error amplification signal, and outputs a photocoupler light emitting diode 25-. a and the light receiving transistor 25
It is given to the base of the transistor 31 of the discharge circuit 11 through -b.

Therefore, the transistor 31 supplies the current corresponding to the error amplified signal to the voltage clamp capacitor 5
Is discharged to the input power source 1 side, and the discharge current is a current controlled to have a set conversion frequency. The discharge current controls the voltage of the voltage clamp capacitor 5, the voltage applied to the winding of the transformer 2, and the off time of the switching semiconductor element 3 to reach a target conversion frequency.

In this embodiment, the current mode control of the conventional control method is used, and the conversion frequency is fixed or the change width thereof is made small by adding the discharge circuit having the above-described configuration. The same effect can be obtained by using the voltage mode control.
Moreover, since the loss of the discharge circuit can be reduced, this embodiment is particularly suitable for the discharge circuit of the present invention.

As described above, when the switching semiconductor element 3 is turned on, the energy stored in the excitation inductance, the leakage inductance of the transformer 2 and the wiring inductance is consumed, and most of the energy is consumed when the switching semiconductor element 3 is turned off. Without being stored, the energy can be stored in the voltage clamp capacitor 5, and most of the energy can be recirculated to the exciting inductance, the leakage inductance, and the wiring inductance of the transformer 2 to return the energy to the DC power supply 1. It can be turned on at zero voltage by selectively designing. Further, since the switching semiconductor element 3 is not turned on until the reverse conduction of the charge storage diode 4 is completed, noise and loss due to the reverse conduction of the charge storage diode 4 can be reduced.

In the above embodiments, the power supply that rectifies the voltage on the secondary side of the transformer to obtain a DC output is described, but the AC voltage is applied to the load without rectifying the voltage on the secondary side of the transformer. The present invention can be similarly applied to a power supply that supplies In this case, a diode for rectifying the detection voltage signal of the output voltage is required as in a general voltage detection circuit.

[0053]

As described above, in the present invention, the charge storage type diode having a relatively long carrier lifetime and the voltage clamp capacitor are combined with the inductance means, and the voltage clamp is performed in accordance with the change of the input voltage or the load current. By providing a discharge circuit that adjusts the discharge amount of the capacitor and controls its voltage to be almost constant, the energy stored in the excitation inductance, leakage inductance of the transformer, and wiring inductance is switched when the switching semiconductor element is turned on. When the semiconductor element is turned off, the energy is stored in the voltage clamp capacitor, and most of the energy can be recirculated to the exciting inductance of the transformer, the leakage inductance, and the wiring inductance. As The transformer can be reliably reset without the need for a reset switch.

Further, by selectively designing the constant, the constant voltage can be turned on at zero voltage, and the switching semiconductor element is not turned on until the reverse conduction of the charge storage type diode is completed. Generation of noise and power loss due to reverse conduction can be reduced. Furthermore, these effects can be obtained without complicating the circuit configuration.

Furthermore, the impedance of the discharge circuit is controlled according to the conversion frequency of the switching power supply,
By adjusting the discharge amount of the voltage clamp capacitor, the conversion frequency of the switching power supply can be set to a target value or its fluctuation range can be made sufficiently small.

[Brief description of drawings]

FIG. 1 is a diagram showing an embodiment of a switching power supply according to the present invention.

FIG. 2 is a diagram showing a waveform of each part for explaining the embodiment.

FIG. 3 is a diagram showing a power recovery rate depending on a carrier lifetime of a diode.

FIG. 4 is a diagram showing a second embodiment of the switching power supply according to the present invention.

FIG. 5 is a diagram showing a third embodiment of the switching power supply according to the present invention.

FIG. 6 is a diagram showing another separate embodiment of the switching power supply according to the present invention.

FIG. 7 is a diagram showing an example of each of the discharge circuits used in the present invention.

FIG. 8 is a diagram showing another embodiment of the switching power supply according to the present invention.

FIG. 9 is a diagram showing an example of a conventional control circuit for a switching power supply.

FIG. 10 is a diagram showing waveforms of respective parts for explaining a conventional control method.

FIG. 11 is a diagram showing an example of a conventional switching power supply.

FIG. 12 is a diagram showing waveforms of various parts for explaining a conventional switching power supply.

[Explanation of symbols]

1 ...- DC power supply 2 ...- Transformer 3 having primary winding 2A and secondary winding 2B ... Switching semiconductor element 4, 4 '... Charge storage type Diodes 5 and 5 '... Voltage clamp capacitor 6 ... Rectifying diode 7 ... Free wheeling diode 8 ... Smoothing inductor 9 ... Smoothing capacitor 10 ... Load resistors 11, 11 '... Discharge circuits 11-a, 11-b, 11'-a, 11'-b ...
Discharge circuit terminal 11 "... Discharge resistor 12 ... Second capacitor 13 ... Control circuit 16 ...
-Control circuit 14 ...- Second diode 17-19
.... resistance 15 ... transistor 20 ...
・ Zener diode 21 ・ ・ ・ ・ Pulse generator 22 ・ ・ ・
・ Smoothing circuit 23 ・ ・ ・ ・ Differential amplifier 24 ・ ・ ・
-Reference voltage 25-a --- Photocoupler light-emitting diode 25-b --- Photocoupler light-receiving diode

Claims (5)

[Claims] 1. A switching semiconductor element that selectively opens and closes a current path from a DC power supply to a load, an inductance means connected in series to the switching semiconductor element, and a parallel connection to the switching semiconductor element, and mutually. A charge storage diode connected in series, a voltage clamp capacitor that acts to clamp the voltage of the switching semiconductor element, and a connection point between the charge storage diode and the voltage clamp capacitor and the DC power supply. A discharge circuit, and a control circuit for controlling switching of the switching semiconductor element, wherein the charge storage diode is first conducted in the forward direction and stored in the inductance means during a period in which the switching semiconductor element is off. Store the stored energy in the voltage clamp capacitor Next, when the current flowing in the forward direction becomes zero, the current is conducted in the reverse direction and the energy stored in the voltage clamp capacitor is returned to the DC power supply through the inductance means, and the discharge circuit is the charge storage diode. Of the energy charged in the voltage clamp capacitor by the forward conduction of <1>, which is not released by the reverse conduction of the charge storage diode. 2. A switching semiconductor element for selectively opening and closing a current path from a DC power source to a load, an inductance means connected in series with the switching semiconductor element, and an inductance means connected in parallel with the inductance means and in series with each other. A connected charge storage diode and a voltage clamp capacitor, a discharge circuit connected between the DC power supply and a connection point between the charge storage diode and the voltage clamp means, and switching of the switching semiconductor element. The charge storage diode is first conducted in the forward direction to store the energy stored in the inductance means in the voltage clamp capacitor during the period in which the switching semiconductor element is off, and When the current flowing in the forward direction becomes zero, it conducts in the reverse direction and the voltage The energy stored in the clamp capacitor is returned to the DC power source through the inductance means, and the discharge circuit is the charge storage diode among the energy charged in the voltage clamp capacitor by the forward conduction of the charge storage diode. A switching power supply, which discharges energy that has not been released due to reverse conduction of the switching power supply. 3. The discharge circuit according to claim 1, wherein a discharge circuit capable of adjusting the energy amount of the voltage clamp capacitor is provided in parallel with the voltage clamp capacitor so that the voltage of the voltage clamp capacitor does not exceed a set voltage. A switching power supply characterized by being equipped with. 4. The switching power supply according to claim 1, wherein the discharge circuit has a variable impedance corresponding to a frequency of a signal synchronized with a driving signal for driving. 5. The switching power supply according to claim 1, wherein the switching semiconductor element is turned on after the reverse conduction of the charge storage diode is completed.