JP2001177734A - High voltage control circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a high voltage control circuit that can conduct high voltage control without having effect on horizontal deflection amplitude and can effectively conduct image distortion correction such as horizontal pincushion distortion correction almost without any power loss in the high voltage control with a very simple configuration without needing any large-sized component. SOLUTION: An inductor 10 is equivalently connected in parallel with a ternary winding 17c in terms of AC for a conduction period t-on of a diode 11 at a top of a pulse Vp, the horizontal resonance frequency is decreased, the peak value of a collector pulse Vc is increased and a value of a DC high voltage HV is also increased. Changing the phase of a square wave pulse Vsw allows an electronic switch 17 to change on ON/OFF phase so as to change a conductive period of the diode 11 thereby freely adjusting the value of the DC high voltage HV.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a high voltage control circuit for controlling a high DC voltage supplied to a cathode of a picture tube, and more particularly to a high voltage control circuit capable of efficiently controlling only a DC high voltage with little effect on a horizontal deflection operation. It relates to a control circuit.

[0002]

2. Description of the Related Art FIG. 8 is a circuit diagram showing a conventional horizontal deflection circuit, FIG. 9 is a waveform diagram for explaining the operation of FIG.
FIG. 9 is a characteristic diagram for explaining the operation of FIG. 8 and will be described together. The conventional example shown in FIG. 8 has been previously filed by the present applicant. In FIG. 8, an excitation pulse Vdr from a horizontal excitation circuit in a preceding stage (not shown) is supplied to a primary winding of an excitation transformer 1, and a base current Ib of a horizontal output transistor (switching element) 2 is supplied from the secondary winding. are doing. In this case, the horizontal output transistor 2 is connected to the damper diode 3 according to a well-known principle.
The switching operation is performed in cooperation with.

During the off period of the switching operation, a half-sine-wave collector pulse Vc is generated at the collector terminal of the horizontal output transistor 2. The retrace resonance capacitor 4 has a pulse width (return time) of the collector pulse Vc.
It is for determining. Horizontal output transistor 2
The horizontal deflection coil 5 and S
A series circuit with the character correction capacitor 6 is connected, and a sawtooth-shaped deflection current Iy having a horizontal deflection cycle flows through the series circuit to deflect the electron beam of the picture tube (not shown) in the horizontal direction.

The collector pulse Vc is supplied to one end of a primary winding 7a of a flyback transformer 7, and a boosted high-voltage pulse Vhv is generated in a secondary winding 7b. The other end of the primary winding 7a is connected to a DC power supply voltage (first DC voltage) Eb, from which power is supplied to the circuit. The capacitor 9 is for smoothing the ripple current of the circuit. The high-voltage pulse Vhv is supplied to the high-voltage rectifier diode 8, rectified to obtain a DC high-voltage HV, and supplied to the anode of a picture tube (not shown). Also, flyback transformer 7
The DC control voltage Ec0 is supplied to one end of the tertiary winding 7c. The other end p of the tertiary winding 7c and the DC power supply voltage E
b, a series circuit of an inductor 10 and a diode (first diode) 11 is connected, and a small pulse Vp obtained by transforming the collector pulse Vc is applied to a point p of the tertiary winding 7c. It has occurred.

The control voltage Ec0 is different from the DC power supply voltage Ec.
c through the voltage control circuit 12. The voltage control circuit 12 is a chopper-type control circuit using an electronic switch 13, and includes an electronic switch 13 that repeats on / off operation by a control pulse Vch, a flywheel operating diode 14, a smoothing choke coil 15, and a smoothing choke coil 15. Of the capacitor 16. In the voltage control circuit 12, the DC power supply voltage Ec
At 3, the pulse is intermittently turned on and off by the control pulse Vch, and becomes a square wave pulse reciprocating between zero and the voltage Ec. This square wave pulse is converted into a direct current by the choke coil 15 and the capacitor 16, and becomes a control voltage Ec0. Therefore, when the duty cycle of the control pulse Vch is changed, the value of the control voltage Ec0 changes accordingly.

Here, the change in the control voltage Ec0 is caused by the DC high voltage H
How it affects V will be described. First, it is assumed that the peak value of the small pulse Vp is set to be substantially the same as or slightly smaller than the value of the DC power supply voltage Eb, as shown in FIG. The DC control voltage Ec0 is superimposed on the pulse Vp. If the control voltage Ec0 is zero, the peak value of the pulse Vp does not exceed the DC power supply voltage Eb, so that the diode 11 is off and no current flows.

Next, as shown in FIG. 9B, the control voltage Ec0
Has a positive voltage value, the voltage level of the entire pulse Vp rises, so that the top of the pulse Vp temporarily exceeds the DC power supply voltage Eb. Then, during that time, the diode 11 conducts, the current Id flows, and the DC power supply voltage Eb
Charge the capacitor 9 connected to. Inductor 10
The waveform Vp1 at the connection point between the diode 11 and the diode 11 has a shape in which the top is sliced by the voltage Eb during the conduction period ton1 of the diode 11. This is equivalent to the fact that the inductor 10 is connected in parallel to the tertiary winding 7c of the flyback transformer 7 during the conduction period ton1. This is equivalent to the addition of a new inductor in parallel. At this time, the resonance period naturally becomes shorter by that amount.

Then, the pulse width of the pulse Vp in FIG. 9 (B), that is, the retrace time tr1 in FIG. 9 (A), which is the original retrace time in FIG.
shorter than r. As a result, the peak value of the collector pulse Vc increases, and the value of the DC high voltage HV in the case of FIG. 9B becomes higher than that in the case of FIG. 9A in proportion thereto. Further, as shown in FIG. 9 (C), when the control voltage Ec0 further rises, the conduction period of the diode 11 becomes longer as ton2 and the retrace time becomes shorter as tr2. As a result, the DC high voltage HV further rises on the same principle.

As described above, the circuit of the conventional example shown in FIG. 8 can freely adjust the value of the DC high voltage HV by moving the value of the control voltage Ec0 by the operation of the voltage control circuit 12. At this time, the current Id flowing through the diode 11 charges the capacitor 9 connected to the DC power supply voltage Eb, and the energy is stored in the primary winding 7a of the flyback transformer 7.
Is used for the operation of the circuit. Therefore, DC high voltage H
If the control voltage Ec0 is increased to increase V,
As shown in FIG. 10, the circuit current Ib flowing from the DC power supply voltage Eb tends to decrease. However, as the control voltage Ec0 increases, the current Id flowing from the diode 11 increases. As a result, the total power consumption of the entire circuit hardly changes, and no extra power is required for high-voltage control.

[0010]

The conventional example shown in FIG. 8 has an advantage that the high voltage value can be controlled with almost no loss, but for that purpose, it is necessary to change the control voltage Ec0. In the case of FIG. 8, this is performed by the chopper type voltage control circuit 12, but the circuit is accordingly complicated. In particular, the choke coil 15 often needs to be large in both volume and weight. Also, the smoothing capacitor 16 needs to flow a large ripple current, so that the size thereof is increased. These were the drawbacks of this circuit.

Although not shown, when a series regulator using a class-A operation semiconductor element is used instead of the chopper method of the voltage control circuit 12 in FIG.
No choke coil is needed. However, in this case, a power loss always occurs in the semiconductor element, which has a disadvantage in that the features of the circuit of FIG. 8 which originally operates with low loss are diminished. The present invention has been made to solve the above-described problems, and can perform high-pressure control without affecting the horizontal deflection amplitude, has almost no power loss during high-pressure control, and has image distortion such as left and right pincushion distortion correction. It is an object of the present invention to provide a high-voltage control circuit that can perform correction effectively, does not require large components, and can be realized with a very simple configuration.

[0012]

In order to achieve the above object, a switching element for performing an on / off operation in a horizontal deflection cycle, and a series circuit of a horizontal deflection coil and an S-shaped correction capacitor connected in parallel to the switching element. A flyback transformer having one end of a primary winding connected to one end of the switching element and the other end connected to a first DC voltage. A high-voltage rectifier diode connected to one end of a secondary winding of the flyback transformer and outputting a high DC voltage, and a tertiary rectifier diode of the flyback transformer having one end connected to a second DC voltage via an electronic switch. A first winding and an inductor connected between the other end of the tertiary winding of the flyback transformer and the first DC voltage;
And a control signal generating circuit for generating a square wave pulse for turning on and off the electronic switch, and by changing the phase of the square wave pulse, the conduction of the first diode is controlled. Another object of the present invention is to provide a high voltage control circuit characterized by controlling a value of the DC high voltage by changing a period.

[0013]

FIG. 1 is a circuit diagram showing a first embodiment of the present invention, and FIGS. 2 and 3 are waveform diagrams for explaining the operation of FIG. 1. In FIG. 1,
The same reference numerals are given to the same parts as those in FIG. In FIG. 1, the main difference from FIG. 8 of the conventional example is that an electronic switch 17 is used instead of the voltage control circuit 12 in FIG. As the electronic switch 17,
For example, a p-type FET or the like can be used. The electronic switch 17 performs an on / off operation in accordance with the square wave pulse Vsw of the horizontal deflection period supplied from the control signal generation circuit 19.

The diode 18 is a stabilizing diode and corresponds to the flywheel diode 14 in the conventional example of FIG. 8, but the operating conditions are slightly different as described later, and may be omitted in some cases. It is a thing. The control signal generation circuit 19 is formed of, for example, a microcomputer or the like, and generates a square wave pulse Vsw having a horizontal deflection period having an appropriate phase and a duty cycle according to the high voltage condition of the circuit.

Next, in FIG. 2, the square wave pulse Vsw
2 (A) to FIG. 2 (E), the pulse width is generated at the connection point q between the inductor 10 and the diode (first diode) 11 when the phase is sequentially shifted as shown in FIGS. The state of the pulse Vq, the current Ids of the diode 18 and the DC high voltage HV are shown. The electronic switch 17 is
It is assumed that the switch is turned on during the low level period of the square wave pulse Vsw. First, in FIG. 2A, the low level period of the square wave pulse Vsw is the pulse width of the pulse Vq, that is, the retrace time t.
The electronic switch 17 is in an off state during a pulse Vp (pulse Vq) period tr, and the tertiary winding 7c and the inductor 10 are switched from the DC power supply voltage (second DC voltage) Ec. DC power supply voltage through diode 11
Since the path leading to the (first DC voltage) Eb is blocked, the current Id of the diode 11 does not flow. Accordingly, there is no effect of shortening the retrace time according to the above-described principle, the retrace time remains at the original tr, and the pulse Vq has a pulse Vq shape.
It is almost the same as p.

Next, as shown in FIG. 2B, consider a case where the square wave pulse Vsw changes from a low level to a high level near the tip of the pulse Vp, ie, near the center of the retrace time. Then, there is a period ton1 in which the electronic switch 17 is on and the tip of the pulse Vp exceeds the DC power supply voltage Eb. At this time, the diode 11 is turned on, and the top of the pulse Vq is sliced by the conduction period ton1. Then, during this conduction period ton1,
As described above, the inductor 10 is connected in parallel to the tertiary winding 7c of the flyback transformer 7 and equivalently to the resonance capacitor 4c.
2A, the retrace time tr1 is shorter than in the case of FIG. As a result, the DC high voltage HV slightly increases as compared with the case of FIG.

Next, as shown in FIG. 2C, it is assumed that the low-level portion of the square-wave pulse Vsw further moves and extends over the entire pulse width of the pulse Vp. Then, the conduction period ton2 in which the tip of the pulse Vp exceeds the power supply voltage Eb is maximized.
At this time, the retrace time tr2 is further shortened, and the DC high voltage HV becomes the maximum value of this circuit. Further, as shown in FIG. 2D, when the position of the square wave pulse Vsw moves to the rear of the pulse Vp and the transition point from the high level to the low level is near the top of the Vp pulse, this transition point Only after that the current I
Since d does not flow, the time during which the top of the Vq pulse is sliced, that is, the conduction period ton3 during which the inductor 10 is effectively working, becomes shorter again. And return time tr3
Is longer than in the case of FIG. 2 (C), and the DC high voltage HV also decreases.

Finally, as shown in FIG. 2E, when the low-level portion of the square wave pulse Vsw is located after the retrace time, there is no period during which the diode 11 is turned on, and the inductor 10 is effective. Does not work. Accordingly, the retrace time returns to the original maximum value tr, and the value of the DC high voltage HV also becomes minimum. As described above, it is understood that the value of the DC high voltage HV can be continuously changed depending on the phase of the square wave pulse Vsw. For this control, the range shown in FIGS. 2A to 2C may be used, or the range shown in FIGS. 2E to 2C in the latter half may be used. In any case, the DC high voltage HV can be changed from the lowest value to the highest value.

The diode 18 for stabilization includes:
This is to prevent a transient voltage from being generated at the point r due to the inductance value formed by the tertiary winding 7c and the inductor 10 when the current Isw of the electronic switch 17 is rapidly turned off. Due to the function of the diode 18, even if the current Isw suddenly disappears, the current Ids automatically flows to take over, so that there is no discontinuity in the current flowing to the tertiary winding 7c and the inductor 10, and the transient voltage Does not occur. However, in the case of FIG. 2A and FIG. 2E, a circuit from the DC power supply voltage Ec to the DC power supply voltage Eb via the electronic switch 17, the tertiary winding 7c, the inductor 10, and the diode 11 is as follows. Either the electronic switch 17 or the diode 11 is always off, and no current flows. Therefore, no transient voltage is generated at the point r, so that no current flows to the diode 18.

2 (C) and 2 (D), the current is cut off by the diode 11 and no current has already flowed into the circuit when the electronic switch 17 is turned off. Therefore, also in this case, no transient voltage is generated, and the current of the diode 18 is zero. After all, electronic switch 1
7 interrupts the current only in the intermediate stage of the control shown in FIG. 2B. At this time, the current Ids flows to the diode 18, but this current is generated when the energy stored in the inductor 10 is emptied. Disappear. For this reason, this current Ids is not so large in both the circulation time and the peak value.

On the other hand, the conventional flywheel diode 14 shown in FIG. 8 continues to flow throughout the off period of the switch 13, and its peak value reaches the same magnitude as the peak value of the switch 13. Compared to this, the value of the current flowing to the stabilizing diode 18 in FIG.
That is, a small device is required. In some cases, the diode 18 can be omitted. In the case where the diode 18 is not provided and the phase relationship is as shown in FIG. 2B, the waveform of the voltage Vr at the output terminal r of the electronic switch 17 is as shown in FIG. Vibrate. However, this does not cause a serious problem unless the negative peak value of the vibration waveform does not exceed the withstand voltage of the electronic switch 17 and this vibration component is induced to another circuit to exert an adverse effect.

In particular, if the control of the DC high voltage HV is performed in the range of FIGS. 2 (E) to 2 (C), the current Ids of the diode 18 does not originally flow, and can be omitted without any problem. However, since the peak value of the pulse Vp is set slightly closer to the DC power supply voltage Eb, the pulse
There may be a case where the pulse peak of p exceeds the DC power supply voltage Eb. At this time, a current Ids slightly flows through the diode 18 even when the electronic switch 17 is off, as shown by the broken line in FIG. However, even in such a case, the current value of the current Ids is still smaller, and the effect is small even if the diode 18 is omitted. Thus, in FIG. 1 of the present invention, as compared with FIG. 8 of the conventional example, not only the choke coil 15 and the capacitor 16 of FIG. Can be greatly reduced in size or can be omitted. This greatly contributes to miniaturization and power saving of equipment.

FIG. 4 is a circuit diagram showing a second embodiment of the present invention. In FIG. 4, a DC high voltage HV is a voltage dividing resistor 2.
The voltage is divided by 0 and 21 to obtain a small voltage Ehv proportional to the DC high voltage HV. This small voltage Ehv is supplied to the comparator 22 together with the DC reference voltage Es. The comparator 22 outputs the comparison result voltage Erf and supplies it to the control signal generation circuit 19. The control signal generation circuit 19 is basically the same as the one shown in FIG. 1, except that the time position of the rising or falling point of the square wave pulse Vsw to be output is moved forward or backward according to the value of the voltage Erf. It is moving. The square wave pulse Vsw, which is the output of the control signal generation circuit 19, is supplied to the electronic switch 17 and controls on / off of the electronic switch 17, as in FIG.

The electronic switch 17 comprises a p-type FET 24, a diode (second diode) 25, resistors 26 and 27 for gate bias, and a capacitor 28 for blocking direct current. The diode 25 is built in the FET 24 and does not need to be provided separately. In this configuration, for example, the high voltage load current I flowing to the picture tube anode
When hv increases and the DC high voltage HV attempts to decrease, the falling point of the square wave pulse Vsw is moved to the front as shown in FIGS. The effect of shortening the retrace time is increased. As a result,
The pulse Vc rises and acts to increase the DC high voltage HV. As a result, the feedback loop operates so that the voltage Ehv obtained by dividing the DC high voltage HV always matches the DC reference voltage Es, and the DC high voltage HV is stabilized.

In a display device using a normal picture tube, it is common to correct image distortion by modulating a horizontal deflection current Iy with a waveform of a vertical deflection cycle. For example,
An appropriate amplitude modulation circuit may be added to a series circuit of the horizontal deflection coil 5 and the S-shaped correction capacitor 6. In particular,
A known saturable reactor, diode modulator, or the like may be used to modulate the envelope of the sawtooth-shaped deflection current Iy with a parabolic wave Vpb having a vertical deflection period. Then
The so-called right and left pincushion distortion of the image can be corrected.
At this time, not only the deflection current Iy but also the DC high voltage HV is modulated and the ripple voltage is superimposed, so that the right and left pinsion distortion correction effect may be reduced or the correction may not be performed accurately. This phenomenon can be reduced by adding a capacitor 23 as shown by a broken line in FIG.
Capacitors that withstand this type of high voltage are large and very expensive.

FIG. 5 is a circuit diagram showing a third embodiment of the present invention. 5, the main difference from FIG. 4 is that an amplitude modulation circuit 29 is added in a series circuit of the horizontal deflection coil 5 and the S-shaped correction capacitor 6, and the amplitude of the deflection current Iy is modulated by the parabolic waveform Vpb. And the waveform V which cancels the fluctuation of the DC high voltage HV caused by the amplitude modulation circuit 29.
c1 is input to the control signal generation circuit 19 simultaneously with the voltage Erf. Then, the time position of the leading edge or the trailing edge of the pulse Vsw is modulated according to the waveform Vc1, and the DC high voltage HV changes at the same time. This offsets the movement of the DC high voltage HV by the amplitude modulation circuit 29, and the ripple waveform of the DC high voltage HV decreases.

FIG. 6 is a circuit diagram showing a fourth embodiment of the present invention. 6, the main difference from FIGS. 1, 4, and 5 is that the inductor 10 and the diode (first diode)
This is the point that a diode (third diode) 30 for damping is newly provided between the connection point q with the reference numeral 11 and the ground.
The smoothing capacitor 31 connected to the DC power supply voltage Ec was present in FIGS. 1, 4 and 5, but was not particularly shown as being included in the DC power supply voltage Ec. is there. In FIG. 6, if the diode 3
When 0 does not exist, the pulse Vq generated here may vibrate violently at the trailing edge of the pulse as shown in FIG. This occurs when the pulse Vp passes the top, its value drops below the DC power supply voltage Eb, and the diode 11 switches from on to off. This cannot be avoided to some extent as long as there is a distributed capacitance of the inductor 10 and a stray capacitance of the circuit.

This phenomenon does not adversely affect the essential operation of the circuit unless the reverse withstand voltage of the diode 11 is exceeded or the vibration itself does not affect other circuits due to the negative peak of the vibration. . However, I would like to suppress this vibration if there are simple measures. Usually, although not shown, damping by a resistor and a capacitor can be considered to reduce such a vibration phenomenon. However, since power is consumed by the resistor for a small effect, it is not very advantageous.

In FIG. 6, this problem is solved by connecting a diode 30 between point q and ground. That is, in this case, when the pulse Vq swings negatively, the diode 30 turns on, and the point q does not become more negative than almost zero. Therefore, as shown in FIG. 7 (B), the amount of vibration in the waveform of the pulse Vq is effectively suppressed. When the diode 30 is turned on, the current Id2 charges the smoothing capacitor 31 connected to the DC power supply voltage Ec via the inductor 10, the tertiary winding 7c, and the diode 25. Since the charged energy becomes a part of the current flowing when the diode 11 is turned on, the current originally flowing from the DC power supply voltage Ec can be saved correspondingly.

As described above, the diode 3 shown in FIG.
0 not only has an effective damping action, but also has the effect of improving the power efficiency of the entire circuit. Further, since the peak value and the distribution period of the current flowing through the diode 30 are extremely small, a smaller one can be used as compared with the diode 11. In FIG. 6, FET2 is used as the electronic switch element.
4 is used, but it is of course possible to use a bipolar transistor. In that case, the diode 30
In order to obtain the energy recovery effect due to the above, there is no built-in diode, so the diode 25
Must be connected between collectors.

[0031]

The high-voltage control circuit of the present invention can perform high-voltage control without affecting the horizontal deflection amplitude, has little power loss during high-voltage control, and is effective in correcting image distortion such as left and right pincushion distortion. It has an extremely excellent effect that it can be realized with a very simple configuration without requiring large parts and with a very simple configuration.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a first embodiment of the present invention.

FIG. 2 is a waveform chart for explaining the operation of FIG.

FIG. 3 is a waveform chart for explaining the operation of FIG. 1;

FIG. 4 is a circuit diagram showing a second embodiment of the present invention.

FIG. 5 is a circuit diagram showing a third embodiment of the present invention.

FIG. 6 is a circuit diagram showing a fourth embodiment of the present invention.

FIG. 7 is a waveform chart for explaining the operation of FIG. 6;

FIG. 8 is a circuit diagram showing a conventional horizontal deflection circuit.

FIG. 9 is a waveform chart for explaining the operation of FIG.

FIG. 10 is a characteristic diagram for explaining the operation of FIG. 8;

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Horizontal excitation transformer 2 Horizontal output transistor (switching element) 3 Damper diode 4 Retrace resonance capacitor 5 Horizontal deflection coil 6 S-shaped correction capacitor 7 Flyback transformer 8 High voltage rectifier diode 9,16,23,28,31 Capacitor 10,15 Inductor 11 Diode (first diode) 12 Voltage control circuit 13, 17 Electronic switch 14, 18 Diode 19 Control signal generation circuit 20, 21, 26, 27 Resistance 22 Comparator 24 FET 25 Diode (second diode) 29 Amplitude Modulation circuit 30 Diode (third diode) Vc Collector pulse Vp, Vp1 pulse Vpb Distortion correction waveform Vsw Square wave pulse Eb DC power supply voltage (first DC voltage) Ec DC power supply voltage (second DC voltage) Ec0 Control voltage Ehv, Erf voltage Es DC reference voltage Iy deflection voltage Ib DC power supply current Id, Ids current HV DC high tr, tr1, tr2, tr3 retrace time ton, ton1, ton2 conduction period of diode

Claims (4)

[Claims] 1. A switching element for performing an on / off operation in a horizontal deflection cycle, a series circuit of a horizontal deflection coil and an S-shaped correction capacitor connected in parallel to the switching element, and a flyback connected in parallel to the switching element. A resonance capacitor, a flyback transformer having one end of a primary winding connected to one end of the switching element, and the other end connected to a first DC voltage; and one end of a secondary winding of the flyback transformer. A high-voltage rectifier diode connected to output a DC high voltage, a third winding of the flyback transformer having one end connected to a second DC voltage via an electronic switch, and a third winding of the flyback transformer The other end and the first
A series circuit of an inductor and a first diode connected between the square wave pulse and a control signal generation circuit that generates a square wave pulse for turning on and off the electronic switch. A high-voltage control circuit which changes a phase to change a conduction period of the first diode, thereby controlling a value of the DC high voltage. 2. A comparator for comparing a voltage proportional to the DC high voltage with a DC reference voltage, and a control signal generating circuit for generating the square wave pulse for changing the phase according to an output of the comparator. The high-voltage control circuit according to claim 1, wherein 3. An amplitude modulation circuit for modulating a deflection current flowing through the horizontal deflection coil, and an output of the comparator and a modulation waveform for canceling the DC high voltage modulation effect generated by the amplitude modulation circuit. The high-voltage control circuit according to claim 2, further comprising the control signal generation circuit. 4. A second diode connected in parallel with the electronic switch and flowing a current in a direction opposite to a current flowing through the electronic switch, from a ground point to a connection point between the inductor and the first diode. And a third diode connected in such a direction as to be forward.
4. The high-voltage control circuit according to 3.