JP2001099914A - Method of processing signal, signal processor, and sonar device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To remarkably reduce the number of amplifiers and AD converters in a submarine prospecting sonar using a reception transducer having a large number (160 channels) of elements. SOLUTION: The elements of the 160 channels are multiplexed into 10 systems by 10 pieces of multiplexers. The ten AD converters AD-convert signals of the 16 channels respectively by time sharing to make concurrent switching timing of the multiplexers and to make 10 pieces of data respectively the sampling data of the same timing. The 160 pieces of sampling data sampled steppedly by this manner are phase-shifted diagonally. The signal of the same channel is sampled twice times with 90 deg. of phase difference to complexify the sampling data without increasing a processing circuit.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a signal processing method and a signal processing device used for a communication device, a sonar device, an ultrasonic diagnostic device or a flaw detection device, and to the application of the signal processing method and the signal processing device. The present invention relates to a scanning sonar device and a cross-fan beam type sonar device for detecting a seafloor bottom, in which a receiving beam is sequentially formed in different directions to search in a wide range direction. Hereinafter, the present invention will be described using a cross fan beam type sonar device as an example.

[0002]

2. Description of the Related Art A cross-fan beam type sonar apparatus used in the prior art has been used to convert echo signals received by, for example, 160 ultrasonic transducers into analog signals using 160 mixers. The desired receiving beam was formed by combining after the quantitative phase shift.

[0003]

As described above, the input stage of the sonar device is composed of a large number of ultrasonic transducer elements such as 160 channels, so that the device must be large and the error of the analog circuit must be increased. There is a problem that the distance resolution cannot be increased to a certain degree or more due to deterioration over time or the like.

[0004] The present invention simplifies the circuit configuration by multiplexing and time division, and provides a signal processing method and a signal processing apparatus applied to a communication device, a sonar device, an ultrasonic diagnostic device, a flaw detection device, and the like. It is an object of the present invention to provide a sonar device using the same, particularly a scanning sonar device which forms a receiving beam sequentially in different directions to search in a wide range direction.

[0005]

According to the present invention, a signal having a predetermined frequency f (= 1 / T) is received, a predetermined sampling time, and (a + 1 / T) from the predetermined sampling time.
4) Time after T (a = 0, 0.5, 1, 1.5,...:
The same is true in the claims), and the data sampled at these times are output as real part data and imaginary part data of complex sampling data.

The present invention is characterized in that the time of (a + ／) T is shorter than の of the period of the predetermined sampling time.

The present invention is characterized in that the two samplings are repeated every (m + 1/2) T (m = 1, 2,...).

According to the present invention, a signal of a predetermined frequency f (= 1 / T) is received, a predetermined sampling time (nT), and (n + 1/4) T after the predetermined sampling time (n = 0, Any of 1, 2,...: The same in the claims), data sampled at a time after (n + 1/2) T, (n + 3/4) T, and 0 ° sampling data and 90 ° sampling data, respectively. , 180 °
The sampling data is 270 ° sampling data, and the average value of the 0 ° sampling data and the 180 ° sampling data is the real part data of the complex sampling data, and the value obtained by averaging the 90 ° sampling data and the 270 ° sampling data is the complex sampling data. The data is output as imaginary part data of the data.

The present invention is characterized in that the time of (n + ／) T is shorter than １ ／ of the period of the predetermined sampling time.

The present invention is characterized in that signals of a plurality of channels are divided into a plurality of groups, the above-mentioned sampling is performed for each signal of one group at a predetermined time, and this processing is sequentially repeated for a plurality of groups.

The present invention relates to a plurality of channel signal input means and a plurality of multiplexers for multiplexing a plurality of channel signals input from the signal input means into a system having less than the number of channels. A plurality of A / D converters for performing A / D conversion of a signal input from each multiplexer, wherein the sampling timing of each A / D converter is synchronized. And

According to the present invention, there are provided a plurality of channels of signal input means and a plurality of multiplexers for multiplexing a plurality of channels of signals input from the signal input means into a system having less than the number of channels. A plurality of A / D converters for performing AD conversion on signals input from the respective multiplexers, wherein the timing is synchronized, and a plurality of A / D converters for which the sampling timings of the respective A / D converters are synchronized; Phase shifting means for shifting the phase of each sampled data so that the phase relationship of the data obtained is predetermined.

According to the present invention, each multiplexer is provided with each AD.
(A + ／) from a predetermined sampling time in the converter
At a time after T, switching is performed so that a signal of the same channel as the predetermined sampling time is input.
The converter samples the signal of the same channel at a predetermined sampling time and at a time (a +）) T after the predetermined sampling time, and converts the signal into real part data and imaginary part of complex sampling data of the channel. It is characterized in that it is output as data.

According to the present invention, there is provided a sonar apparatus for forming a receiving beam sequentially in different directions to search in a wide range direction, a receiving transducer having a plurality of elements each receiving an echo signal of a predetermined frequency, and each element receiving the echo signal. A multiplexer that multiplexes the obtained echo signals into a system having less than the number of elements, an AD converter that samples the echo signal of each element in each system and outputs complex sampling data, and an A / D converter that outputs complex sampling data of each element. While shifting each sampling data so that the phase relationship becomes predetermined,
A signal processing unit for forming a reception beam in a predetermined direction.

In the present invention, the signal processing unit stores a multiplier for previously multiplying a coefficient for shifting a phase of the complex sampling data of each element and a coefficient for forming a reception beam with the complex sampling data of each element. The multiplier is multiplied by the complex sampling data of each element to simultaneously perform the phase shift and the formation of the received beam.

According to the present invention, there is provided a sonar apparatus for forming a receiving beam sequentially in different directions to search in a wide range direction, comprising a plurality of elements for receiving echo signals of a predetermined frequency f (= 1 / T). A transducer, a multiplexer for multiplexing the echo signals received by each element into a system having less than the number of the elements, and an echo signal of each element in each system at a predetermined sampling time and (n + ／) from the predetermined sampling time. 2) sampling at two times after T, outputting these as complex sampling data of each element, and shifting each sampling data so that the phase relationship between the complex sampling data of each element becomes predetermined. Signal processing to form a receiving beam in a predetermined direction Characterized by comprising a and.

[0017] In the case of sonar exploration sonar or the like, since the Doppler effect hardly works, the frequency of the echo signal is almost the same as the transmission frequency. Therefore, the frequency does not change even if some time passes. Therefore, even if IQ sampling is not performed at the same sampling time, 90
If the phase is shifted and sampling is performed twice with a delay of (a + ／) T, real-part data and imaginary-part data similar to complex sampling can be obtained. Thereby, real part data and imaginary part data can be generated in a time sharing manner without providing two channels of a real part and an imaginary part.

In this case, by setting the interval between the sampling timing of the real part data and the sampling timing of the imaginary part data shorter than the sampling timing of the next real part data as shown in FIG.
That is, by setting the interval between the sampling timing of the real part data and the sampling timing of the imaginary part data to be shorter than の of the sampling period, the case where there is a fluctuation in amplitude as shown in FIG. Even when the frequency is shifted due to an effect or the like, complex sampling data with a small error can be obtained.

The narrower the interval between the sampling timing of the real part data and the sampling timing of the imaginary part data, the better the interval of a = 0, that is, the interval of T / 4. The example of FIG. 11B shows a case where the phase of the sampling data is 45 °, and the sample value at the sampling timing t ₀ of the real part data is 0.4861359 and T
The sample value of t ₁ delayed by / 4 is 0.5155987, and the phase obtained from these values is 43.32 °, which is an error of 0.0026 wavelength. On the other hand, when the sampling intervals of the real part data and the imaginary part data are set at equal intervals (1.25 T in the example of FIG. 1), the following is obtained.
The sample value of the sampling timing t ₁ ′ after 25T is 0.6334498, and the phase obtained based on this and the value of t ₀ is 37.50 °, which is an error of 0.021 wavelength. As described above, by setting the interval between the sampling timing of the real part data and the sampling timing of the imaginary part data to be shorter than サ ン プ リ ン グ of the sampling period, even when the input signal has amplitude fluctuation or frequency deviation, Complex sampling data with few errors can be obtained.

Further, by setting the sampling repetition cycle of the two sampling data (real part data and imaginary part data) to (m + 1/2) T, the phase of the signal is inverted every time the sampling is repeated. The DC bias component can be cancelled. Also, (n
The DC bias component superimposed on the sampling data can also be canceled by averaging using the data of (+1/2) T and the data of (n + 3/4) T.

In the case where signals input from a plurality of signal input means are multiplexed by a multiplexer, if the multiplexers are sequentially switched at predetermined time intervals, noise generated by the switching of the multiplexers may cause other channels to generate noise. May have an adverse effect on the sampling data. For this reason, in the present invention, the switching timing of all multiplexers and the AD converter sampling timing are synchronized to prevent noise from being superimposed on the sampling data.

In a submarine exploration sonar or the like, signals input from a plurality of channels are sequentially sampled at fixed time intervals, and the obliquely arranged data (on the time axis) is input to a matched filter to thereby perform submarine exploration. However, even when sampling is performed at the above-mentioned synchronized timing, a data sequence arranged (obliquely) at the above-mentioned fixed time interval is realized by shifting the phase of the sampled data. Further, data sampled in a stepwise manner may be shifted so as to become data of the same time sample. In this case, the data is 2nπ
The phase shift may be shifted to (nλ). Even in such a data string, 2nπ can be canceled and processed as the same time data.

[0023]

FIG. 1 is a block diagram of a submarine exploration sonar according to an embodiment of the present invention, FIG. 2 is a diagram showing an installation form of a transducer of the submarine exploration sonar, and a transmission beam formed by the transducer. FIG.

First, in FIG. 2, each of the transmission transducer 11 and the reception transducer 122 is composed of an ultrasonic transducer array in which a plurality of ultrasonic transducers are arranged in one line. The transmitting transducer 11 is installed on the bottom of the ship so that the transducers are arranged in the bow and stern directions,
The receiving transducer 12 is installed on the bottom of the ship such that the arrangement direction of the transducers is on the ship side.

At the bottom of the ship, the transmitting transducer 1
1. In addition to the transducer unit 1 comprising the receiving transducer 12, a transmitting / receiving unit 2 for applying an ultrasonic burst signal to the transmitting transducer 11 and receiving a reflected echo and converting it into digital sampling data is provided. The cabin is provided with an arithmetic processing unit 3. The arithmetic processing unit 3 performs beamform detection, seafloor detection, and the like based on the sampling data transmitted and input from the transmission / reception unit 2. For this purpose, a phase shifter 31 shown in FIG. 7 and a beamformer 32 shown in FIG. 8 are provided.

The transmitting circuit 26 of the transmitting / receiving section 2 applies a pulse signal to each element of the transmitting transducer 11. Each element of the transmitting transducer 11
It is driven by this pulse signal and sends out an ultrasonic signal into the sea. The transmission circuit 26 has a built-in oscillator that oscillates a signal of 320 kHz, and controls the timing of each element so that the transmission beam is formed in a fan shape directly below the hull as shown in FIG. 2B. To apply a pulse signal. The transmission beam formed in this manner includes:
The fan shape is 5 °, 150 ° left and right. The pulse width of the pulse signal input to each element is 1 at 320 kHz.
It is about 0 to 50 waves. Since the beam is formed just below, even if the ship is moving, there is almost no effect of the Doppler effect, and the reflected echo from the sea floor is the same as that at the time of transmission.
It becomes a kHz burst wave.

The receiving transducer 12 has a cylindrical shape in which 160 elements are arranged on the circumference as shown in FIG. The transmission / reception unit 2 and the arithmetic processing unit 3 connected to the reception transducer 12 sample the reflected echo received by each element and compare it with a reference by a matched filter to obtain a signal shown in FIG.
As shown in (B), a receiving beam of about 20 ° before and after and about 1.5 ° right and left is formed. The receiving beam is scanned from right to left at a high speed, and this scanning is repeated many times for one pulse transmission to perform the seabed exploration.
In FIG. 3, the receiving transducer 12 has a radius 12
Since it has a cylindrical shape of 5 mm and 160 ultrasonic transducer elements are arranged at 1.5 ° intervals, a central angle of 2
At 38.5 °, a part of the cylinder is cut away.

The signal received by each element of the receiving transducer 12 is input to the transmitting / receiving section 2 as a signal of the corresponding receiving channel. In the transmission / reception unit 2, each channel is amplified by the preamplifier 13, filtered by the filter 14, and amplified by the TVG amplifier 15. Filter 14
Is a band-pass filter that removes frequencies other than frequencies near the frequency (320 kHz) of the ultrasonic beam transmitted from the transmission transducer 11. As described above, the reflected echo signal is a signal in a narrow band of about 320 kHz, and the band-pass filter removes signals such as out-of-band ultrasonic equipment signals and out-of-band sea noise.

The TVG amplifier 15 is a time variable gain amplifier, and increases the gain as time elapses after the transmission transducer 11 emits a burst wave. This is because it is necessary to receive a reflected echo with a long propagation distance and a small signal level, which is reflected at a distance as time elapses after the burst wave is emitted. is there.
A simple filter 16 for removing noise of the TVG amplifier 15 is interposed at the subsequent stage of the TVG amplifier 15. Thereafter, the multiplexer 17 outputs 160
Channel signals are time-division multiplexed into 10 channels. The k-th (= 0 to 9) multiplexer has 10n (=
0-15) + k signals are input. That is, channels 0, 10, 20,...
, 141, 151 to the first multiplexer,..., The ninth multiplexer.
, 149, 159 are input. The 0th to 9th multiplexers sequentially switch the selection n of the input signal at the same timing in synchronization.

The reflected echo signals multiplexed on the ten channels are amplified again by the TVG amplifier 18. A general TVG amplifier has a gain control range of about 40 dB,
If you want to perform a wide range of seafloor exploration, T of 40dB or more
Since a VG range is required, the TVG amplifier has two stages as described above. Even if both TVG amplifiers are arranged at the subsequent stage of the multiplexer, the transient response characteristics cannot be adjusted in time. However, it is necessary to increase the bandwidth after the multiplexer, and noise generated by the first stage TVG amplifier becomes a problem. The first stage TVG amplifier is provided for each channel in front of the multiplexer, and is connected to the multiplexer 17 via a simple filter 16 for limiting amplifier noise.

The signal amplified by the TVG amplifier 18 is A
The data is sampled by the D converter 19 and converted into digital sampling data. The sampling timing of the AD converter 19 and the switching timing of the multiplexer 17 are created based on the signal oscillated by the oscillator of the transmission circuit. That is, the frequency of the transmission pulse (reflected echo signal) is completely synchronized with the multiplexer switching timing and the sampling timing.

FIG. 4 is a diagram for explaining the sampling timing of the AD converter 19.

In the latter processing unit, it is desirable to convert the reflected echo signal into complex data in sampling in order to process it as complex data. However, mixing a cos signal and a sine signal into a real-valued signal to separate them into signals of a real part I and an imaginary part Q and separately sampling the signal complicates the circuit configuration and reduces measurement errors due to phase shift and the like. Cause invites.

Therefore, in this apparatus, by taking advantage of the fact that the frequency of the received reflected echo signal is stable and the sampling clock is completely synchronized with this, sampling is performed twice with a phase difference of 90 °. One is the real part (In-phase) data, and the other is the imaginary part (Qu-phase).
(adrature) data,
Complex sampling data is generated. Further, in this apparatus, the reflected echo signal is sampled four times (0 °, 90 °, 180 °, 270 °) with a phase difference of 90 °, the 0 ° sampling data and the 180 ° sampling data are combined, and 90 ° The DC bias component of the reflected echo signal is removed by combining the sampling data and the 270 ° sampling data.

As described above, since 160 channels are multiplexed into ten systems in a time-division manner, each system has 16 channels.
Channel. In each system, a signal of four channels is sampled at 1λ (one cycle) at a frequency of 320 kHz of a reflected echo, and a signal of 16 channels is sampled at 4λ.

The sampling timing will be described in detail with reference to FIG. Channels 10n + 0, (n = 0,
,..., 15: the same applies hereinafter). Further, a signal of 10n + 1 is selectively input to the AD converter AD1 via the multiplexer 16. Similarly, AD converters ADk, (k = 0, 1,..., 9: the same applies hereinafter)
, A signal of the channel 10n + k is selectively input. Each AD converter is 1 / 16λ (0.195625μ
The input signal is sampled every second). Therefore 1λ
During this period, sampling is performed 16 times.

During the first 1λ, each AD converter ADk
Channel k, channel 10 + k, channel 20+
k, channel 30 + k is switched every sample, and 4
Sample each time. As a result, each channel is sampled four times at intervals of 1 / 16λ × 4 = 1 / 4λ, that is, 90 °, so that each channel is (relatively) 0 °, 90 °, 180 ° 270 °
Can be obtained.

During the next 1λ, each of the AD converters ADk has a channel 40 + k, a channel 50 + k, and a channel 6 + k.
0 + k and channel 70 + k are switched every one sample, and sampling is performed four times. During the next 1λ, each AD converter ADk is connected to channel 80 + k, channel 90 + k, channel 100 + k, channel 110+
k is switched for each sample, and sampling is performed four times. Further, during the next 1λ, each AD converter ADk switches the channels 120 + k, the channels 130 + k, the channels 140 + k, and the channels 150 + k every one sample, and performs sampling four times. In this way,
During 4λ, 4 (0 °, 9) for all channels
0 °, 180 °, 270 °). This is called a 4λ cycle.

Each of the AD converters ADk, (k = 0 to 0)
The sampling timing of 9) is completely synchronized, and the switching of the multiplexer 16 after the sampling is completed is simultaneous. If the AD converter ADk uses a high-speed AD converter of about 20 MHz, only the input immediately before sampling affects the sampling data, and even if the multiplexer is switched immediately after sampling, the switching noise adversely affects the next sampling data. Has no effect.

As described above, immediately after the input signal is sampled, the multiplexer is switched, and the TVG at the subsequent stage is switched.
The response of the amplifier is executed, and by the next sampling timing (after 0.195625 μsec), the noise generated by the operation of the selection signal of the multiplexer and the change of the output data of the AD converter is sufficiently attenuated, and Does not adversely affect sampling. In addition, since the switching of the multiplexers and the AD converters of the ten systems is performed in synchronization as described above, there is no possibility that the switching noise of one system invades another system and adversely affects it.

The sampling data of each channel is input to the averaging circuit 20. The averaging circuit 20 performs an averaging process on a pair of 0 ° sampling data and 180 ° sampling data and a pair of 90 ° sampling data and 270 ° sampling data for each channel. Since it is sampling data whose timing is set by the same clock as the transmission frequency (reflection echo frequency), 0 °
The sampling data and 180 ° sampling data, and the 90 ° sampling data and 270 ° sampling data should have almost the same amplitude level and different polarities. Therefore, (0 ° sampling data−180 ° sampling data) / 2
0 ° sampling data (real part data R) in which the DC offset component has been canceled by averaging
Can be calculated. It should be noted that the DC offset component is generated due to AC coupling with positive / negative asymmetry or an offset error of the AD converter. Further, the 90 ° sampling data and the 270 ° sampling data are also averaged by (90 ° sampling data−270 ° sampling data) / 2, thereby canceling the DC offset component of the 90 ° sampling data (imaginary part data I). ) Can be calculated. The real part data R and the imaginary part data I are output as complex sampling data.

The complex sampling data is sent to the arithmetic processing unit 3 in the cabin through a high-speed link connected by an optical fiber or the like.
Is transmitted to Since the digital processing is performed after the AD converter 19 of the transmission / reception unit 2, the transmission timing of the sampling data does not need to exactly match the timing of the step-shaped broken line a in FIG. From the transmission / reception unit 2 to the arithmetic processing unit 3 so that it can be executed in real time
Should just be input. That is, of the stepped sampling data, for example, the data of channel 0 to the data of channel 9 have the same timing, but the transmission from the transmission / reception unit 2 to the arithmetic processing unit 3 is performed serially. In the processing, these data are processed at the same timing.

As shown in FIG. 3, the receiving transducer 12 has a cylindrical shape with a central angle of 238.5 ° having 160 elements at 1.5 ° intervals. 60 elements in a range of about 90 ° are used. Hereinafter, each element is indicated by a channel number. When a reception beam is formed by channels 0 to 59, the beam direction is the direction between channels 29 and 30. If this direction is 0 °, channel 0 will be in the direction of 44.25 °, Channel 59 is oriented at -44.25 °.

FIGS. 5 and 6 are diagrams showing the principle of the phase shift and the receiving beamform performed by the arithmetic processing unit 3. FIG. The sampling data input from the transmitting / receiving unit 2 to the arithmetic processing unit 3 is at the sampling time as shown by the step-shaped broken line a in FIG. That is, 0 °, 90 °, 180 °,
Although sampling is performed four times at 270 °, 180 ° sampling data and 270 ° sampling data are used to remove a DC offset component, and 90 ° sampling data is used as imaginary part data. The data is input to the arithmetic processing unit 3 as complex sampling data of the data timing.

The arithmetic processing unit 3 forms a reception beam on the continuous 60 channels, and scans this from right to left. That is, 101 receiving beams from the receiving beam (0) of the channels 0 to 59 to the receiving beam (100) of the channels 100 to 159 are continuously formed. In order to form such a continuous reception beam using a matched filter, it is necessary that the temporal relationship between the data of channel 0 to the data of channel 159 be continuous. For this reason, the phase of the data of each channel is shifted so that it can be handled as the data of the timing of the stair-like horizontal line a, diagonal line b or horizontal line c in FIG. An oblique line b is a phase straight line when the sampling time of each sampled data is set at a constant time interval, and continuity from channel 159 data to the next channel 0 data is also ensured. The horizontal line c is a phase straight line when all the sampling times of the sampling data of the channels 0 to 159 are the same. Further, the stair-like horizontal line a indicates the phase when the sampling time of each sampling data is the same for each of channels 0 to 39, channels 40 to 79, channels 80 to 119, and channels 120 to 159. It is a straight line.
The phase difference of each group is shifted so as to be 2π. The phase shift is performed by rotating the phase of the input sampling data to the hatched phase. The timing at which the sampling data is phase-shifted, that is, the positions of the stair-like horizontal line a, oblique line b, and horizontal line c are arbitrary.

In the case of oblique sampling, it is conceivable to perform sampling at the timing of the oblique line b at the time of actual sampling. However, as described above, if the switching timing of the transducer or the AD converter is shifted in each system, other sampling is performed. In this case, the switching timing is aligned on such a staircase because adverse effects such as switching noise are exerted on this system.

The phase-shifted sampling data is compared with the reference in order from channel 0 data. FIG. 6 shows an example of the reference. The reference is
This represents the reception level of each channel when the reflected echo as a parallel wave arrives along the circumference of the cylinder in order from the channel closest to the arrival direction of the reflected echo.
When the sampled data groups from the beam (0) of the channel 0 to the channel 59 to the beam (100) of the channel 100 to the channel 159 are sequentially compared with the reference, a large correlation is found when the beam is in the direction in which the reflected echo actually arrives. , Which indicates that the reflected echo is coming from that direction.

Since the receiving transducer 12 has a radius of 125 mm as described above, the front channels 29 and 30 and the rear channels 0 and 59 in the beam direction are approximately 7. There is a distance of 5 wavelengths. That is, 125 × (1-1 / {2) / (1500/320)}
7.5.

On the other hand, in order to improve the search accuracy, the seafloor sonar, a general sonar device, and the like tend to shorten the pulse width of the burst wave transmitted from the transmission transducer 11, and the pulse width of the reflected echo also increases. Be shorter. When a reflected echo signal arrives from the beam direction and the pulse width of the reflected echo is shorter than the 7.5 wavelength, one reflected echo is used for forming a received beam.
The channels 0 are not simultaneously applied. Therefore, it is assumed that a beam is formed by sampling every four wavelengths along the circumference, and the beamforming is performed by dividing the 7.5 wavelengths into two equal groups.

That is, the rear group 1 (channel 0 to channel 9, channel 53 to channel 5)
9) and the former group 2 (channels 10 to 52). The group 1 uses the current sampling data, and the group 2 uses the data sampled one time earlier to form a beamform (with reference). By performing the comparison, the sampling data of the reflected echoes from all the channels forming the receiving beam can be used, and the detection accuracy can be improved.

The above groups 1 and 2 are groupings when the phase of the sampling data is shifted to oblique sampling. When the phases are shifted to the data of the same timing, the group 1 is divided into channels 0 to 8 and channels Channels 51 to 59 and group 2 are channels 9 to 50.

FIG. 7 is a diagram showing the configuration of the phase shifter 31 of the arithmetic processing unit 3. The 0 ° sampling data input from the transmission / reception unit 2, that is, the real part data of the complex sampling data is input to the multiplier 43 and the multiplier 45. The multiplier 41 is connected to the memory 41 in which the real terms of the phase shift coefficients corresponding to each of the 160 elements are stored.
The memory 45 in which the imaginary terms of the phase shift coefficients corresponding to each of the 160 elements are stored is connected to the multiplier 45. When 0 ° sampling data is input to the multipliers 43 and 45, the sampling data (element)
Are read out from the memories 41 and 42, and are multiplied by the multipliers 43 and 45. The 0 ° sampling data multiplied by the real term of the correction coefficient in the multiplier 43 is output to the adder 47. Also, the multiplier 4
The 0 ° sampling data multiplied by the imaginary term of the correction coefficient by 5 is output to the adder 48.

The 90 ° sampling data, ie, the imaginary part data of the complex sampling data input from the transmission / reception unit 2 is input to the multipliers 44 and 46. The multiplier 41 is connected to the memory 41 storing the real terms of the phase shift coefficients corresponding to the 160 elements. Further, the memory 42 in which the imaginary term of the phase shift coefficient corresponding to each of the 160 elements is stored is connected to the multiplier 46. When the 90 ° sampling data is input to the multipliers 44 and 46, the phase shift coefficients corresponding to the sampling data (elements) are stored in the memories 41 and 4.
2 and these are multiplied in multipliers 44 and 46. Note that the 0 ° sampling data and the 90 ° sampling data are input to the multipliers 43 to 46 in synchronization. 90 ° multiplied by the real term of the correction coefficient in the multiplier 44
The sampling data is output to the adder 48. Also,
The 90 ° sampling data multiplied by the imaginary term of the correction coefficient in the multiplier 46 is output to the adder 47.

The adder 47 converts the 0 ° sampling data multiplied by the real term of the correction coefficient input from the multiplier 43 to the 90 ° sampling data multiplied by the imaginary term of the correction coefficient input from the multiplier 46. Is subtracted to calculate 0 ° correction data, which is input to the beam former 32 at the subsequent stage. Further, the adder 48 includes 0 ° sampling data multiplied by the imaginary term of the correction coefficient input from the multiplier 45 and 90 ° sampling data multiplied by the real term of the correction coefficient input from the multiplier 44. 90 °
The correction data is calculated, and the correction data is
To enter. By the above processing of the phase shifter 31, the phase of the sampling data of each element can be aligned as shown in b or c of FIG.

FIG. 8 shows the beam former 3 of the arithmetic processing unit 3.
FIG. 2 is a diagram illustrating a configuration of a second embodiment. This beamform section is composed of a complex matched filter. The 0 ° sample time series, that is, the 0 ° correction data is sequentially input to a shift register 51 of 60 stages, a shift register 52 of 107 stages, and a shift register 53 of 43 stages. In addition, the above 90 °
The correction data is sequentially input to a 60-stage shift register 61, a 107-stage shift register 62, and a 43-stage shift register 63.

[0056] In the figure, R _N, R _O represents a 0 ° sampling data (real data), R _N is data sampled input current 4λ cycle, R _O is sampled input in the preceding 4λ cycle The data is shown.
Further, I _N and I _O are 90 ° sampling data (imaginary part data), I _N indicates data sampled and input in the current 4λ cycle, and I _O indicates data sampled and input in the previous 4λ cycle. And C _R , C _I
Indicates a reference coefficient (complex matched data) of the complex matched filter, C _R indicates a real part coefficient of the reference, and C _I indicates an imaginary part coefficient of the reference. The subscripted numbers are the element (channel) numbers in the beam. Incidentally, the reference coefficient C _R, C _I is those fixed represented by numbers 0 to 59, the sampling data R _{N inputted, R O, I N, I} O is 0 in FIG. 5
Although the number 9 is assigned, the data of channels 0 to 159 are sequentially shifted and input.

As shown, the matched filters are RR,
It consists of four series: IR, RI, and II. RR is R
_This is a filter for calculating the degree of correlation between _N , R _O (real part data) and C _R (real part coefficient). The 60 multipliers 55 use the reference coefficient C _R and its timing (beam direction).
And the corresponding 0 ° sampling data is multiplied, and the adder 56 adds the multiplication result. Also, II
This filter calculates the degree of correlation between I _N , I _O (imaginary part data) and C _I (imaginary part coefficient), and 60 multipliers 57 use the reference coefficient C _I and its timing (beam direction). Multiplication with the corresponding 90 ° sampling data is performed, and the adder 58 adds the multiplication result. Adder 55
RR system filter output (RR) and II system filter output (II) are input to the subtractor 71, and the operation of (RR)-(II) is performed, and the real part of the complex sampling data and the complex reference The correlation value of the phase with the real part of the coefficient is calculated. That is,

[0058]

(Equation 1)

Is performed, and the correlation between the real part data and the real part coefficient is calculated.

On the other hand, IR is a filter for calculating the degree of correlation between I _N , I _O (imaginary part data) and C _R (real part coefficient), and 60 multipliers 65 use the reference coefficient C _R
Is multiplied with the corresponding 90 ° sampling data at that timing (beam direction), and the adder 66 adds the multiplication result. R · I is a filter for calculating the degree of correlation between R _N , R _O (real part data) and C _I (imaginary part data), and 60 multipliers 67 use reference coefficients C _I and 0 corresponding to the timing (beam direction)
° Multiplication with the sampling data is performed, and the adder 68 adds the result of the multiplication. The addition result of the adder 65, that is, the filter output (IR) of the IR system and the filter output (RI) of the RI system are input to the adder 72 (IR) +
The calculation of (RI) is performed, and the correlation value of the phase between the real part of the complex sampling data and the real part of the complex reference coefficient is calculated. That is, the operation of [Equation 1] is executed.

The operation results of the subtractor 71 and the adder 72 are input to the amplitude detector 73. The amplitude detector 73
The amplitude of the received beam is obtained based on the calculation result. This amplitude is

[0062]

(Equation 2)

In the case of hardware processing, a table or a circuit for approximation processing may be used.
The extraction circuit 74 is necessary because there are no elements on the entire circumference of the receiving transducer, and extracts 101 beams of clocks 59 to 159 of the shift register. This 101 beam is applied to channels 29-30 as described above.
There are 101 beams from the beam (0) in the direction between the channels to the beam (100) in the direction between the channels 129 and 130. This process is repeated every 4λ cycle of the 320 kHz pulse wave.

In this embodiment, the beam former 3
2, a phase shifter 31 is separately provided in the preceding stage, but the phase shifter for shifting each sampling data may be included in the reference coefficient of the beamformer 32, and the phase shifter 31 may be omitted. It is. In this case, when the phase shifter 31 and the beamformer 32 are separated, the complex matched data (reference coefficient) may be one for all beams as shown in FIG.
In the case where the function of the phase shifter 31 is also used, the complex matched data corresponding to the beam to be formed having complex matched data for each beam is read out as shown in FIG. Multiply 0 ° sampling data and 90 ° sampling data. When such multiplication is performed, the arrangement of a plurality of elements may not have regularity. Any of a one-dimensional array, a two-dimensional array, and a three-dimensional array may be used. For example, they may be arranged in a cylinder as shown in FIG. 13, or may be arranged on the surface of a sphere. Further, the intervals between the elements do not have to be equal.

In the above embodiment, as shown in FIG. 4, multiplexing is performed using ten multiplexers and AD converters, and data of 160 channels is sampled so as to be repeated at a cycle of 4λ. Alternatively, sampling can be performed using a sampling pattern as shown in FIG. The sampling pattern of FIG.
160 channels are multiplexed into 7 systems, each with 0 °,
Sampling is performed at 90 ° and is repeated at a period of 3.5λ. Real part data is only 0 ° data, imaginary part data is 90 °
Since the data is only data, the phase is inverted every time the data is repeated, so that a DC bias component hardly occurs. The remaining bias component can be removed by signal processing. Further, the sampling pattern of FIG.
The signals are multiplexed into the system and are sampled at 0 ° and 90 °, respectively, but are repeated at a period of 4.5λ similarly to the pattern of FIG.

In the above embodiment, a so-called cross fan beam type sonar device has been described. However, the present invention is not limited to this, and is applied to, for example, a scanning sonar using a cylindrical or spherical transducer (transducer / receiver). You can also.

FIG. 12 is a block diagram of a scanning sonar to which the present invention is applied, and FIG. 13 is a diagram for explaining a data string input to a beamformer in the scanning sonar. In this figure, the same components as those in the block diagram shown in FIG. Each element 27 of the transducer is a trap circuit 28
Are connected to the transmitting circuit 26 and the amplifier 13 of the receiving circuit. Although only one channel is shown between the amplifier 13 and the filter 16, they are provided corresponding to the respective elements. One multiplexer 17 and one AD converter 19 are provided for a plurality of channels. AD converter 1
The data sampled by 9 is input to the beamformer 32 in the order shown in FIG. The beamformer 32 stores complex matched data for each beam to be formed in the same manner as in FIG.
The complex matched data corresponding to the data string (element number) input to the above is read and multiplied by each sampling data to form a reception beam. Note that a plurality of types of complex matched data may be prepared according to the tilt angle, the beam direction, and the beam width.

[0068]

As described above, according to the present invention, (n +
1/4) By sampling twice with the T phase shifted and using the real part data and the imaginary part data, it is possible to generate complex sampling data without increasing the configuration of AD converters and mixers. become.

According to the present invention, by using the 180 ° sampling data and the 270 ° sampling data, even if the DC bias component is superimposed on the data, it can be removed.

According to the present invention, electrical noise is generated when a multiplexer or the like is switched and adversely affects sampling of other systems. However, by switching all the multiplexers simultaneously, any system can be switched when a noise occurs. Sampling is not performed, and no system is switched at the time of sampling, so that the influence of noise can be eliminated. Further, in the present invention, continuous sampling can be performed by a matched filter or the like by aligning the sampling times, which are stepped by such sampling, at the same time or obliquely.

[Brief description of the drawings]

FIG. 1 is a block diagram of a seabed exploration sonar according to an embodiment of the present invention.

FIG. 2 is a diagram showing a mounting form of a transducer of the seabed exploration sonar and a transmission beam and a reception beam.

FIG. 3 is a diagram showing a configuration of a receiving transducer of the seafloor sonar.

FIG. 4 is a diagram showing a sampling timing chart of an AD converter of the seafloor sonar.

FIG. 5 is a diagram illustrating a phase shift method of an arithmetic processing unit of the seabed exploration sonar.

FIG. 6 is a diagram illustrating a beam forming method of the arithmetic processing unit.

FIG. 7 is a diagram showing a configuration of a phase shifter of the arithmetic processing unit.

FIG. 8 is a diagram illustrating a configuration of a beam former of the arithmetic processing unit.

FIG. 9 is a diagram illustrating complex matched data stored in a beamformer.

FIG. 10 is a diagram showing a timing chart of another method of sampling timing of the AD converter.

FIG. 11 is a diagram showing a timing chart of another method of the sampling timing of the AD converter.

FIG. 12 is a block diagram of a scanning sonar to which the present invention is applied.

FIG. 13 is a diagram illustrating a sampling data sequence of the scanning sonar.

FIG. 14 is a diagram illustrating a change in an occurrence error due to a difference in sampling timing.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 ... Transducer part 2 ... Transmission / reception part 3 ... Operation processing part 11 ... Transmission transducer 12 ... Reception transducer 13 ... Preamplifier 14 ... Bandpass filter 15 ... TVG amplifier 16 ... Bandpass filter 17 ... Multiplexer 18 ... TVG amplifier 19 ... AD converter Reference Signs List 20 averaging processing unit 31 Phaser 32 Beamformer 41, 42 (Memory in which phase coefficient is stored) 43-46 Multipliers 47, 48 Adders 51, 61 60-stage shift registers 52, 62 ... 107-stage shift register 53, 53 ... 43-stage shift register 55, 57, 65, 67 ... multipliers 56, 58, 66, 68 ... adder 71 ... subtracter 72 ... adder 73 ... amplitude detector 74 ... take-out circuit

 ──────────────────────────────────────────────────続 き Continuing on the front page (72) Inventor Yasushi Nishimori 9-52, Ashihara-cho, Nishinomiya-shi, Hyogo Furuno Electric Co., Ltd. (72) Inventor Mitsuaki Watanabe 9-52, Ashihara-cho, Nishinomiya-shi, Hyogo Furuno Electric Co. In the formula company

Claims (12)

[Claims] 1. A signal of a predetermined frequency f (= 1 / T) is received, and a predetermined sampling time and a time (a = 0, 0.5, 1, 1) after (a + ／) T after the predetermined sampling time.
1.5, the same as in the claims), and outputs the data sampled at these times as real part data and imaginary part data of complex sampling data. 2. The signal processing method according to claim 1, wherein the time of (a + ／) T is shorter than の of the period of the predetermined sampling time. 3. The method according to claim 1, wherein the two samplings are (m + 1 /
2) The signal processing method according to claim 1 or 2, wherein the method is repeatedly performed for each T (m = 1, 2,...). 4. A signal of a predetermined frequency f (= 1 / T) is received, and a predetermined sampling time (nT) and (n + ／) T after the predetermined sampling time (n = 0, 1, 1)
2, any of: ... the same in the claims),
The data sampled at the time after (n + 1/2) T and after (n + 3/4) T are defined as 0 ° sampling data, 90 ° sampling data, 180 ° sampling data, and 270 ° sampling data, respectively. A signal processing method for outputting a value obtained by averaging the sampled data and the 180 ° sampling data as real part data of complex sampling data, and outputting a value obtained by averaging 90 ° sampling data and 270 ° sampling data as imaginary part data of the complex sampled data. 5. The signal processing method according to claim 4, wherein the time of (n + ／) T is shorter than １ ／ of a cycle of the predetermined sampling time. 6. A signal of a plurality of channels is divided into a plurality of groups, and the sampling according to any one of claims 1 to 5 is performed at predetermined time for each signal of one group, and this processing is performed for a plurality of groups. A signal processing method performed repeatedly and sequentially. 7. A plurality of channel signal input means, and a plurality of multiplexers for multiplexing the plurality of channel signals input from the signal input means into a system having less than the number of channels, wherein the switching timing of each multiplexer is A signal processing device comprising: a synchronous device; and a plurality of A / D converters for performing A / D conversion of a signal input from each multiplexer, wherein the sampling timing of each A / D converter is synchronized. 8. A signal input means of a plurality of channels, and a plurality of multiplexers for multiplexing the signals of the plurality of channels input from the signal input means into a system having less than the number of channels, wherein the switching timing of each multiplexer is A plurality of A / D converters for performing A / D conversion of a signal input from each multiplexer, wherein a sampling timing of each A / D converter is synchronized; and a data sampled by the A / D converter And a phase shifting means for shifting the phase of each sampled data so that the phase relationship of (i) becomes predetermined. 9. Each of the multiplexers includes:
At a time after (a + ／) T from the predetermined sampling time, switching is performed so that a signal of the same channel as the predetermined sampling time is input. Each AD converter performs the predetermined sampling time and the predetermined sampling time. 8. The signal of the same channel is sampled at a time (a + ／) T after the sampling time, and this is output as real part data and imaginary part data of complex sampling data of the channel.
The signal processing device according to claim 1. 10. A sonar device for forming a receiving beam sequentially in different directions to search a wide range direction, comprising: a receiving transducer having a plurality of elements each receiving an echo signal of a predetermined frequency; and an echo received by each element. A multiplexer for multiplexing signals into systems having less than the number of elements, an AD converter for sampling echo signals of each element in each system and outputting complex sampling data, and a phase relationship between the complex sampling data of each element And a signal processing unit that forms a reception beam in a predetermined direction while shifting the phase of each sampled data so that is predetermined. 11. The signal processing unit stores a multiplier for preliminarily multiplying a coefficient for shifting a phase of the complex sampling data of each element and a coefficient for forming a reception beam with the complex sampling data of each element. 10. The sonar device according to claim 9, wherein the multiplier is multiplied by the complex sampling data of each element to simultaneously perform the phase shift and the formation of the received beam. 12. A sonar device for forming a receiving beam sequentially in mutually different directions to search in a wide range direction, comprising: a receiving transducer having a plurality of elements each receiving an echo signal of a predetermined frequency f (= 1 / T); A multiplexer for multiplexing the echo signals received by each element into a system having less than the number of elements, and a multiplexer which divides the echo signal of each element in each system from a predetermined sampling time and from the predetermined sampling time to (n
An AD converter that samples at two times after (+) T and outputs them as complex sampling data of each element; and converts each sampling data so that the phase relationship between the complex sampling data of each element becomes predetermined. And a signal processing unit that forms a received beam in a predetermined direction while shifting the phase.