JP2001031339A - Elevator controller - Google Patents

Abstract

PROBLEM TO BE SOLVED: To restrain the worsening of comfortableness of an elevator even in case torque generated by a permanent magnet synchronous motor driving an elevator pulsates. SOLUTION: A speed control system 6 computes a torque current command value and a magnetic flux current command value based on a signal of speed deviation of the actual speed of a permanent magnet synchronous motor 3 from a speed command value and also computes the primary frequency angle of an inverter 7 based on the actual speed. A torque ripple elimination control computer 19 computes a torque ripple compensation value for restraining torque ripple of the permanent magnetic synchronous motor 3 based on the torque current command value and the primary frequency angle obtained from the speed control system 6 and outputs the torque current command value including the compensation value to a current control system 8. The current control system 8 outputs a gate signal to the inverter 7 so as to obtain the torque current command value to which the torque ripple compensation value is included.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an elevator control device for controlling an elevator by driving a permanent magnet synchronous motor connected to a hoist with an inverter.

[0002]

2. Description of the Related Art Permanent magnet synchronous motor PMSM (Permanent Ma
The gnet Synchronus Motor) has recently been used as a drive unit of an elevator system because it can be manufactured in a smaller size for the same output than an induction motor. The permanent magnet synchronous motor is driven by a variable voltage frequency AC power supplied from an inverter.

FIG. 11 is a configuration diagram of an elevator control device that controls an elevator by driving a permanent magnet synchronous motor with an inverter. The elevator car 1 is connected to a counterweight 2 by a main rope, and is suspended via a pulley. The pulley is driven by the permanent magnet synchronous motor 3, and the car 1 is moved up and down. A pulse generator 4 (hereinafter referred to as PG4) is attached to the pulley, and detects the rotation speed of the permanent magnet synchronous motor 3 by a pulse signal. That is, the rotation speed of the permanent magnet synchronous motor 3 is detected by a pulse signal via the PG 4 provided on the pulley. Also,
The car 1 is provided with a load detector 5 for detecting a car load.

[0004] The pulse signal detected by the PG 4 and the car load detected by the load detector 5 correspond to the car 1 of the elevator.
Is input to a speed control system 6 for controlling the speed of the motor. The speed control system 6 outputs a gate signal to the inverter 7 by adding a car load to a speed deviation signal between the actual speed of the permanent magnet synchronous motor 3 and the speed command value, and outputs a current command value to the current control system 8. Calculate Idc and Iqc.

The current command values Idc and Iqc are a d-axis current command value Idc and a q-axis current command value Iqc in an orthogonal rotating coordinate system when the permanent magnet synchronous motor 3 is vector-controlled. The d-axis current command value Idc is a magnetic flux current command value for generating a magnetic flux, and the q-axis current command value Iqc is a torque current command value for generating a torque. The speed control system 6 calculates the actual speed of the permanent magnet synchronous motor 3 based on the pulse signal input from the PG 4, and calculates the primary frequency angle θe of the inverter 7 based on the actual speed.

In the current control system 8, a d-axis current command value Idc, a q-axis current command value Iqc, and a primary frequency angle θe in the orthogonal rotation coordinate system are input from the speed control system 6 and input via the current sensor 9. Output current Iu of inverter 7
A gate signal is output to the inverter 7 so that f and Iwf satisfy the d-axis current command value Idc, the q-axis current command value Iqc, and the primary frequency angle θe. Thereby, the permanent magnet synchronous motor 3 is driven and controlled based on the current command value from the speed control system 6.

[0007]

However, the permanent magnet synchronous motor 3 used in such an elevator control device
One of the embodiments is a thin flat structure. In the permanent magnet synchronous motor 3 having this structure, a change in magnetic flux when the magnetic pole rotates is distorted due to the effect of slot ripple.
That is, the magnetic flux change, which should be sinusoidal, is distorted due to the effect of slot ripples, which causes the torque generated by the permanent magnet synchronous motor 3 to pulsate, thereby deteriorating the ride comfort of the elevator. Has become. Further, torque ripple generated due to an unbalance of the winding of the permanent magnet synchronous motor 3 or an error in a sensor system for detecting current also causes torque ripple.

Hereinafter, slot ripple will be described by taking as an example a case where the number of slots is six times the number of poles. FIG. 12 shows the armature winding 11 formed in the slot of the iron core 10 of the permanent magnet synchronous motor 3 and the armature winding 1 as the magnetic pole moves.
FIG. 2 is an explanatory diagram showing a relationship of a magnetic field generated in FIG. Theoretically, the spatial distribution of the magnetic flux generated by the armature winding current is sinusoidal as shown by the characteristic curve S1, but actually the number of the armature windings 11 is finite, so that the characteristic curve S2
The magnetic flux changes almost stepwise as shown in FIG.

FIG. 13 is a characteristic diagram showing the relationship between the actual magnetic flux and the magnetic pole position over time. As the magnetic pole passes through the slot, the actual magnetic flux distribution changes while changing the step shape as shown in FIG. 13 every time. This causes the torque generated as slot ripple to be distorted. In the induction motor, measures such as winding skew are taken, and the level is practically suppressed. However, the skew effect is reduced in the permanent magnet synchronous motor 3 having a thin structure, and the skew effect is reduced. Slot ripple occurs.

FIG. 14 shows the relationship between the fundamental component and the harmonic component of the actual magnetic flux distribution. 14 (a), 14 (b) and 14 (c), it is assumed that the fundamental wave component proceeds to the left with time, and the frequency of the fifth / seventh harmonic component with respect to the fundamental wave is 5/7 times. It is assumed that the phase changes by about 360 ° while the fundamental wave travels by 60 °.

In the state shown in FIG. 14A, the magnetic pole is at a position of 180 ° in electrical angle. At this time, the phase of the magnetic flux of the fifth / seventh order component, which is the harmonic component, matches the fundamental wave. ,
The magnetic flux at the magnetic pole position is zero. At this time, the magnetic pole is located between the slots. FIG.
In the state shown in FIG. 14C in which the magnetic pole position has advanced by 30 ° in electrical angle from the state shown in FIG. 14A, the 5th / 7th order component is 18 electrical degrees.
0 ° changes. Even in this state, the fifth order / 7 at the magnetic pole position
The magnetic flux of the next component is zero. The magnetic pole is at the position of the slot.

Next, the state shown in FIG.
The state during the transition to the state (c), that is, FIG.
In the state (b), the magnetic flux of the fifth / seventh-order component at the magnetic pole position is not 0, and therefore, the total magnetic flux due to the influence of the magnetic pole position increases, and the generated torque becomes maximum. This causes a six-fold component (6f component) of the fundamental frequency of the torque pulsation.

An object of the present invention is to provide an elevator control device capable of suppressing deterioration of the ride quality of an elevator even when the torque generated by a permanent magnet synchronous motor driving the elevator pulsates.

[0014]

An elevator control apparatus according to the first aspect of the present invention supplies a variable voltage frequency AC power supply from an inverter to a permanent magnet synchronous motor to drive a hoist connected to the permanent magnet synchronous motor. An elevator control device that controls the elevator by calculating a torque current command value and a magnetic flux current command value based on a speed deviation signal between the actual speed of the permanent magnet synchronous motor and the speed command value, and based on the actual speed. A speed control system for calculating a primary frequency angle of the inverter by using a current control to output a gate signal to the inverter so that an output current of the inverter satisfies a torque current command value and a magnetic flux current command value at the primary frequency angle. And a torque ripple of the permanent magnet synchronous motor based on the torque current command value and the primary frequency angle. Characterized in that a calculated torque ripple compensation value torque ripple removal control calculator for adding the torque current command value for win.

In the elevator control apparatus according to the first aspect of the present invention, the speed control system converts the torque current command value and the magnetic flux current command value based on a speed deviation signal between the actual speed of the permanent magnet synchronous motor and the speed command value. The calculation is performed, and the primary frequency angle of the inverter is calculated based on the actual speed. The torque ripple elimination control calculator calculates a torque ripple compensation value for suppressing torque ripple of the permanent magnet synchronous motor based on the torque current command value from the speed control system and the primary frequency angle, and performs current control in consideration of the torque current command value. Output to the system. The current control system outputs a gate signal to the inverter so that the torque current command value takes into account the torque ripple compensation value.

In the elevator control apparatus according to a second aspect of the present invention, in the first aspect of the invention, the torque ripple elimination control computing unit adds the torque ripple compensation value to the magnetic flux current command value instead of the torque current command value. It is characterized by doing.

In the elevator control apparatus according to the second aspect of the present invention, a gate signal is output to the inverter so that the magnetic flux current command value takes into account the torque ripple compensation value.

According to a third aspect of the present invention, in the elevator control device according to the first aspect of the present invention, the torque ripple elimination control arithmetic unit adds the torque ripple compensation value to the primary frequency angle instead of the torque current command value. It is characterized by the following.

According to a third aspect of the present invention, in the elevator control device according to the first aspect of the present invention, the torque ripple elimination control arithmetic unit outputs a gate signal to the inverter so as to have a primary frequency angle to which a torque ripple compensation value is added. I do.

According to a fourth aspect of the present invention, in the elevator control apparatus according to the first aspect of the present invention, the torque ripple elimination control computing unit computes an amplitude having a value proportional to the torque current command value from the primary frequency angle. And calculating a sine wave having a phase of N times the fundamental frequency as the torque ripple compensation value.

In the elevator control apparatus according to the fourth aspect of the invention, the torque ripple compensation value calculated by the torque ripple elimination control calculator has a value proportional to the torque current command value as an amplitude, and the fundamental frequency calculated from the primary frequency angle. Is represented by a sine wave whose phase is N times the component of

According to a fifth aspect of the present invention, in the elevator control apparatus according to the fourth aspect of the present invention, the torque ripple elimination control calculator is configured to adjust the torque ripple compensation value in accordance with an unbalanced load torque between an elevator car and a counterweight. Is characterized by having variable amplitude.

In the elevator control apparatus according to the fifth aspect of the present invention, the amplitude of the torque ripple compensation value is made variable in accordance with the unbalanced load torque between the elevator car and the counterweight, and the vibration is suppressed in accordance with the car load. .

According to a sixth aspect of the present invention, in the elevator control apparatus according to the fourth aspect, the torque ripple elimination control computing unit is configured to adjust the torque ripple compensation value in correspondence with an unbalanced load torque between the elevator car and a counterweight. Is characterized by having a variable phase.

In the elevator control apparatus according to the present invention, the phase of the torque ripple compensation value is made variable in accordance with the unbalanced load torque between the elevator car and the counterweight, and the vibration is suppressed in accordance with the car load. .

According to a seventh aspect of the present invention, in the elevator control apparatus according to the fourth aspect, the torque ripple elimination control computing unit varies the amplitude of the torque ripple compensation value according to the actual speed of the permanent magnet synchronous motor. It is characterized by doing.

In the elevator control apparatus according to the present invention, the amplitude of the torque ripple compensation value is made variable in accordance with the actual speed of the permanent magnet synchronous motor, and stable vibration suppression control is performed within a controllable range.

According to an eighth aspect of the present invention, in the elevator control apparatus according to the fourth aspect, the torque ripple elimination control calculator is set in advance in correspondence with an unbalanced load torque between the elevator car and the counterweight. An amplitude and a phase in the torque ripple compensation value are selected, and the amplitude and the phase in the torque ripple compensation value are made variable.

According to an eighth aspect of the present invention, in the elevator control device, the amplitude in the torque ripple compensation value is determined based on the amplitude and the phase in the torque ripple compensation value predetermined in correspondence with the unbalanced load torque between the elevator car and the counterweight. And make the phase variable.

[0030]

Embodiments of the present invention will be described below. FIG. 1 is a block diagram of an elevator control device according to a first embodiment of the present invention. The elevator car 1 is connected to a counterweight 2 by a main rope, and is suspended from a pulley driven by a permanent magnet synchronous motor 3.
Is moved up and down.

The rotational speed of the permanent magnet synchronous motor 3 is detected by a pulse signal via the PG 4 and the car load of the car 1 is detected by the load detector 5 to control the speed of the car 1 of the elevator. Input to the system 6. The speed control system 6 calculates the actual speed of the permanent magnet synchronous motor 3 from the pulse signal input from the PG 4, adds a car load to a speed deviation signal between the actual speed and the speed command value, and sends the signal to the current control system 8. Of the current command values Idc and Iqc. Further, the primary frequency angle θe of the inverter 7 is calculated based on the actual speed of the permanent magnet synchronous motor 3.

Current command values Idc, Iq to current control system 8
c is a d-axis current command value (magnetic flux current command value) Idc and a q-axis current command value (torque current command value) in the orthogonal rotation coordinate system when the permanent magnet synchronous motor 3 is vector-controlled.
Iqc.

In the current control system 8, the d-axis current command value Idc, the q-axis current command value Iqc, and the primary frequency angle θe in the orthogonal rotation coordinate system are input from the speed control system 6 and input via the current sensor 9. Output current Iu of inverter 7
A gate signal is output to the inverter 7 so that f and Iwf satisfy the d-axis current command value Idc, the q-axis current command value Iqc, and the primary frequency angle θe.

That is, the output current Iu of the inverter 7
f and Iwf are detected by the current sensor 9 and input to the three-phase / two-phase converter 12 of the current control system 8. The three-phase / two-phase converter 12 converts the current signals Iuf and Iwf represented by three phases in the stationary coordinate system from the current sensor 9 into current signals Ix and Iy represented by two phases in the orthogonal stationary coordinate system.

The two-phase current signals Ix and Iy converted by the three-phase / two-phase converter 12 are input to the dq converter 13 and are converted into current signals Idf and Iqf of a rectangular rotating coordinate system.
That is, the dq converter 13 converts the current signals Ix and Iy of the orthogonal stationary coordinate system into the current signals Idf and Iqf of the orthogonal rotating coordinate system based on the primary frequency angle (electrical angle) θe output from the speed control system 6. And adders 14a, 1
4b.

On the other hand, the d-axis current command value Idc in the orthogonal rotation coordinate system from the speed control system 6 is input to the adder 14a, and the adder 14a outputs the d-axis current signal Idf and the d-axis current command value Idc. A current deviation signal is calculated, and similarly, a current deviation signal between the q-axis current signal Iqf and the q-axis current command value Iqc is calculated by the adder 14b.

The q-axis current command value Iqc input to the adder 14b is converted by the adder 14c into the speed control system 6
Is the q-axis current command value Iqc obtained by adding the torque ripple compensation value from the torque ripple removal control calculator 19 to the q-axis current command value Iqc from. Torque ripple elimination control calculator 1
9 calculates a torque ripple compensation value for removing torque ripple generated by the permanent magnet synchronous motor 3. Based on the q-axis current command value Iqc output from the speed control system 6 and the primary frequency angle θe, a permanent magnet A torque ripple compensation value for suppressing torque ripple of the synchronous motor 3 is calculated. Then, the torque ripple compensation value is added to the adder 14.
At c, the value is added to the q-axis current command value Iqc from the speed control system 6. That is, the q-axis current command value I from the speed control system 6
A new q-axis current command value Iqc is created by adding the torque ripple compensation value to qc, and vibration suppression control is performed as an input to the adder 14b.

The current deviation signals obtained by the adders 14a and 14b are input to PI controllers 15a and 15b, respectively, and are PI-calculated (proportional-integral) to obtain d-axis voltage command values Vdc and q in the orthogonal rotating coordinate system. A shaft voltage command value Vqc is generated.

The d-axis voltage command value Vdc and q-axis voltage command value Vqc from the PI controllers 15a and 15b
Is input to the inverse dq converter 16. Inverse dq converter 16
Then, the d-axis voltage command value Vdc and the q-axis voltage command value Vq
c is converted into voltage command values Vx and Vy in the orthogonal stationary coordinate system. That is, based on the primary frequency angle (electrical angle) θe from the speed control system 6, the voltage command value Vd of the orthogonal rotation coordinate system
c and Vqc are converted into voltage command values Vx and Vy in the orthogonal stationary coordinate system, and output to the two-phase / three-phase converter 17. The two-phase / three-phase converter 17 converts the voltage command values Vx, Vy of the rectangular stationary coordinate system represented by two phases into the voltage command values Vu, Vv, Vw of the rectangular stationary coordinate system represented by three phases. Then, the PWM circuit 18 creates and outputs a gate signal to the inverter 7 by pulse width modulation of the voltage command signals Vu, Vv, Vw.

As described above, the torque ripple elimination control calculator 19 calculates the torque ripple compensation value and adds it to the q-axis current command value (torque current command value) Iqc to generate the permanent magnet synchronous motor 3 Suppresses torque distortion (vibration component).

FIG. 2 shows a torque ripple elimination control calculator 19.
FIG. 3 is a block diagram showing an example of the configuration. The primary frequency angle θe output from the speed control system 6 is input to the N-times period calculator 20 of the torque ripple elimination control calculator 19, and a phase (_Nf_θe) N times the primary frequency angle θe is calculated. For example, a six-fold phase (_6f_θe) is calculated.

This is because torque ripple of the permanent magnet synchronous motor 3 is often caused by a component six times the fundamental frequency (6f component). That is, the slot ripple is determined by the relationship between the number of slots and the number of poles of the motor. For example, if the number of motor slots is 72 and the number of poles is 12, the slot ripple is one of the supply frequencies to the permanent magnet synchronous motor 3.
Double. The N-times phase (_Nf_θe) calculated by the N-times period calculator 20 is used as the sine wave function unit 2 as the sine wave phase Q.
1 is input.

On the other hand, the q-axis current command Iqc output from the speed control system 6 is input to the absolute value circuit 22 of the torque ripple elimination control calculator 19 and converted into an absolute value, and the obtained q
The absolute value of the shaft current command value Iqc is multiplied by a gain constant a (_Nf_Gain_FIX) for setting the amplitude of the sine wave in the multiplier 23 to obtain the amplitude value A. Then, the amplitude value A is input to the sine wave function unit 22. This is slot ripple 6
This is because the f component becomes a sin component based on the slot position.

The sine wave function unit 21 outputs a sine wave AsinQ having the output of the multiplier 23 as the amplitude A and the output of the N-times period calculator 20 as the phase Q. This output signal (sine wave Asin
Q) passes through a limiter circuit 24 and a filter circuit 25 and is output as a torque ripple compensation value α (Nf_Compen). This torque ripple compensation value α is the q-axis current command value I
qc.

According to the first embodiment, a sinusoidal torque compensation value α of an N-times component of the fundamental frequency is calculated and added to the q-axis current command value (torque current command value) Iqc. During traveling of the elevator, distortion (vibration component) of the generated torque due to the effect of the slot ripple can be suppressed.

Next, a second embodiment of the present invention will be described. FIG. 3 is a block diagram of an elevator control device according to a second embodiment of the present invention. The second embodiment is different from the first embodiment shown in FIG. 1 in that the torque ripple elimination control calculator 19 replaces the torque current command value (q-axis current command value) Iqc with the magnetic flux current command value. (D-axis current command value) The torque ripple compensation value is added to Idc. The other configuration is the same as that of the first embodiment shown in FIG. 1, and therefore, the same components are denoted by the same reference symbols and overlapping description will be omitted.

In FIG. 3, the torque ripple elimination control calculator 19 determines the torque of the permanent magnet synchronous motor 3 based on the q-axis current command value (torque current command value) Iqc output from the speed control system 6 and the primary frequency angle θe. A torque ripple compensation value for suppressing the torque ripple is calculated. Then, the torque ripple compensation value is added to the speed control system 6 by the adder 14c.
To the d-axis current command value (magnetic flux current command value) Idc. Thus, a new d-axis current command value (magnetic flux current command value) Idc is created by adding the torque ripple compensation value to the d-axis current command value (magnetic flux current command value) Idc from the speed control system 6, and the adder 14a Vibration suppression control is performed as an input.

By this processing, the d-axis current command value (magnetic flux current command value) Idc includes a component for suppressing N-fold vibration, and the output from the inverter 7 to the permanent magnet synchronous motor 3 by the d-axis current control. Since the current is controlled by including the suppression component, the distortion of the torque generated by the permanent magnet synchronous motor 3 can be suppressed.

Next, a third embodiment of the present invention will be described. FIG. 4 is a block diagram of an elevator control device according to a third embodiment of the present invention. The third embodiment differs from the first embodiment shown in FIG. 1 in that the torque ripple elimination control calculator 19 replaces the torque current command value (q-axis current command value) Iqc with the primary frequency angle θe. And a torque ripple compensation value. The other configuration is the same as that of the first embodiment shown in FIG. 1, and therefore, the same components are denoted by the same reference symbols and overlapping description will be omitted.

In FIG. 4, a torque ripple elimination control calculator 19 calculates the torque of the permanent magnet synchronous motor 3 based on the q-axis current command value (torque current command value) Iqc output from the speed control system 6 and the primary frequency angle θe. A torque ripple compensation value for suppressing the torque ripple is calculated. The torque ripple compensation value calculated by the torque ripple removal control calculator 19 is added to the primary frequency angle θe from the speed control system 6 by the adder 14c to create a new primary frequency angle θe.
The new primary frequency angle θe to which the torque ripple compensation value has been added is calculated by the dq converter 13 and the inverse dq converter 16.
Is input to

By this processing, the primary frequency angle θe includes a component for suppressing N-fold vibration. This means that the d-axis voltage command value Vdc and the q-axis voltage command value Vqc are converted by the inverse dq converter 16 and the 2-phase / 3-phase converter 17 into the three-phase voltage command values Vu, Vv, Vw of the orthogonal stationary coordinate system. Is converted to suppress vibration. Therefore, distortion of the torque generated by the permanent magnet synchronous motor 3 can be suppressed.

FIG. 5 is a block diagram showing another example of the torque ripple elimination control computing unit 19. This is provided with an unbalanced load torque-corresponding gain calculator 26 in addition to the one shown in FIG. 2, and adjusts the amplitude in the torque ripple compensation value corresponding to the unbalanced load torque between the elevator car 1 and the counterweight 2. It is designed to be variable. The other configuration is the same as that of the torque ripple elimination control calculator 19 shown in FIG. 2, and therefore, the same elements are denoted by the same reference numerals and overlapping description will be omitted.

Referring to FIG. 5, a gain calculator 26 corresponding to the car load torque based on the car load WtTq from the load detector 5 attached to the car 1 has a gain constant a1 (Nf_Gain_WT) corresponding to the car load signal. Is output. The gain constant a1 is calculated by the multiplier 23 in the absolute value circuit 2.
Q-axis current command value (torque current command value) Iq obtained in step 2.
The value is multiplied by the absolute value of c, and becomes the amplitude A of the torque ripple compensation value α (Nf_Compen) indicated by the sine wave AsinQ.

This is because the unbalanced load applied to the rotor of the permanent magnet synchronous motor 3 differs depending on the loading condition of the car 1, and if the torque ripple compensation value α is constant, over-compensation occurs depending on the loading condition, and conversely vibration occurs. This is to prevent generation of components.

FIG. 6 is a block diagram showing another example of the torque ripple elimination control computing unit 19. In FIG. This is different from that shown in FIG. 2 in that an unbalanced load torque-corresponding phase calculator 27 and an adder 14d are additionally provided, and the torque ripple compensation value corresponding to the unbalanced load torque between the elevator car and the counterweight is provided. The phase is made variable. Other configurations are the same as those of the torque ripple elimination control calculator 19 shown in FIG.
The same elements will be denoted by the same reference symbols, without redundant description.

In FIG. 6, the unbalanced load torque corresponding phase calculator 27 includes a load detector 5 attached to the car 1.
And outputs a phase gain (_Nf_θe_WT) corresponding to the car load based on the car load WtTq. This phase gain is added by the adder 14d to the phase (_Nf_θe) output from the N-times period calculator 20, and becomes the phase Q of the torque ripple compensation value α (Nf_Compen) indicated by the sine wave AsinQ.

Since the unbalanced load applied to the rotor of the permanent magnet synchronous motor 3 differs depending on the loading condition of the car 1, if the torque ripple compensation value α is constant, over-compensation may occur depending on the loading condition, and conversely vibration may occur. This is to prevent generation of components.

FIG. 7 is a block diagram showing still another example of the torque ripple elimination control computing unit 19. This is different from the one shown in FIG. 2 in that a speed-dependent gain calculator 28 is provided so that the amplitude A of the torque ripple compensation value α can be varied according to the actual speed of the permanent magnet synchronous motor 3. The other configuration is the same as that of the torque ripple elimination control calculator 19 shown in FIG. 2, and therefore, the same elements are denoted by the same reference numerals and overlapping description will be omitted.

In FIG. 7, the q-axis current command Iqc output from the speed control system 6 is input to the absolute value circuit 22 of the torque ripple elimination control calculator 19, converted into an absolute value, and the obtained q-axis current command value The absolute value of Iqc is calculated by a multiplier 23.
Gain constant a (_Nf_Gain_FIX) that sets the amplitude of the sine wave
Is multiplied to obtain the amplitude value A1. This amplitude value A1
Is input to the speed-based gain calculator 28 and multiplied by a gain corresponding to the actual speed of the permanent magnet synchronous motor 3 to obtain the amplitude value A.
Is output.

FIG. 8 is a gain characteristic diagram of the speed-based gain calculator 28. Assuming that the permanent magnet synchronous motor 3 starts rotating at time t1, the gain of the speed-based gain calculator 28 increases from "0" at a predetermined change rate (rate) and becomes constant at "1". This is for the purpose of smoothly starting up until the set time t2 has passed so as not to cause torque disturbance in the control system.

If the speed V of the permanent magnet synchronous motor 3 exceeds the set speed V1 at time t3, the set speed V
The gain is decreased from "1" to "0" between 1 and the set speed V2. That is, between the set speed V1 and the set speed V2, the gain is set to “1” at a predetermined change rate.
From "0" to "0", and when the speed exceeds the set speed V2, the gain is maintained at "0". FIG. 8 shows a case where the gain is maintained at “0” from time t4 to time t5.

On the contrary, when the speed V of the permanent magnet synchronous motor 3 is reduced from the set speed V2 to the set speed V1 from time 5 to time t6, the gain is set to "0" at a predetermined change rate. From "1" to "1", and when the speed falls below the set speed V1, the gain is held at "1".

This is because when the elevator is traveling at the rated speed, the frequency becomes high, so that the response of the current control system 8 exceeds the limit and a phase shift occurs with respect to the ripple, and the effect of the vibration suppression control is reduced. This is to prevent the phenomenon of disappearing or conversely causing vibration.

As described above, the gain of the speed-dependent gain calculator 28 is made a variable gain, and the gain is reduced as the frequency becomes higher (high-speed region). At a certain frequency or higher, the gain is set to "0" to cancel the torque compensation. . It is also possible to use a variable gain of a predetermined function so as to enhance the effect of the vibration suppression control only in a specific frequency range.

FIG. 9 is a block diagram showing still another example of the torque ripple elimination control computing unit 19. 2, a corresponding address selector 29 and a corresponding gain / phase data table 30 are provided, and the amplitude and phase at a predetermined torque ripple compensation value are stored in the corresponding gain / phase data by the corresponding address selector 29. The amplitude and the phase in the torque ripple compensation value are selected from the table 30 so as to be variable. The other configuration is the same as that of the torque ripple elimination control calculator 19 shown in FIG. 2, and therefore, the same elements are denoted by the same reference numerals and overlapping description will be omitted.

In FIG. 9, a corresponding gain / phase data table 30 is an adjustment data table divided into a predetermined number, and stores correction values for determining the amplitude and phase in the torque ripple compensation value in advance according to the car load. Have been. Then, an address index output from the corresponding address selector 29 is input as a table address, and a phase (_Nf_Offset) and a gain (_Nf_Gain) corresponding to the unbalanced load torque are selected. Selected phase (_N
f_Offset) is added to the output of the N-times period calculator 20 by the adder 14e, and the gain (_Nf_Gain) is multiplied by the torque current command value (q-axis current command value) Iqc from the absolute value circuit 22 by the multiplier 23. Is done. As described above, by setting the values in the table based on the respective car loads, it becomes possible to supply the optimum compensation value.

Next, FIG. 10 is a block diagram showing still another example of the torque ripple elimination control calculator 19. This is different from that shown in FIG. 9 in that a corresponding gain / phase data function generator 31 is provided instead of the corresponding address selector 29 and the corresponding gain / phase data table 30,
The corresponding gain / phase data function generator 31 generates an amplitude and phase correction value in the torque ripple compensation value based on the car load, and makes the amplitude and phase in the torque ripple compensation value variable. The other configuration is the same as that of the torque ripple elimination control calculator 19 shown in FIG. 2, and therefore, the same elements are denoted by the same reference numerals and overlapping description will be omitted.

In FIG. 10, the corresponding gain / phase data function generator 31 is set so as to cancel the torque ripple.
Is a mathematical expression of an optimal compensation calculation function for unbalanced load torque.

[0069]

As described above, according to the present invention, even if the torque generated by the permanent magnet synchronous motor pulsates due to the effects of slot ripples, winding imbalances, and errors in the sensor system, the ride comfort of the elevator can be improved. Deterioration can be suppressed.

[Brief description of the drawings]

FIG. 1 is a block configuration diagram of an elevator control device according to a first embodiment of the present invention.

FIG. 2 is a block diagram showing a torque ripple elimination control arithmetic unit (part 1) according to the embodiment of the present invention;

FIG. 3 is a block configuration diagram of an elevator control device according to a second embodiment of the present invention.

FIG. 4 is a block diagram of an elevator control device according to a third embodiment of the present invention.

FIG. 5 is a block diagram showing a torque ripple elimination control calculator (part 2) according to the embodiment of the present invention.

FIG. 6 is a block diagram showing a torque ripple elimination control calculator (part 3) according to the embodiment of the present invention.

FIG. 7 is a block diagram showing a torque ripple elimination control calculator (part 4) according to the embodiment of the present invention.

FIG. 8 is a gain characteristic diagram of the speed-based gain calculator in FIG. 7;

FIG. 9 is a block diagram showing a torque ripple elimination control calculator (part 5) according to the embodiment of the present invention.

FIG. 10 is a block diagram showing a torque ripple elimination control calculator (part 6) according to the embodiment of the present invention.

FIG. 11 is a configuration diagram of an elevator control device that controls an elevator by driving a permanent magnet synchronous motor with an inverter.

FIG. 12 is an explanatory diagram showing a relationship between an armature winding formed in a slot of an iron core of the permanent magnet synchronous motor and a magnetic field generated in the armature winding as the magnetic pole moves.

FIG. 13 is a characteristic diagram showing the relationship between the actual magnetic flux and the magnetic pole position of the permanent magnet synchronous motor over time.

FIG. 14 is an explanatory diagram showing a relationship between a fundamental component and a harmonic component of a magnetic flux distribution of the permanent magnet synchronous motor.

[Explanation of symbols]

Reference Signs List 1 basket 2 counter weight 3 permanent magnet synchronous motor 4 pulse generator 5 load detector 6
Speed control system 7 Inverter 8 Current control system 9 Current sensor 10 Iron core 11 Armature winding 12
3 phase / 2 phase converter 13 dq converter 14 adder 15 PI controller 16 inverse dq converter 17 2 phase / 3 phase converter
18 PWM circuit 19 Torque ripple elimination control calculator 20 N-times period calculator 21 Sine wave function unit 22
Absolute value circuit 23 Multiplier 24 Limiter circuit 25
Filter circuit 26 Gain calculator corresponding to unbalanced load torque 27 Phase calculator corresponding to unbalanced load torque 28 Gain calculator corresponding to speed 29 Corresponding address selector 30
Corresponding gain / phase data table 31 Corresponding gain / phase data function generator

Claims (8)

[Claims] 1. An elevator control apparatus for controlling an elevator by supplying a variable voltage frequency AC power supply from an inverter to a permanent magnet synchronous motor and driving a hoisting machine connected to the permanent magnet synchronous motor to control the elevator. A speed control system that calculates a torque current command value and a magnetic flux current command value based on a speed deviation signal between the actual speed and the speed command value and calculates a primary frequency angle of the inverter based on the actual speed; A current control system that outputs a gate signal to the inverter so that the output current of the inverter satisfies the torque current command value and the magnetic flux current command value at the primary frequency angle, and based on the torque current command value and the primary frequency angle A torque ripple compensation value for suppressing torque ripple of the permanent magnet synchronous motor is calculated and the torque current command value is calculated. An elevator control device comprising: a torque ripple elimination control computing unit that takes into account the following. 2. The elevator control apparatus according to claim 1, wherein the torque ripple elimination control calculator adds the torque ripple compensation value to the magnetic flux current command value instead of the torque current command value. 3. The torque ripple elimination control computing unit adds the torque ripple compensation value to the primary frequency angle instead of the torque current command value.
3. The elevator control device according to claim 1. 4. The torque ripple elimination control computing unit computes the sine wave having a value proportional to the torque current command value as an amplitude and a phase of an N-times component of a fundamental frequency calculated from the primary frequency angle as the torque ripple compensation. The elevator control device according to any one of claims 1 to 3, wherein the operation is performed as a value. 5. The torque ripple elimination control calculator according to claim 4, wherein the amplitude of the torque ripple compensation value is made variable in accordance with the unbalanced load torque between the elevator car and the counterweight. Elevator control device. 6. The torque ripple elimination control computing unit according to claim 4, wherein a phase in the torque ripple compensation value is made variable in accordance with an unbalanced load torque between the elevator car and a counterweight. Elevator control device. 7. The torque ripple elimination control arithmetic unit varies an amplitude of the torque ripple compensation value according to an actual speed of the permanent magnet synchronous motor.
3. The elevator control device according to claim 1. 8. The torque ripple elimination control computing unit selects an amplitude and a phase at a predetermined torque ripple compensation value corresponding to an unbalanced load torque between an elevator car and a counterweight, and calculates the amplitude and phase at the torque ripple compensation value. The elevator control device according to claim 4, wherein the amplitude and the phase are variable.