JP2000135983A - Electric motor control device for motor-driven power steering - Google Patents

Abstract

PROBLEM TO BE SOLVED: To exercise sufficient performance as a motor-driven power steering without using a complicated, high-cost analog circuit. SOLUTION: An actual motor current detected by a motor current sensor 27 is converted into digital data by an analog-to-digital converter 34 and the data is taken into a CPU 35, and the CPU 35 computes the command value to a motor driving circuit 36 on the basis of the actual motor current and the target current value. The motor current can therefore be controlled without using a complicated, high-cost analog circuit. In the case of realizing motor current control by feedback control, if a means for computing the driving command value is constituted to execute filter processing for eliminating higher frequency components than cutoff frequency (=R/L) when approximating the current response of the electric motor to applied voltage with the time lag of first order, the influence of zero-order hold is reduced to realize a nonvibrating current control system.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device for an electric motor used for an electric power steering of a vehicle, which contributes to reducing a steering force of a driver and improving a steering feeling.

[0002]

2. Description of the Related Art In an electric power steering, a rotation direction and a rotation torque of an electric motor are controlled in accordance with an assist signal calculated based on an output signal of a torsion torque sensor for detecting a torsion torque of a steering system. This reduces the driver's steering load. For example, Tokuho 6
In the electric power steering disclosed in JP-A--24942, in order to generate a necessary rotational torque, control is performed to match the actual motor current to a predetermined current value, thereby configuring an electric power steering having excellent responsiveness.

[0003]

However, feedback control by an analog circuit using an operational amplifier or the like is used as a device for controlling the actual motor current to be equal to a predetermined current value, and the circuit is complicated. Had become expensive.

[0004] In view of the above circumstances, an object of the present invention is to provide an electric motor control device capable of exhibiting sufficient performance as an electric power steering without using a complicated and expensive analog circuit.

[0005]

According to a first aspect of the present invention, there is provided a control apparatus for an electric motor for an electric power steering, comprising: a steering torque detecting means for detecting a steering torque of a steering mechanism detected by a steering torque detecting means; The command value calculating means calculates the drive command value based on the calculated drive command value, and the motor drive means energizes and drives the electric motor for assisting the steering mechanism based on the calculated drive command value.

Here, the motor current detecting means detects a current flowing through the electric motor, and the detected value is converted into digital data which can be processed by the microcomputer by the AD converting means. On the other hand, the target motor current calculation means calculates a target motor current for the electric motor as digital data based on the steering torque detected by the steering torque detection means. Then, the calculating means converts the drive command value for the motor driving means with the current detection value converted into digital data by the AD conversion means and the target value so that the current flowing through the electric motor (actual motor current) matches the target motor current. Based on the motor current value, the microcomputer calculates at predetermined time intervals.

Therefore, according to the control device for an electric motor, a complicated and expensive analog circuit is not used.
The electric motor can be controlled. The elimination of the use of the analog circuit leads to the following effects. In other words, when configuring an analog circuit, for example, parts that have individual variations such as capacitors are required, and the performance of the device as a finished product may also vary. It is troublesome. According to the configuration of the control device of the present invention, such variation does not occur, and the adjustment work is not required. When an analog circuit is used, it is necessary to set and manufacture an analog circuit corresponding to the type of the object on which the electric power steering is mounted. For example, in an electric motor for electric power steering for a light vehicle and a large vehicle, it is necessary to cope with a difference in electric current and the like to be used. According to the configuration of the control device of the present invention, it is only necessary to change the program according to the type of the mounting target, and the hardware configuration can be shared.

In controlling the electric motor, the calculating means calculates a deviation between the target motor current and the actual motor current, based on the calculated current deviation. It is conceivable to calculate a feedback drive command value for. Then, the motor driving means energizes and drives the electric motor based on the calculated feedback drive command value. If you do this,
Highly accurate motor control can be realized. In particular, since it is premised on use in electric power steering, it is considered preferable to perform feedback control in consideration of environmental changes such as heat and aging. Also, for mass production,
Since it is practically difficult to perform a precise adjustment work before shipment, it can be said that it is preferable to respond by feedback control, particularly from the viewpoint of electric power steering mounted on a vehicle premised on mass production.

In order to control the actual motor current to a predetermined current value by a microcomputer,
Considering the amount of calculation load, the control cycle (sampling time) is desirably about 1 ms or more. In contrast, standard digital controller design techniques are:
This is a method in which a transfer function of an analog feedback controller having desired performance is bilinearly converted to derive a digital control law, and the digital control law is calculated by a microcomputer.
In order to perform control to match the actual motor current to a predetermined current value by this method, the control cycle is 200 microseconds (μ
s) It has been ascertained by the applicant that it needs to be of the order or less. If the control is performed in a control cycle that is slower than this, there is a problem that the actual motor current becomes oscillating and the steering feeling deteriorates. This is for the following reason.

In the digital control system, a command value for the motor drive circuit takes a constant value between each sampling. So-called zero order hold (zero order hold)
It is. Normally, the actual motor current becomes oscillating as the control cycle becomes slower because of the time constant (= inductance (L) /) of the electric motor used in the electric power steering.
This is because the effect of the zero-order hold cannot be ignored when the control cycle approaches 1 ms while the resistance value (R) is about several ms.

Therefore, in the control device according to the third aspect, the following configuration is employed in order to reduce the influence of zero order hold and to realize a non-oscillating current control system. That is, the calculating means for calculating the feedback drive command value described above approximates the response of the current to the applied voltage of the electric motor as a first-order lag, which is a cutoff frequency (= resistance value (R) / inductance / (L)). The filter processing for removing higher frequency components is performed. As described above, by removing the frequency components higher than the cutoff frequency, the control band of the feedback controller can be narrowed to a lower frequency side than the band of the analog feedback controller. As a result, the effect of zero order hold can be reduced, and a non-oscillating current control system can be realized.

With this configuration, the response of the actual motor current is generally slower than that of the analog feedback controller. However, this effect does not cause a problem in the motor current control of the electric power steering for the following reason. is there. That is, the control band of the electric motor control of the electric power steering which is the object of the present invention is different from the servo system of the machine tool or the like, and if it is about 100 Hz, it is sufficient for practical use. Applicants' knowledge that the cut-off frequency of a motor is several hundred Hz, and that even if high-frequency bands higher than the cut-off frequency of the motor are removed, a practically sufficient control band for electric motor control can be obtained. Based.

[0013] As described above, when the arithmetic means for executing the filter processing for removing the high frequency component is configured, the following can be performed. That is, as described in claim 4, the feedback drive command value is calculated using a transfer function having an element for removing a frequency component higher than the cutoff frequency. In this case, a transfer function having a delay element (a first-order delay element, a second-order delay element, or the like) may be realized. This may be realized by using a transfer function having an element for integrating the deviation of the actual motor current.

If the calculation means is constituted by using a transfer function having an integral element, a steady-state deviation when the target motor current changes in a step-like manner can be eliminated. When the actual motor current does not reach the target motor current even if the calculated feedback drive command value is maximum, it is preferable to stop the integration operation in the direction of increasing the deviation integration. This is for the following reason.

For example, let us consider a case where sudden steering is performed in electric power steering. If the deviation integration operation is not stopped, the following inconvenience occurs. That is, when a DC motor is used as the electric motor,
Back electromotive force is generated in proportion to the motor speed. And
In the case of sudden steering, the back electromotive force becomes a large value. For example, when a drive command value is output by PWM,
Even when the PWM duty is 100%, there is still a state where the actual motor current is lower than the target motor current (FIG. 1).
4 between A and B). The PWM duty value is given as a linear combination of the integral value of the deviation and the deviation.
It is limited to 100% between D and E. In such a case, if the function of integrating the deviation between the target motor current and the actual motor current is operating, the integrated value of the deviation is represented by D in FIG.
→ It follows the route of E. At point B in FIG. 14, the influence of the back electromotive force is reduced, and even if the actual motor current catches up with the target motor current, the deviation integration follows the path from E to F in FIG. Remains. As a result, the overshoot of the actual motor current occurs between B and C in FIG. 14, and the steering feeling deteriorates.

On the other hand, when the deviation integration operation is stopped as set forth in claim 7, when the PWM duty is 10
At a point D in FIG. 14 at which 0% has been reached, the integration operation in the direction of increasing the deviation integration stops, so that the deviation integration value follows the path of DG in FIG. Therefore, at the point B in FIG. 14, when the effect of the back electromotive force is reduced and the actual motor current catches up with the target motor current, the deviation integration follows the G → H path in FIG. Began to decrease,
Overshoot of the actual motor current does not occur, and a good steering feeling is obtained.

Further, as shown in claim 8, claims 2 to
7. The control device for an electric motor for electric power steering according to claim 7, wherein the calculating means calculates a feedforward drive command value for the motor drive means based on the target motor current, and adds the feedforward drive command value to the feedback drive command value. The final drive command value for the motor drive means may be used. By doing so, it is possible to reduce the steady-state deviation of the motor current and improve the response of the current rise, thereby further improving the steering feeling.

[0018]

Embodiments of the present invention will be described below with reference to the drawings. It is needless to say that the embodiments of the present invention are not limited to the following examples, and can take various forms as long as they belong to the technical scope of the present invention.

[First Embodiment] FIG. 1 is a vehicle configuration diagram mainly showing an electric power steering control device of an embodiment to which the above-mentioned invention is applied. A steering wheel (steering wheel) 11 steered by the vehicle driver is connected to a shaft (steering shaft) 12.
A rack 13 corresponding to a “steering mechanism” is provided on the shaft 12.
And the pinion gear 14 are connected. Handle 11
When the shaft 12 rotates in response to the steering, the rotation angle of the shaft 12 becomes the moving amount of the rack 13. Tie rods 15 are provided at both ends of the rack 13, and the tie rods 15 steer the evening ear 16 left and right.

The shaft 12 is provided with a steering torque sensor 21 and a steering angle sensor 24 corresponding to "steering torque detecting means". From the steering torque sensor 21, a torque generated when a vehicle driver steers a steering wheel is provided. (Steering torque signal) corresponding to the steering angle sensor 2
4 outputs a steering angle generated when the steering wheel is steered.

Steering torque sensor 21 and steering angle sensor 2
An electric motor 23 is mounted between the rack 4 and the rack 13 and the pinion gear 14 via a speed reducer 22. The speed reducer 22 is a well-known speed reducer including a worm and a worm wheel. If the electric motor 23 is energized and driven, the force required to rotate the handle 11 is reduced.

The electric motor 23 is provided with a motor rotation angle sensor 25 and a motor current sensor 27.
ECU 30 for executing the power supply control to electric motor 23
Detects the steering torque signal from the steering torque sensor 21, the steering angle from the steering angle sensor 24, the motor rotation angle from the motor rotation angle sensor 25, the motor current from the motor current sensor 27, and the speed of the vehicle. Vehicle speed sensor 5
Based on the vehicle speed signal from 0, the current supplied to the electric motor 23 is controlled.

Next, the internal configuration of the ECU 30 will be described.
This will be described with reference to the processing block diagram of FIG. Since the calculation of the electric motor control is performed digitally based on the signal output from the steering torque sensor 21, inside the ECU 30,
First, the detection signal is passed through a low-pass filter (LPF) 31 composed of an analog circuit, and then the A / D converter 34 provided in the microcomputer 33 converts the A / D signal into an analog signal.
After the D conversion, it is taken into the CPU 35. Where LP
F31 has a Nyquist frequency (1/1 of the sampling frequency) to remove aliasing noise at the time of A / D conversion.
2) It is configured as a filter having the following cutoff frequency.

Similarly, the signal output from the motor current sensor 27 is also converted into digital data by an A / D converter 34 in the microcomputer 33 after passing through the LPF 32 and then taken into the CPU 35. On the other hand, the signal of the vehicle speed sensor 50 is a signal whose pulse frequency generally changes according to the vehicle speed. This signal is subjected to waveform shaping by a waveform shaping section 38 in the ECU 30 and input to a port of the CPU 35. The CPU 35 obtains a signal corresponding to the vehicle speed by measuring the cycle of the vehicle speed pulse. Similarly, signals from the motor rotation angle sensor 25 and the steering angle sensor 24 are
The signals are input to the ports of the CPU 35 via the waveform shaping units 39 and 40 provided correspondingly.

The microcomputer 33 includes a driving circuit 3
6, a PWM (pulse width modulation) duty is output. The drive circuit 36 controls the power transistor 3 based on the PWN signal so that the current of the electric motor 23 follows.
7 is performed. Next, a control calculation block of the electric power steering control device of the first embodiment (FIG. 3)
The operation will be described with reference to a control flowchart (FIG. 4) corresponding to the control operation block. The operation shown in the flowchart of FIG. 4 is executed in the microcomputer 33 at each predetermined control cycle.

In the first step 10, a detection signal (steering torque signal) from the steering torque sensor 21 is converted into digital data by the A / D converter 34, and the CPU 35
It is taken in. Subsequent steps 20 to 50 correspond to the target motor torque calculation in the control calculation block of FIG. First, in step 20, a basic assist torque is calculated based on the steering torque data. In the following step 30, the basic assist torque is corrected so that the assist torque decreases as the vehicle speed increases.

In step 40, the assist torque for improving the return of the steering wheel is corrected based on the motor rotation angle and the steering angle. Then, in step 50, in order to improve convergence when the steering wheel is returned, the assist torque is corrected based on the motor rotational angular velocity and the steering angular velocity.
The result obtained in this way is the target motor torque.

In the subsequent step 60, the above-mentioned steps 20 to st
The motor current corresponding to the target motor torque calculated in ep50 is calculated. Then, in step 70, the motor current signal is A / D converted and becomes digital data,
5 is taken in. In the subsequent step 80, a command value (PWM duty value) for the drive circuit is calculated based on the actual motor current value and the target motor current value. And the last
In step 90, the calculated PWM duty value is output to the drive circuit 36 to control the motor current.

As described above, the drive circuit 36 switches the power transistor 37 based on the PWM signal so that the current of the electric motor 23 follows. As described above, according to the electric power steering control device of the first embodiment, the actual motor current detected by the motor current sensor 27 is converted into digital data by the AD converter 34 and is taken into the CPU 35. A command value for the motor drive circuit 36 is calculated by the CPU 35 based on the motor current and the target current value. As a result, the motor current can be controlled without using a complicated and expensive analog circuit.

The elimination of the need to use an analog circuit leads to the following effects. In other words, in the case of configuring an analog circuit, for example, a component such as a capacitor or the like having a variation between individual components is required, so that the performance of the device as a finished product may also vary.
The adjustment work is troublesome. With the configuration as in the present embodiment, such variations do not occur, and adjustment work is not required. When an analog circuit is used, it is necessary to set and manufacture an analog circuit corresponding to the type of the object on which the electric power steering is mounted. For example, in the electric motor 23 for electric power steering for a light vehicle and a large vehicle, it is necessary to cope with a difference in electric current and the like to be used. With the configuration as in the present embodiment, it is only necessary to change the program according to the type of the mounting target, and the hardware configuration can be shared.

[Second Embodiment] In the second embodiment, a feedback drive command value is calculated as a drive command value for the electric motor 23. The operation will be described with reference to a control calculation block (FIG. 5) and a control flowchart (FIG. 6) corresponding to the control calculation block of the electric power steering control device of the second embodiment. The calculation shown in the flowchart of FIG. 6 is executed in the microcomputer 33 at every predetermined control cycle. The vehicle configuration (FIG. 1) mainly showing the electric power steering control device in the case of the first embodiment described above and E
A processing block diagram showing the internal configuration of the CU 30 (FIG. 2)
The same applies to the case of the second embodiment, and therefore, the description thereof will not be repeated.

Step 110 in the flowchart of FIG.
Steps 170 to 190 and step 190 have the same contents as steps 10 to 70 and step 90 shown in FIG. 4 in the case of the first embodiment, and a description thereof will be omitted, and only step 180 and step 185 will be described. In step 180, the difference (current deviation) between the target motor current and the actual motor current is calculated. And the following step
In 185, according to the recurrence formula shown in [Equation 1] below, P
The WM duty value d (n) is calculated.

[0033]

(Equation 1)

Here, d (k) is the PWM duty value in the k-th cycle, e (k) is the current deviation in the k-th cycle, and a
(K) and b (k) are predetermined constants. Thereafter, the PWM duty value d (n) obtained by the above calculation
Are subjected to upper and lower limiter processing to obtain a final PWM duty value. In step 190, the final PWM duty value is output to the drive circuit 36 to control the motor current.

As described above, in the case of the second embodiment, the feedback drive command value is calculated based on the current deviation.
Based on the feedback drive command value, the electric motor 2
3 is energized, so that highly accurate motor control can be realized. In particular, since it is premised on use in electric power steering, it is considered preferable to perform feedback control in consideration of environmental changes such as heat and aging.
In addition, when mass-producing, it is practically difficult to perform precise adjustment work before shipping.From the viewpoint of electric power steering mounted on vehicles that are premised on mass production, feedback control is particularly necessary. It can be said that the corresponding is preferable.

[Third Embodiment] By the way, in order for the microcomputer 33 to perform control for matching the actual motor current to a predetermined current value, the control cycle must be about 1 ms or more in view of the calculation load. Is desirable. On the other hand, a standard digital controller design method bilinearly converts the transfer function of an analog feedback controller having desired performance, derives a digital control law, and converts it into a microcomputer. This is a method for performing calculations. In order to perform control to match the actual motor current to a predetermined current value by this method, the control cycle is 200 microseconds (μ
s) It is necessary to be less than about. If the control is performed in a control cycle that is slower than this, there is a problem that the actual motor current becomes oscillating and the steering feeling deteriorates. This is because the time constant (= inductance (L) / resistance value (R)) of the electric motor used for the electric power steering is usually about several milliseconds, but becomes zero when the control cycle approaches 1 ms. This is because the influence of the order hold cannot be ignored.

The third embodiment is an embodiment for reducing the influence of zero order hold and realizing a non-oscillating current control system. [First Mode of Third Embodiment] The control operation block shown in FIG. 7 is a portion corresponding to the "motor current feedback loop" in the control operation block of FIG. 5 used for the description of the second embodiment. Is taken out. Cd (z) shown in FIG. 7 is a pulse transfer function corresponding to the above [Equation 1], and z represents a delay operator. ZOH is a zero-order hold, P (s) is a continuous-time transfer function representing the electric motor 23 and the drive circuit portion (motor drive circuit 36, power transistor 37), and s is a Laplace operator.

In the first embodiment, the motor current feedback controller in the control calculation block of FIG. 7 removes a frequency component higher than the cutoff frequency (= resistance value (R) / inductance (L)). It is characterized by having a filtering function. This is particularly effective when the control cycle (sampling time) is not negligible compared to the time constant (L / R) of the electric motor 23.

Regarding the structure of the motor current feedback controller, the structure of the pulse transfer function Cd (z) is as follows.
This will be specifically described with reference to an actual example. Assuming that a DC motor having an inductance (L) of 83.45 μH and a resistance (R) of 53.95 mΩ is assumed as the electric motor 23, and a motor current feedback control system as shown in FIG. . The control cycle is 1 ms
And The motor time constant in this case is L / R = 1.5 ms
Thus, the control cycle 1 ms has a size that cannot be ignored.

As described above, in the case where the control cycle cannot be ignored compared to the time constant of the control object, as a method for considering the influence of zero order hold, for example, a document “Introduction to Digital Control Theory” (Mitsuhiko Araki, Asakura Shoten, P137) ~ P13
There is a known technique described in 9). Hereinafter, the main points will be described.

In FIG. 8, Fd (z) is a pulse transfer function of the feedback controller, ZOH is a zero order hold, and Gc (s) is a continuous time transfer function of the control object. In the digital controller as shown in FIG. 8, when discussing stability in controlling a continuous-time control system,
First, considering Gd (z) shown in the following [Equation 2],

[0042]

(Equation 2)

Further, by introducing w shown in the following [Equation 3],

[0044]

(Equation 3)

The continuous time transfer function Gc (w) is expressed as the following [Equation 4].

[0046]

(Equation 4)

A control system as shown in FIG. 9 is configured for the control target represented by the continuous time transfer function Gc (w). The stability margin realized by the controller represented by the continuous time transfer function Fc (w) is obtained by using the continuous time transfer function Fd (z) defined as in the following [Equation 5]. It matches the stability margin when configured. This is the content of the known technology described in the above document.

[0048]

(Equation 5)

That is, in order to obtain a digital controller having a desired stability margin in consideration of the zero order hold for the control system including the zero order hold (ZOH) shown in FIG. A continuous-time transfer function Fc (w) so that the control system has a desired stability margin.
And then form the continuous-time transfer function Fd (z) according to the above [Equation 5].

However, how to configure an appropriate continuous time transfer function Fc (w) for a specific control object is not a known technique. In the first embodiment, the electric motor 23 for electric power steering is considered as a specific control object, and an appropriate continuous-time transfer function Fc (w) for the electric motor 23 is configured on the premise of such known technology. Things.

The DC motor taken as an example (L = 8
The transfer function P (s) from the equivalent applied voltage to the motor current of 3.45 μH, R = 53.95 mΩ) is as shown in the following [Equation 6]. P (s) = 1 / (L * s + R) [Equation 6] According to the above [Equation 2], [Equation 3], and [Equation 4], Gc shown in FIG. When the continuous-time transfer function Pc (w) corresponding to (w) is obtained, it becomes as shown in the following [Equation 7].

[0052]

(Equation 6)

A Bode diagram of the continuous time transfer function Pc (w) is shown in FIG. From FIG. 10, the gain crossover frequency for Pc (w) cannot be strictly defined in consideration of the zero-order hold of P (s), which was the first-order lag. It can be seen that the phase is delayed by 180 degrees and the system has a small phase margin.

In the first embodiment, a controller corresponding to Fc (w) in FIG.
The transfer functions shown in [Equation 9] and [Equation 10] are used.

[0055]

(Equation 7)

The Bode diagram is shown in FIG. Among them, the transfer function Fc1 (w) uses a first-order lag as an element by which the motor current feedback controller removes frequency components higher than the cutoff frequency (R / L). The loop transfer function Fc shown by (c) in FIG.
As can be seen from the Bode diagram of (w) * Pc (w), Fc
With the effect of 1 (w), the loop transfer function Fc (w) * Pc
It can be seen that the cutoff frequency of (w) moves to a lower frequency range and the stability margin is increased. Fc2 (w) is Fc1
This has the effect of reducing the phase delay due to (w).

Here, according to the above [Equation 5], FIG.
When the pulse transfer function Cd (z) of the motor current feedback controller at is obtained, the following equation (11) is obtained.

[0058]

(Equation 8)

Using this pulse transfer function Cd (z),
The above-described DC motor (L = 83.4 at a control cycle of 1 ms)
As an example of performing current feedback control of 5 μH, R = 53.95 mΩ), a response waveform when the target current is changed from OA to 10 A in a step-like manner is shown in FIG.
Indicated by (B) in FIG. 11 is a response waveform when simple proportional control is performed without an element for removing high frequency components. It can be seen that the response in (A) in FIG. 11 is non-oscillating and the steady-state deviation is smaller than that in (B).

[Second Embodiment of Third Embodiment] Although the delay element is used in the first embodiment, an integral element is used in the second embodiment. That is, the continuous time transfer function Fc shown in FIG.
Instead of (w), the one shown in the following [Equation 12] is used.

[0061]

(Equation 9)

A Bode diagram in that case is shown in FIG.
Indicated by The continuous-time transfer function F represented by this [Equation 12]
For c (w), an integral element is used as an element by which the motor current feedback controller removes frequency components higher than the cutoff frequency (L / R). As can be seen from the Bode diagram of the loop transfer function Fc (w) * Pc (w) shown in FIG.
With the effect of (w), the loop transfer function Fc (w) * Pc (w)
It can be seen that the cutoff frequency moves to a lower range, and the stability margin increases.

Further, the continuous time transfer function Fd (z) defined by the above [Equation 5] is obtained by using the continuous time transfer function Fc (w) shown by the [Equation 12]. When the pulse transfer function Cd (z) of the motor current feedback controller is obtained, it becomes as shown in the following [Equation 13].

[0064]

(Equation 10)

Using this pulse transfer function Cd (z),
The above-described DC motor (L = 83.4 at a control cycle of 1 ms)
As an example of performing the current feedback control of 5 μH, R = 53.95 mΩ), the response waveform when the target current is changed from OA to 10 A in a step shape is shown in FIG.
Indicated by

In this case as well, it can be seen that the response is non-oscillatory, as in the case of using the delay element shown in FIG. In this case, it is also found that the steady-state deviation is zero due to the effect of the integral element. In each of the first embodiment using the delay element described above (see the transfer function shown in Equation 8) and the second embodiment using the integral element (see the transfer function shown in Equation 12), the non-synchronous sampling time and the non- In order to realize an oscillating current response, it is essential to have an element for removing a frequency component higher than the cutoff frequency of the electric motor 23.

[Contrivance in Second Embodiment] When the integral element is used as in the second embodiment, considering the following points, even if the calculated feedback drive command value is the maximum, When the actual motor current does not reach the target motor current, it is preferable to stop the integration operation in the direction to increase the deviation integration. This is for the following reason.

If the deviation integration operation is not stopped in a situation where the electric power steering is suddenly steered,
When the following inconvenience occurs, that is, when a back electromotive force proportional to the motor rotation speed of the electric motor 23 using the DC motor is generated and the vehicle is steered suddenly, the back electromotive force has a large value. Therefore, there is a state where the actual motor current is lower than the target motor current even when the PWM duty is 100% (see a section between AB in FIG. 14). The PWM duty value is given as a linear combination of the integral value of the deviation and the deviation, but is limited to 100% between D and E in FIG. In such a case, if the function of integrating the difference between the target motor current and the actual motor current is operating, the integrated value of the difference follows the path from D to E in FIG. FIG.
At point B in FIG. 4, the influence of the back electromotive force is reduced, and even if the actual motor current catches up with the target motor current, the deviation integration follows the path from E to F in FIG. Remains. As a result, BC in FIG.
Since the overshoot of the actual motor current occurs between the two, the steering feeling deteriorates.

Therefore, the control as shown in the flowchart of FIG. 15 is executed, and if necessary, the deviation integration operation is stopped. Prior to the description of the flowchart of FIG. 15, a description related to the processing will be given. The pulse transfer function Cd of the motor current feedback controller shown in the above [Equation 13]
(Z) can be modified as shown in the following [Equation 14].

[0070]

[Equation 11]

That is, the part [0.
05] and the part corresponding to the deviation integration [0.05 / (1-
Z ^-1 )]. The calculation for each of the deviation proportional part and the deviation integral part is performed in FIG. Step 110 in the flowchart of FIG.
Steps 180 to 190 and step 190 are the same as those shown in FIG. 6 in the second embodiment, except that steps 181 to tep 184 are executed instead of step 185 in FIG. Therefore, that part will be mainly described.

After calculating the difference (current deviation) between the target motor current and the actual motor current in step 180, in subsequent step 181 it is checked (saturation check) whether or not the PWM duty value of the previous cycle is the maximum value. . According to the check result, the PWM duty value d in step 182
The deviation integration calculation of (n) changes.

That is, in step 182, the PW
When the M duty value is the maximum value, the integral calculation in the direction in which the integral value increases (the part corresponding to the deviation integral [0.05 / (1-Z ^-1 ) shown in the above [Equation 14]) is Not performed. It should be noted that the integral calculation is performed in the direction in which the integral value decreases. On the other hand, when the PWM duty value of the previous cycle is not the maximum value, the integration operation (the part corresponding to the deviation integration in [Equation 14]) is performed as it is.

In the following step 183, the calculation of the deviation proportional part [0.05] shown in [Equation 14] is executed. Then, in step 184, the results of step 182 and step 183 are added, necessary guard processing is performed, and the final PW
Let it be M duty value.

By executing such processing, the PWM
At the point D in FIG. 14 where the duty reaches 100%, the integration operation in the direction of increasing the deviation integration is stopped. Therefore, the deviation integral value follows the path from D to G in FIG. Therefore, at the point B in FIG. 14, when the influence of the back electromotive force decreases and the actual motor current catches up with the target motor current, the deviation integration follows the G → H path in FIG.
The PWM duty starts to decrease immediately, no overshoot of the actual motor current occurs, and a good steering feeling is obtained.

[Fourth Embodiment] In the fourth embodiment, a command value for the motor drive circuit 36 is fed-forward term based on the target current in parallel with the motor current feedback controller in the second or third embodiment. And a motor current feed-forward controller is provided. The operation will be described with reference to a control calculation block (FIG. 16) and a control flowchart (FIG. 17) corresponding to the control calculation block of the electric power steering control device of the fourth embodiment. The operation shown in the flowchart of FIG. 15 is executed in the microcomputer 33 at each predetermined control cycle. In addition, the first
The vehicle configuration (FIG. 1) mainly showing the electric power steering control device in the case of the embodiment and the processing block diagram (FIG. 2) showing the internal configuration of the ECU 30 are the same in the case of the second embodiment. Do not.

Step 21 in the flowchart of FIG.
0 to 270 and step 290 correspond to FIG. 4 in the case of the first embodiment.
Since the contents are the same as those of steps 10 to 70 and step 90 shown in FIG. In step 280, the difference (current deviation) between the target motor current and the actual motor current is calculated, similarly to step 180 in the case of the above-described second embodiment. And continue
In step 285, the PWM duty feedback value dfb (n) is calculated according to the recurrence formula shown in [Equation 15] below.

[0078]

(Equation 12)

Here, dfb (k) is the PWM of the k-th cycle.
The duty feedback value, e (k), is the current deviation in the k-th cycle, and a (k), b (k) are predetermined constants. In the following step 286, the PWM duty feedforward value dff (n) is calculated according to the recurrence formula shown in [Equation 16] below.

Dff (n) = f (R * Iref (n)) [Expression 16] where R is the resistance value of the DC motor, lref (n) is the target motor current in the n-th cycle, and f (...) ) Is a preset map. Then, in step 287, the PWM duty final value d is calculated as shown in the following [Equation 17].
(N) is calculated.

D (n) = dff (n) + dfb (n) [Expression 17] Then, the PWM duty value d obtained by the above calculation
(N) is subjected to upper and lower limiter processing, and the final PWM
Duty value. In step 290, the final P
The WM duty value is output to the drive circuit 36 to control the motor current.

As described above, in the case of the fourth embodiment, P
WM duty feedback value dfb (n) and PWM
Since the PWM duty final value d (n) is calculated based on the duty feed forward value dff (n) and the motor current is controlled based on the calculated value, the steady-state deviation of the motor current is reduced, The response can be improved, and the steering feeling can be further improved.

[Others] (1) In the above-described embodiment, the electric motor 23 is
The configuration is premised on a configuration in which the shaft is rotated via the shaft 2 to perform the auxiliary steering. However, in addition to this, the same applies to a configuration based on a “rack coaxial type” in which the rack 13 is directly driven by electromagnetic force or the like. Applicable.

(2) In the first embodiment of the third embodiment described above, the transfer function having the first-order lag element is used. However, the transfer function is not limited to the first-order lag element but has a second-order or higher-order lag element. The same can be realized by using a transfer function.

[Brief description of the drawings]

FIG. 1 is a vehicle configuration diagram mainly showing an electric power steering control device according to an embodiment to which the present invention is applied.

FIG. 2 is a processing block diagram illustrating an internal configuration of an ECU according to the embodiment.

FIG. 3 is a control calculation block diagram of the electric power steering control device of the first embodiment.

FIG. 4 is a control flowchart corresponding to a control calculation block of the first embodiment.

FIG. 5 is a control arithmetic block diagram of an electric power steering control device according to a second embodiment.

FIG. 6 is a control flowchart corresponding to the control calculation block of the second embodiment.

FIG. 7 shows a portion corresponding to a “motor current feedback loop” in the control calculation block of FIG. 5;

FIG. 8 is a control calculation block diagram for illustrating a first mode of the third embodiment.

FIG. 9 is a control calculation block diagram for illustrating a first mode of the third embodiment.

FIG. 10 is a Bode diagram of a transfer function in a first mode of the third embodiment.

FIG. 11 is an explanatory diagram of a response waveform when the target current is changed in a step-like manner when the current F / B control of the DC motor is performed using the pulse transfer function in the first mode of the third embodiment. is there.

FIG. 12 is a Bode diagram of a transfer function in a second mode of the third embodiment.

FIG. 13 is an explanatory diagram of a response waveform when the target current is changed in a stepwise manner when the current F / B control of the DC motor is performed using the pulse transfer function in the second mode of the third embodiment. is there.

FIG. 14 is an explanatory diagram showing a comparison between a case where the deviation integration operation is not stopped and a case where the deviation integration operation is stopped in a situation where the electric power steering is suddenly steered, in the second mode of the third embodiment.

FIG. 15 is a flowchart showing control when the deviation integration operation is stopped in the second mode of the third embodiment.

FIG. 16 is a control calculation block diagram of the electric power steering control device of the fourth embodiment.

FIG. 17 is a control flowchart corresponding to the control calculation block of the fourth embodiment.

[Explanation of symbols]

DESCRIPTION OF SYMBOLS 11 ... Handle 12 ... Shaft 13 ... Rack 14 ... Pinion gear 15 ... Tie rod 16 ... Evening ear 21 ... Steering torque sensor 22 ... Reduction gear 23 ... Electric motor 24 ... Steering angle sensor 25 ... Motor rotation angle sensor 27 ... Motor current sensor 30 ... Microcomputers 31, 32 LPF 34 A / D converter 35 CPU 36 Motor drive circuit 37 Power transistors 38, 39, 40 Waveform shaping unit 50 Vehicle speed sensor

 ────────────────────────────────────────────────── ─── Continuing on the front page (72) Inventor Takehito Fujii 1-1-1 Showa-cho, Kariya-shi, Aichi F-term in DENSO Corporation (Reference) 3D032 CC08 CC12 CC48 DA03 DA15 DA23 DA63 DA64 DC01 DC02 DC09 DC12 DD02 DD06 DD07 DD10 DD17 DD18 EA01 EB11 EC23 GG01 GG02 3D033 CA13 CA16 CA17 CA20 CA21 CA27

Claims (8)

[Claims] An electric motor for assisting steering of a steering mechanism; a motor driving means for energizing the electric motor; a steering torque detecting means for detecting a steering torque in the steering mechanism; and a detecting means for detecting the steering torque. And a command value calculating means for calculating a drive command value for the motor driving means based on the steering torque obtained, further comprising: detecting a current flowing in the electric motor. A target motor current calculating means for calculating a target motor current for the electric motor as a digital value based on a steering torque detected by the steering torque detecting means; Digital data that can be processed by a microcomputer with the value detected by the motor current detection means A / D converting means for converting, and a drive command value for the motor driving means is converted into digital data by the A / D converting means such that a current (actual motor current) flowing through the electric motor matches the target motor current. A control device for an electric motor for electric power steering, comprising: a calculation unit that performs calculation by a microcomputer at predetermined time intervals based on a current detection value and the target motor current value. 2. The control device for an electric motor for electric power steering according to claim 1, wherein said calculating means calculates a deviation between said target motor current and an actual motor current, and based on the calculated current deviation, A control device for an electric motor for electric power steering, wherein a feedback drive command value for said motor driving means is calculated. 3. The control device for an electric motor for electric power steering according to claim 2, wherein said calculating means is higher than a cutoff frequency when a response of a current to an applied voltage of said electric motor is approximated as a first-order lag. A control device for an electric motor for electric power steering, wherein a digital filter process for removing a frequency component in a frequency range is executed. 4. The control apparatus for an electric motor for electric power steering according to claim 3, wherein said calculating means uses a transfer function having an element for removing a frequency component higher than the cutoff frequency. A control device for an electric motor for electric power steering, wherein the digital filter processing is realized by calculating a command value. 5. The control device for an electric motor for electric power steering according to claim 4, wherein said calculating means uses a transfer function having a delay element as an element for removing a frequency component higher than the cutoff frequency. A control device for an electric motor for electric power steering. 6. The control device for an electric motor for electric power steering according to claim 4, wherein the calculating means removes the target motor current and the actual motor current as elements for removing a frequency component higher than the cutoff frequency. A control device for an electric motor for electric power steering, wherein a transfer function having an element for integrating a deviation of the electric power is used. 7. The control device for an electric motor for electric power steering according to claim 6, wherein the calculating means sets the actual motor current to the target motor current even if the calculated feedback drive command value is maximum. The control device for an electric motor for electric power steering, wherein the integration operation in the direction to increase the deviation integration is stopped when the deviation does not reach. 8. The control apparatus for an electric motor for electric power steering according to claim 2, wherein said calculating means calculates a feedforward drive command value for said motor driving means based on said target motor current. An electric motor control apparatus for an electric power steering, wherein the control signal is added to the feedback drive command value to obtain a final drive command value for the motor drive means.