GB2558296A - Radio frequency modulator - Google Patents

Abstract

A radio frequency modulator which requires at least two carrier signals V1a-V2b to be modulated by at least two waveforms and combined in a single summation. The modulator may alternatively be a mixer or demodulator and phase or amplitude modulation may be applied to the carriers V1a-V2b, which may be phase offset. Each phase offset may be 90 degrees out of phase relative to another carrier. There may be two, three or four carrier waves V1a-V2b in some embodiments. The transmitter may be quadrature amplitude modulated (QAB) or a single sideband (SSB) frequency transmitter. The output of the modulator may be filtered before transmission. Each carrier V1a-V2b may be paired within the summation with an auxiliary balanced switching pulse waveform arranged to facilitate suppression of third and fifth harmonics of each carrier wave V1a-V2b.

Description

(54) Title of the Invention: Radio frequency modulator

Abstract Title: Radio frequency modulator with a single direct summation (57) A radio frequency modulator which requires at least two carrier signals V1a-V2b to be modulated by at least two waveforms and combined in a single summation. The modulator may alternatively be a mixer or demodulator and phase or amplitude modulation may be applied to the carriers V1a-V2b, which may be phase offset. Each phase offset may be 90 degrees out of phase relative to another carrier. There may be two, three or four carrier waves V1a-V2b in some embodiments. The transmitter may be quadrature amplitude modulated (QAB) or a single sideband (SSB) frequency transmitter. The output of the modulator may be filtered before transmission. Each carrier V1a-V2b may be paired within the summation with an auxiliary balanced switching pulse waveform arranged to facilitate suppression of third and fifth harmonics of each carrier wave V1a-V2b.

V1ac

T1

T2_

V2a

V1bo

V2b

Output

Figure 5

1/5

	/ / V	c I
	- r

jfi k

2/5

a Main + Auxiliary

Composite

Composite b Main + Auxiliary

c Main + Auxiliary

Figure 2

Composite

3/5 a Main b Main

Main

+ Auxiliary = Composite

+ Auxiliary = Composite

+ Auxiliary Figure 3

Composite

4/5

V1ao

V1b

V2ao

V2b

T1 T3 •Η H · l· l·· p2 Ii s s |

T2 j ·

PX) 3 s S

Output

-ό V3a

V3b

T4

Figure 4

V4a

| . Q-
=3
CL	LL 0_	-Ω	_Q	-Ω	-Ω
=3	00		CD	LD
OQ		• 1—	• I—	• 1—	• 1—
	ΟΤΠΓΟ	mrn	Ο7ΩΓΟ	iTOTT

η in

o o

Figure 6

Q.

''Φ rnrmi

Q.

CO ρππππππ

Q.

CM

ΓΟΤΟΤΠ

Q.

ρπππππρ

Vs+ cr E E > >

Title: Radio Frequency Modulator

Description of Invention and the Prior Art

This invention relates to radio frequency modulating systems and to improved means and techniques for generating quadrature amplitude modulated (QAM) radio frequency transmissions and more particularly single sideband (SSB) radio frequency transmissions. It also relates to mixers and demodulators.

There is a need for power efficiency in transmitters generating radio frequency power. Various techniques are proposed and in use to this end with varying degrees of success and complexity. For efficiency they generally use non-linear techniques to minimize the losses. An example of this is an amplitude modulation (AM) transmitter using a square wave carrier (class D). This is particularly efficient when the modulation is supplied by means of a modulating switch mode supply. Another example of efficient AM is the Chireix Outphasing technique ¹¹¹ where AM is generated by the phase modulation (PM) and combination of a pair of constant amplitude carriers. This allows efficient non-linear stages to be used in the output ^pl. QAM improves bandwidth efficiency over AM by increasing the information carried and suppressing the carrier. SSB, a form of QAM, suppresses one sideband and is used extensively because of its bandwidth efficiency. Techniques for the generation of approximations to SSB at higher efficiency use techniques such as FM/AM balancing ^P1 and complex envelope reconstruction techniques such as Envelope Elimination and Restoration (EER)^[4]. These techniques are surveyed in recent papers ^[5·⁶·⁷¹.

A key component of QAM and SSB generation schemes in the prior art is a balanced modulator i^{S 9 |l,} i. which is a form of amplitude modulator. The principal output from an amplitude modulator producing AM is a carrier wave and upper and lower sidebands. It is acting as a mixing device (frequency translation) in that difference and sum frequencies are produced as well as both the original input signals. The output from a balanced modulator suppresses the input carrier wave and if doubly balanced then both carrier and modulation are suppressed. The result is double sideband (DSB) transmission with a suppressed carrier (DSBSC). Balanced modulators such as the Cowan modulator¹¹⁰¹ and derivatives are almost invariably low power devices and, when used for generating the signal for a transmitter, necessitate careful linear amplification strategies that are either inefficient or complex. An added level of complexity occurs in the generation of QAM and SSB or the related image rejection mixer where at least two balanced modulators are combined. A high power balanced modulator has been proposed ¹¹¹¹ for generation of DSBSC at high power. This technique does not facilitate further combination to generate SSBSC at high power.

It is an object of this invention to facilitate the production of multiple modulation modes such as QAM or SSB in a transmitter directly at high power and with high efficiency. To this end the invention describes a multiple input summation modulator. It is also an object of this invention that its use may be extended to providing the function of a mixer or frequency translator and a demodulator. A key difference between this invention and the prior art is the characteristic of taking multiple inputs and producing the required output directly in a single summation that can be implemented at high power.

The Figures referred to in the descriptions that follow are:

Figure 1: Phasor diagrams.

Figure 2: Schematic representation of pulse waveforms facilitating harmonic suppression.

Figure 3: Schematic representation of preferred pulse waveforms with phase relationships.

Figure 4: Schematic representation of the summation modulator based on the 4 PM method.

Figure 5: Schematic representation of the summation modulator based on the 2 PM method.

Figure 6: An outline design for a transmitter based on the 4 AM method.

The operating principal of the present invention will be shown mathematically in comparison with the prior art modulation techniques. Sinusoidal components are utilized in this analysis it being understood that Fourier analysis and superposition permits this. The radio frequency carrier is represented as radian frequency οχ and the modulation m is exemplified as sinusoidal modulation at radian frequency ω,„.

The process of Amplitude Modulation (AM) is exemplified as follows:

Carrier signal: Acos(co_ct)

Modulation signal with offset: 1+m == 1+Mcos(w,„t) where |m| < 1 and |M| < 1

AM signal: Acos(<O_ct)( 1+Mcos(w,„t))

This can be expanded to: A(cos(ort)+ ^l/2Mcos(or-w,„)t+ 'ZMcosrir+Midt)

i.e. the AM characteristic of a fixed carrier and amplitude modulated upper and lower sidebands with the modulation frequency offset from the carrier.

In typical high power AM schemes the amplitude information of the input carrier (A) is lost due to saturation and the modulation signal may be present at the output but is easily removed if the modulation frequency is well separated from the carrier frequency.

The typical output is then:

cos(co_ct)+ ¹/2Mcos(cDc-coin)t+ ¹/2Mcos(coc+co_in)t+1 +Mcos(ra_mt) which is usually filtered to the required AM signal: cos(co_ct)+ ¹/2Meos(m_c-ra_m)t+ 'ZMcosioy+oyJt

Outphasing propounded by Henri Chireix ^[1·²·⁶¹ is a technique used to generate AM by combining two constant amplitude PM carriers.

AM signal: (l+m)cos(co_ct) == 2cos((i)cos(oyt) == cosioyt-β) + cosioyt+f)

Where β = arccos((l+m)/2) for |m| < 1 and 0 < β < π/2

The phase modulation offset β is applied symmetrically to the carrier. The Chireix technique is used particularly in high power AM transmitters. To achieve linear modulation m must be transformed as shown to derive the phase angle β for the phase modulation. Figure la is a phasor example depicting the summation of 2 Outphasing PM carriers to create AM for m = 0.45. The PM carrier amplitudes are each 1 unit and the resultant (dashed line) is an in-phase carrier with instantaneous amplitude 1.45. The phase modulation angle β is indicated.

An ideal balanced modulator is considered as a multiplier of two signals. It can be seen as a combination of two AM modulators where the carrier and modulation applied to the second AM modulator are both inverted with respect to the first. The outputs from the two AM modulators when balanced are summed to leave a double sideband suppressed carrier (DSBSC) signal. This process is most commonly implemented with a switched or saturated carrier and will be shown as such.

AM signal 1: cos(<O_ct)(l+m)

AM signal 2: -cos(co_ct)(l-m))

Adding these two AM signals cancels the carrier and reinforces the sidebands producing DSBSC:

2m.cos(co_ct) == 2Meos(ra_mt)cos(ra_ct) == Mcos(ra_c-ra_m)t+Mcos(ra_c+ra_m)t for m = Mcos(ra_mt)

There are many devices that produce DSBSC based on balanced modulators ^[8·⁹·^10,111.

SSB can be derived from DSB or AM by rejecting the unwanted sideband and carrier if present with a selective filter^[8]. This is known as the Filter Method of producing SSB. This is almost invariably a low power scheme as the filtering requirement is severe for audio band modulation. A low level balanced modulator is normally used to facilitate the rejection of the carrier before the filter ¹⁸¹.

A second method to generate SSB is known as the Hartley Modulator¹¹²¹ or Phasing Method. This method uses the output from a first balanced modulator with an in-phase carrier and in-phase modulation and adds the output from a second balanced modulator with quadrature phase carrier and quadrature phase modulation. For SSB the quadrature modulation components are taken as generated by the Hilbert transform of the in-phase components. The Hilbert transform of cos(x) is sin(x) and the Hilbert transform of sin(x) is -cos(x). Using this definition this method of generating SSB will be shown mathematically:

First balanced modulator in-phase carrier: cos(oyt)

First balanced modulator in-phase modulation: mi = Mcos(ra_mt)

Second balanced modulator quadrature carrier: -sin(oyt)

Second balanced modulator quadrature modulation: m_q= Msin(ra_mt)

First balanced modulator DSBSC output:

2Meos(ra_mt)cos(ra_ct) == Mcos(ra_c+ra_m)t + Meos(ra_c-ra_m)t

Second balanced modulator DSBSC output:

-2Msin(ra_mt)sin(ra_ct) == Mcos(oy+oi,„)t - Meos(ra_c-m_m)t

Summing these two DSBSC outputs results in SSBSC (here USB): 2Mcos(oy+M,„)t

An inversion of phase of either of the modulation components produces the other sideband.

When the modulating waveforms are independent, the summation of the two DSB signals are considered as: m,cos(oyt) - m_qsin(co_ct) == Rc((m, + j.m_q) cxp(joyt)) or (mi + j.m_q) as a phasor.

This is the general form of quadrature amplitude modulation (QAM). Figure lb depicts an example of summation of 2 phasors, representing in-phase and quadrature phase DSB signals, for an instant when (mi + j.m_q) = (0.45 + 0.6 j). The resultant QAM signal with instantaneous amplitude 0.75 is shown as a dashed line.

A third method known as the Third Method or Weaver Method ¹¹³¹ combines the Filter Method and Phasing Method and involves a secondary carrier and additional balanced modulators.

The Phasing Method and variants ¹¹⁴¹ and the Third Method and variants ^[15·¹⁶·¹⁷¹ use balanced modulators and are invariably implemented at low power due to the complexity and linearity requirements.

The present invention presents a further method for generating QAM or SSB amongst other modulation modes by the use of multiple AM or PM carriers which are separately modulated utilizing quadrature relationships if required and combined as a single direct summation at the output. The principal of the invention of direct single stage summation of the outputs from a number of independent AM or PM signals is extendable to an arbitrary number of AM or PM signals and need not involve quadrature relationships allowing for example a multiple channel transmitter. The preferred embodiment is as a SSBSC or QAM generator at high power using three or four AM or two or four PM carriers and the following analysis will focus on these approaches.

The process of generating SSBSC or QAM utilizing four AM signals is as follows:

First AM in-phase carrier and modulation signal: cos(co_ct)( 1+ mi)

Second AM quadrature carrier and modulation signal: -sin(co_ct) (1+ m,)

Third AM anti-phase carrier signal and modulation signal: -cos(co_ct)( 1- m_ia)

Fourth AM anti-quadrature phase carrier signal and modulation signal: sin(co_ct)( 1- m,_|a)

The summation of these four signals is:

(mi + m_ia)cos(co_ct) - (m_q + ηι,^βίη/ωΤ) which is the USB where m_q and m_qq are the Hilbert transforms of mi and m_ia respectively or QAM if the modulations are not related by the Hilbert transform.

The LSB is obtained by inverting m_q and m_qa. mi and m_ia can be independent and where the opposite sign is taken for the quadrature components Independent Sideband (ISB) results.

SSBSC will be exemplified with sinusoidal modulation as previously.

Where mi = Mcos(<B_mt); m_q = Msin(ro_mt); m_ia = mi; m_qa = m_q Expanding to reveal the spectral components:

First AM signal: cos(co_ct)( 1+ mj) = cos(ort) + AMcostor+orJt + AMcos(co_c-co,n)t

Second AM signal: sin(oj_ct) ( 1+ m_q) = -sin(co_ct) + AMcostov+orJt - AMcos(co_c-co,n)t

Third AM signal: -cos(co_ct)( 1- m_ia) = -cos(co_ct) + AMcos(co_c+cOm)t + AMcos(co_c-cOm)t

Fourth AM signal: sin(oj_ct)( 1- m_qa) = sin(co_ct) + AMcos(co_c+<O_m)t - AMcos(co_c-0)_m)t

Simultaneous direct summing of these twelve spectral components of the four AM signals yields SSB directly (in this case USB): 2Meos(ro_c+ro_m)t. The carrier and opposite sideband signals cancel.

Figure lc depicts the phasor example for the summation of 4 AM signals corresponding to the QAM example of mi = m_ia = 0.45 and m_q = m_qa = 0.6 j. The 4 AM technique produces an output of double the output from the 2 DSB technique. The phasors in the diagram are therefore scaled by A for ease of comparison. The 4 AM signals are shown in a sequence for clarity only, as summation is a single simultaneous operation. The sequence shown is: in-phase AM of instantaneous amplitude A (1.45), quadrature AM of instantaneous amplitude A (1.6j), inphase AM of instantaneous amplitude A (-0.55) and quadrature AM of instantaneous amplitude A (-0.4j) with the same resultant QAM signal of instantaneous amplitude A (1.5) = 0.75.

A simpler situation results when m_ia and m_qq are zero i.e. unmodulated. The output summation result, Mcos(ro_c+ro_m)t, is still SSB but at half the amplitude of the 4 AM method. In this circumstance a three AM signal form can be achieved by substituting a single unmodulated carrier signal for the unmodulated third and fourth AM signals. The identity that is used is:

cos(co_ct) - sin(oU) == /2cos(co_ct + π/4)

The third signal, the anti-mid-phase carrier, is then a single signal: -/2cos(co_ct + π/4) == /2cos(co_ct - 3π/4)

The output summation with m_ia and m_qa set to zero is unchanged by the substitution of this equivalent signal and therefore permits an implementation with 3 AM signals one of which is unmodulated. Figure Id depicts the phasor diagram for this situation for mj = 0.45 and m_q = 0.6 j. No scaling is required for this technique as there is no secondary modulation (m_ia and m_qa) applied and the resultant is the same amplitude as the 2 DSB case. Again the sequence is for clarity only. The sequence depicted is an unmodulated carrier of amplitude /2 at an angle

-3π/4, an in-phase AM of instantaneous amplitude 1.45 and a quadrature AM of instantaneous amplitude 1.6j with the same resultant QAM of instantaneous amplitude 0.75.

The Chireix Outphasing technique ¹¹¹ could be applied to the summation modulator and used to generate the 4 AM carriers required to generate SSBSC by creating 8 phase modulated constant amplitude carriers. Similarly the Outphasing technique could be applied to the 3 AM form requiring two AM carriers and one unmodulated carrier. Using Outphasing there would then be four PM carriers and one unmodulated carrier presented to the summation. The advantage is there is no requirement to apply direct AM to the output stages. The equivalent effect of the AM modulations is realized within the summation without being generated specifically. The disadvantage is the increase in the number of output devices and summation elements and the waveform shaping required for modulation linearity.

A novel solution expounded within this invention is to generate two or four constant amplitude PM carriers which when summed produce SSBSC or more generally QAM.

The 4 PM signals required are:

The in-phase pair: $ίη(ω4+β,): -sin(oyt-|i,) where βι = arcsin(nii) for |mi|<l and |βι|< π/2

The quadrature phase pair: εοβ/ωΤ+β,); -cos(co_ct^_q) where β_(| = arcsin(m_q) for |m_q|<l and |β„|< π/2

Analyzing this generally:

The 4 PM carriers can be expanded as:

βίη/ωΤ+βι) == sin(cirt) cos(β,) + cos(co_ct) siη(β,)

-βίη/ωΤ-βί) == -sin(oyt) cos(β,) + cos(oyt) sin(β,) cos(ortNfi) == cos(ort) cos^_q) - sin(ort) si η(β_(|) —cos(co_ct—β_ς) == -cos(co_ct) cos^_q) - si ii(oj_qt) βίη/β,)

In the summation the resultant is:

2(cos(oyt) εΐη/βί) - sin(oyt) sin(β,)) == 2(m, cos(oyt) - m_q sin(oyt)) The general expression for QAM.

For SSB mi and m_q are the pair of quadrature (Hilbert transform) modulations.

These 4 PM signals will be shown to generate SSB by analysis with sinusoidal modulation as previously.

The example sinusoidal modulations for USB are: mi = Mcos(co,nt); ni,, = Msin(ro_mt) with |M| < 1 The required transformations are:

βι = arcsin(Mcos(co_mt)) and β, = arcsin(Msin(ro_mt))

The resultant in this case is then

2(cos(oyt) si n( [$,) - sin(oyt) si η(β_(|)) == 2M(cos(co_ct) cos(ro_mt) - sin(oyt) sin(ro_mt)) == 2Mcos((co_c + 0J„,)t) which is the required SSBSC signal (in this example USB).

Figure le depicts the summation of 4 PM signals as phasors. A scaling factor of /2 is again applied for ease of comparison as the 4 PM method produces double the output of the 2 D SB method. The phasor sequence shown in the summation is for clarity only. The sequence is the in-phase pair of PM signals of amplitude /2 (1) and the quadrature pair of PM signals of amplitude /2 (1). The resultant QAM signal is again of instantaneous amplitude /2(1.5) = 0.75.

There are a number of advantages to this method of generating SSB with four PM carriers. The 4 PM carriers have constant and equal amplitudes which means the final stages feeding the summation can advantageously be operated saturated and with a common supply. Combining the in-phase signals as a pair and the quadrature phase signals as a pair has the advantage of no negative power flow for each pair over a complete modulation cycle. Utilizing the Chireix Outphasing method for producing AM from two PM would require 8 PM carriers rather than the 4 in this innovation also unlike the Chireix method the amplitude to phase transform is symmetrical and substantially linear about zero amplitude as this corresponds to the inverse sine function with phase deviation about zero phase.

The 2 PM approach with the summation modulator generates SSB when the modulating signals mi and m_q, are a pair of quadrature (Hilbert transform) modulations. When the modulations are unrelated then generalized Quadrature Amplitude Modulation (QAM) is generated. The transformations to derive the phase modulation are more complex than the 4 PM case but the reduction in output devices and drives is an advantage. The use of the summation modulator with the two PM approach applies independent phase modulation to each carrier as opposed to the prior art Chireix method of generating AM where one phase modulation is symmetrically equal and opposite to the other as depicted in Figure la.

The 2 PM carriers are:

-sin(co_ct + β_α - Pb) and sin(o\t + β„ + βι») where β_;ι = arctan(m_q/mi) where -π < β„ < π and βι, = arcsin(¹/2/(m_i ² + m/)) where 0 < βι, < π/2 and /(mi² + m,/)< 2

These 2 PM signals will be shown to generate SSB by analysis with sinusoidal modulation as previously.

Using example sinusoidal modulations: mi = Mcos(oj,„t); m_q = Msin(oj,„t) with |M| < 2 It follows that: β_;ι = arctan(Msin(ro_mt) / Mcos(oj,„t)) = arctan(tan(oj,„t)) = oj,„t and βι, = arcsin(^l/2M''/(cos(ojlllt) ²+ sin(romt)²)) = arcsin(M/2)

Summation of the 2 PM carriers:

sin(o\t + βα + βι,) - sin(o\t + β„ - Pb) == 2.sin^).cos(co_ct + p_a) == 2.sin(arcsin(M/2)).cos(co_ct + 0J„,t) == Mcos(co_ct + oj,„t) which is the required SSBSC signal (in this example USB). Inversion of the sign of p_a yields the LSB.

A more general analysis is as follows:

For β_α = arctan(m_q/nii) and Pb = arcsin(¹/N(mi²+ m,²)) where -π < pa < π

It follows that tan(p_a) = m_q/nii, sin(p_a) = m_q //(m/+ m,/) and cos(p_a) = mi //(m/+ m,/) and that sin(Pb) = A(m,²+m_q ²)

The summed PM signals are:

sin(o\t + p_a+ Pb) - sin(o\t + p_a - Pb) = 2.sin(p_b).eos(ro_ct + p_a) = /(111/+ m/j.icostPaj.costwT) - sin(p_a).sin(co_ct)) = /(m/+111/).((1¾ //(m/ + m_q ²)).cos(co_ct) - (in, //(m/+ m,/)).sin(wT)) = mi cos(wT) - m_q sin(wT)

i.e. generalized Quadrature Amplitude Modulation (QAM)

This is the analytic expression for the USB where m_qis the Hilbert transform of mi. As before the inversion of the sign of p_a yields the LSB: mi cos(wT) + m_q sin(co_ct)

Figure If depicts the phasor diagram for the same QAM example as used previously. A scaling factor is not required as the 2 PM method produces an output of the same amplitude as the 2 DSB method. The 2 PM signals in the example are therefore of unit amplitude. The resultant is the QAM signal of instantaneous amplitude 0.75.

A preferred embodiment for a SSB or QAM transmitter using either PM technique would use a Digital Signal Processor (DSP) or equivalent to generate the phase modulation.

The essential difference between this invention and the prior art is that there are no intermediate stages, filters or processes requiring amplitude linearity and specifically there is no intermediate DSB and no two input balanced modulators. There are only AM or PM modulators and the SSB or QAM signal is derived directly from and in the output summation. Advantages accrue in that the AM or PM generators can be at any power level required with no further amplification needed and efficient non-linear techniques can be used throughout. The independence of the AM or PM signals allows carrier suppression to whatever degree is required including no suppression and the ability to apply multiple modes of modulation simultaneously. The efficiency is also maximized in that with the modulator set to produce for example SSBSC there is ideally no dissipation associated with the unwanted sideband and carrier components as they are cancelled and effectively no loading is presented at those spectral components within the modulator. The output load is actually present but no current flows for those spectral components as the net voltage output is zero for those spectral components due to the cancellation. Therefore no individual AM or PM generator is producing power for these spectral components. This ideal can be closely approached even with switching waveforms. In the preferred embodiment using pulse waveforms rather than sinusoidal waves for the carriers the even harmonics are cancelled by using balanced waves and a composite wave of main and auxiliary pulse waveforms cancels the 3^rd and 5^th harmonics. The higher order odd harmonics can be rejected with a rising input impedance output filter (e.g. inductive input low pass filter) leaving the loading on the AM or PM sources without dissipation at all these unwanted spectral components. The main and auxiliary wave technique will be described in detail later.

The descriptions of this invention have focused on the SSB application utilizing a Hilbert transform. The schemes described here utilize in-phase carrier modulation mi and quadrature phase carrier modulation m_qand can accommodate the situation where the modulations are independent (that is not related by the Hilbert transform). This then facilitates the more general modulation families of Quadrature Amplitude Modulation (QAM) and Quadrature Phase Shift Keying (QPSK). More than 4 modulated carriers can be summed permitting a variety of modulation scenarios including multiple transmissions. The limits are practical rather than theoretical.

The modulator will be described in terms of series summation of low impedance sources as being the preferred embodiment. Parallel summation based on current generators is also possible. An important requirement for the high power use of this modulator in the preferred embodiment is that the entire summation series is continuously and individually at low impedance. In order to provide the low impedance sources for the series summation each individual source must be able to sink or supply current as required. The output of each summation component should be low impedance whether the individual output is in-phase, anti-phase or zero with respect to the summed output. This is most easily achieved by having no zero output states for each component in the summation and is a preferred implementation for high power. A second desirable aspect for a preferred implementation is the use of pulse waveforms to maximise efficiency by using switching techniques.

A disadvantage with square wave carrier (class D) transmitters is the high levels of harmonics in the output. The presence of odd harmonics is an intrinsic characteristic of square waves. The preferred implementation of the invention described suppresses the even harmonics by the use of the well known balanced waveform technique and suppresses the 3^rd and 5^th harmonics of the main square wave by the use of an auxiliary pulse waveform at a lower amplitude and which preferably is also arranged to have no zero states and is also a balanced waveform. This arrangement means the difficult filtering requirements for efficient and clean output from a class D transmitter are much reduced as many of the harmonics are intrinsically suppressed. It should be emphasized that a composite carrier of auxiliary and main wave is not actually generated prior to the single direct summation.

The composite carrier only appears in principal within this summation with all the other components to produce the single output.

Some waveforms that suppress 3^rd and 5^th harmonics are depicted in figure 2.

Figure 2a: Main pulse train + Auxiliary pulse train scaled at 0.707 = Composite pulse train

Zero voltage states are present in the main, auxiliary and composite pulse trains.

Figure 2b: Main pulse train + Auxiliary pulse train scaled at 0.414 = Composite pulse train

Zero voltage states are present in main and auxiliary but not in the composite pulse train.

Figure 2c: Main pulse train + Auxiliary pulse train scaled at 0.414 = Composite pulse train

No zero voltage states are present.

The pulse train strategy of Figure 2c is preferred as the implementation is straightforward. The process of suppression of the 3^rd and 5^th harmonics will be shown mathematically using the waveforms of figure 2c as an example though the same principal applies to all 3 waveforms of figure 2. The design basis exampled subdivides the carrier cycle into 8 equal durations.

The amplitudes of the spectral components of the Main wave of fig. 2c (a balanced square wave) are: (8/(mt)).sin(mt/8).(l-cos(mt)).cos(n;t/4).cos(mt/8) where n is the harmonic number (n = 1 being the fundamental, n = 3 being the 3^rd harmonic etc.)

The amplitudes of the spectral components of the Auxiliary wave of fig. 2c (a balanced pulse train) are:

(8/(ηπ)). sin(mt/8). (l-cos(mt)). sin(mt/4). sin(mt/8)

Both are balanced waves and eliminate even harmonics through the function (l-cos(mt)) which is zero for n even.

The ratio of the Auxiliary wave to Main wave spectral components is: tan(mr/4).tan(mt/8)

Function tan(ιιπ/4) alternates between +1 and -1 as n is stepped through the odd harmonic values 1,3,5,7 etc. and specifically for n = 3, tan(3W4) = -1 and for n = 5, tan(5W4) = 1

Function tan(nW8) alternates in sign and value and specifically tan(3W8) = - tan(5W8) == 2.414

The ratio function tan(mr/4).tan(mt/8) is then equal to -tan(3W8) = - l/tan(W8) for both n = 3 and n = 5 Therefore function tan(mt/4).tan(mr/8).tan(jt/8) = -1 for n = 3 or 5 (and 11,13 etc.)

The technique to suppress the 3¹ and 5^th harmonics (and cyclically 11^th and 13^th etc.) is to scale the Auxiliary wave by tan(WS) == 0.414 and add it to the Main wave.

The spectral components of the summed Main and Auxiliaiy wave composite are then:

(8/(mt)).sin(mt/8).(l-cos(mt)).cos(njt/4).cos(mt/8).(l+tan(mt/4).tan(mi/8).tan(jt/8))

The components at n = 3,5 (and 11,13 etc.) are zero as well as for all n even.

This technique to suppress the 3rd and 5th harmonics using a Main and Auxiliary wave is applied in principal to each AM or PM carrier separately. Phase modulation is applied to both Main and Auxiliary waves equally therefore the relationships above are continuously valid. AM has no effect on the suppression as equal modulation is applied to each Main and Auxiliary wave pair and simply scales both Main and Auxiliary components together. Each spectral component of a Main and Auxiliary wave pair can be considered separately so this is mathematically equivalent to amplitude modulating the composite wave

i.e. {main}.(l+m) + {aux}.(l+m) == {main+aux}.(l+m).

The principal utilized here is key to this invention in that the AM has taken place at the component level at the same time as the summation. It should be realized that the important aspect of this invention is the summation of all the individual components in a single direct output summation. This is the extension of the above principal to all the carriers. To be clear for example with the 3 AM carrier form created from 3 sets of Main and Auxiliary waves appropriately scaled and modulated there are 6 components. All 6 components of this example are then summed in a single direct summation that may also act as part or all of the component scaling. In principal each of the 6 components consists of a fundamental carrier plus two modulation sidebands and harmonics of the carrier with modulation sidebands. The even harmonics cancel in the balanced form of the pulse waveforms. So, exampling the 3 AM carrier form of SSBSC modulator with a main and auxiliary wave adjusted for suppression of 3^rd and 5^th harmonics, the following spectral components are present in the summation of the 6 components: Anti-mid-phase main carrier + odd harmonics (no modulation applied)

Anti-mid-phase auxiliary carrier + odd harmonics (no modulation applied)

In-phase main carrier + odd harmonics all with In-phase modulation sidebands

In-phase auxiliaiy carrier + odd harmonics all with In-phase modulation sidebands

Quadrature-phase main carrier + odd harmonics all with Quadrature-phase modulation sidebands

Quadrature-phase auxiliary carrier + odd harmonics all with Quadrature-phase modulation sidebands

In the single output summation the carriers cancel, the unwanted sidebands cancel and the 3 ^rd and 5^th harmonics and their sidebands cancel. This leaves the wanted SSBSC and harmonics of order 7^th and higher.

The use of a DSP or equivalent to generate the required PM for a two or four PM carrier system could be extended to the direct generation of the phase shifted pulse waveforms of main and auxiliary pulse waveforms. It is therefore envisaged that a preferred embodiment of this invention would use a DSP or equivalent to generate the pulse waveforms required i.e. a main and auxiliary waveform for each of the PM carriers as well as perform the Hilbert transform for in-phase/quadrature modulation and the appropriate wave shaping for linearity and conversion to phase shift modulation. The same approach of using a DSP can be applied to the 3 or 4 AM scheme. Other solutions could use a Hilbert transform codec and logic circuits to obtain the pulse waveforms.

Modulators, mixers and demodulators are essentially the same process the terminology indicating the purpose. This modulator can likewise be seen as a mixer or demodulator for frequency changing or detection purposes. The terms in co_ct and ω,,Ι in the analysis above do not have any fixed relationship and can for example be a radio receiver local oscillator and signal frequency with the output being an intermediate frequency or the baseband of the required single sideband. This is a direct form of image rejection mixer or demodulator. The prior art can achieve this by the use of two or more balanced modulators yielding intermediate DSB outputs that are then recombined to reject the image or provide the sideband selective demodulation. They are seen as complex devices. The use of the summation modulator of the present invention allows the direct output of the desired intermediate frequency or baseband without any intermediate stages. To fulfill these roles the elements used to implement the summation modulator of the present invention need adjusting appropriately to accord with the frequencies and power being handled.

The present invention can be utilized in many different designs whilst retaining the essential aspect of a single summation of multiple AM or PM signals. More generally any generator and summation means for the AM or PM signals can be used. The invention resides in the direct single summation at the output and the characteristics of the input signals to generate the modulated carriers. The waveforms for the carrier waves can be sinusoidal, part sinusoid or pulse forms.

Summation of the multiple signals will be exemplified using one specific combination of waveforms and transformer topology however other topologies and techniques to achieve this single direct summation can be used within the scope of this invention. Specifically implementation using only square waves (class D) without the addition of the auxiliary pulse train is viable with the consequence of reduced harmonic suppression. For high power or efficiency there is a practical need that the signal sources are individually continuously low impedance for series summation or high impedance for parallel summation.

The series summation case for the 4 PM modulator with pulse form carriers is illustrated schematically in Figure 4. Here the various inputs are summed by both difference and addition. Low output impedance buffers are used to drive the primaries of the 4 transformers, T1 to T4, differentially with the required input signals. A common supply is implied and the ratio of main to auxiliary waves is achieved using transformer ratios.

Inputs Via and Vlb are Main (square) wave for sin(ort+[f) and sintojt-β,) respectively.

Inputs V2a and V2b are Auxiliary pulse train for sin(oj_ct+|f) and s i n(oj_ct($,) respectively.

Inputs V3a and V3b are Main (square) wave for cos(co_ct+3_q) and cos(co_ct-3_q) respectively.

Inputs V4a and V4b are Auxiliary pulse train for cos(co_ct+3_q) and eos(co_ct-3_q) respectively.

Transformers T1 and T3 (Main waves) use a nominal ratio of 12 primary to 12 secondary.

Transformers T2 and T4 (Auxiliary waves) use a nominal ratio of 29 primary to 12 secondary.

The output is from the series summation of the secondaries of T1 to T4. The ideal ratio for suppression of 3^rd and 5^th harmonics is an Auxiliary wave to Main wave ratio of tan(WS). This is closely approximated by the 12/29 ratio of the Main to Auxiliary primary windings.

Similarly the 2 PM series summation case is illustrated schematically in Figure 5.

Inputs Via and Vlb are Main (square) wave for sin(or_qt + β_;ι + βι,) and sin(oj_ct + β_;ι - βι,) respectively.

Inputs V2a and V2b are Auxiliary pulse train for si n(or_qt + β_;ι + βι») and si ir(or_ct + β_;ι βι,) respectively.

Transformer T1 (Main wave) uses a nominal ratio of 12 primary to 12 secondary.

Transformer T2 (Auxiliary wave) uses a nominal ratio of 29 primary to 12 secondary.

The output is from the series summation of the secondaries of T1 and T2.

The invention will be described further by way of a simple outline transmitter design example. This is illustrated in Figure 6. The basis is the 4 AM summation modulator with Main and Auxiliary pulse waveforms applied as in-phase and quadrature phase. There are therefore 4 carrier inputs. The Main and Auxiliary waveforms used are depicted in Figure 3. This figure depicts the preferred waveform of Figure 2c in the required phase relationships.

Figure 3a In-phase carrier, cos(ort): Main + Auxiliary = Composite

Figure 3b Quadrature phase carrier, si n(or_qt): Main + Auxiliary = Composite

Figure 3c Anti-mid-phase carrier - used in the 3 AM technique not required in the 4 AM technique

Note the scaling shown in Figure 2 and Figure 3 is purely to illustrate the eventual amplitude relationships referred to the output rather than the amplitude of the applied input pulses.

The switch devices, shown in Figure 6 as MOSFETs for low impedance on state, are driven by transformer secondaries applying the in-phase or quadrature phase, Main or Auxiliary wave as appropriate. The carrier inputs shown in Figure 6 are:

In-phase carrier: Main input to VT1, Auxiliary input to VT2 (see Figure 3a for waveforms)

Quadrature phase carrier: Main input to VT3, Auxiliary input to VT4 (see Figure 3b for waveforms)

Positive and negative voltage power supplies are applied to Vs+ and Vs- respectively.

The modulations, mi and m,. are applied to inputs Vmi and Vmq respectively so as to modulate the switching pairs differentially. That is the upper switch pairs receive amplitude modulation Vs(l - m_x) and the lower switch pairs receive amplitude modulation Vs(l + m_x).

In the same way as the 4 PM example of Figure 4 the four output transformers, T5 to T8, combine the 8 signals differentially in pairs and use transformer ratios to optimize the suppression of 3 ^rd and 5^th harmonics within the overall series summation.

Transformers T5 and T7 (Main waves) use a nominal ratio of 12 primary to 12 secondaiy.

Transformers T6 and T8 (Auxiliary waves) use a nominal ratio of 29 primary to 12 secondary.

The series summation of the secondaries of T5 to T8 is applied to an inductive input low pass filter LPF1 to obtain the transmitter output.

In this example outline design for a 4 AM based transmitter the output is general QAM. SSB results when the modulations, mi and m,. are related by the Hilbert transform.

References:

1: US Patent 1946308 Apparatus for Radio Communication

Henri Chireix, Feb 1934, Priority July 1931

2: US Patent 6836183 Chireix Architecture Using Low Impedance Amplifiers

James S Wight, Dec 2004, Filed 16 Oct 2002

3: US Patent 2020327 Single Side Band Transmission

Ellison S.Purington, Nov 1935, Filed 6 Sep 1930

4: US Patent 2666133 Single Sideband Transmitter

Leonard R.Kahn, Jan 1954, Filed 16 Aug 1951

5: Dynamic Power Supply Design for High-Efficiency Wireless Transmitters

Jason T. Stauth and Seth R. Sanders

Masters Research Project: Submitted to the Department of Electrical Engineering and Computer Sciences, University of California at Berkeley (2006)

6: RF Power Amplifier Techniques for Spectral Efficiency and Software-Defined Radio

Rameswor Shrestha

ISBN: 978-90-365-3076-7, ISSN: 1381-3617 (CTIT Ph.D. thesis series No. 10-175) (2010)

7: High Average-Efficiency Multimode RF Transmitter Using a Hybrid Quadrature Polar Modulator Chien-Jung Li, Student Member, IEEE, Chi-Tsan Chen, Tzyy-Sheng Homg, Senior Member, IEEE, Je-Kuan Jau, Member, IEEE, and Jian-Yu Li

IEEE Transactions on Circuits and Systems—II: Express Briefs, Vol. 55, No. 3, March 2008

8: US Patent 1449382 Method and Means for Signalling with High Frequency Waves

JohnR.Carson, Mar 1923, Filed Dec 1 1915

9: US Patent 1984156 Modulation System

Ellison S.Purington Dec 1934, Filed 18 May 1929

10: US Patent 2025158 Modulating System

Frank A.Cowan, Dec 1935, Filed Jun 7 1934

11: US Patent 3275950 Double Sideband Suppressed Carrier Balanced Modulator Circuit

Edmund E.Birr, Sept 1966, Filed Apr 29 1963

12: US Patent 1666206 Modulation System

Ralph V.L.Hartley, Apr 1928, Filed Jan 15 1925

13: US Patent 2928055 Single Sideband Modulator

Donald K.Weaver, Mar 1960, Filed 17 Dec 1956

14: US Patent 3195073 Single-Sideband Suppressed Carrier Signal Generator

Thomas C.Penn, Jul 1965, Filed 26 Jul 1961

15: US Patent 1719052 Single Sideband Carrier System

Estill I. Green, Jul 1929, Filed 20 Sep 1926

16: US Patent 1994048 Single Side Band Transmission

Ellison S.Purington Mar 1935, Filed 6 Sep 1930

17: US Patent 2173145 Single Side-band Transmitter

Walter H.Wirkler, Sep 1939, Filed 26 Nov 1937

Claims (3)

Claims:

1 I claim a radio frequency modulator, mixer or demodulator combining in a single direct output summation three or more carrier waves independently amplitude modulated with two or more modulating waveforms.

2 I claim a radio frequency modulator, mixer or demodulator combining in a single direct output summation two or more carrier waves independently phase modulated with two or more modulating waveforms.

3. I claim a radio frequency modulator according to claim 1 or claim 2 where the output of a transmitter is obtained directly from the filtered output from the said composite output from the said modulator.

Intellectual

Property

Office

Application No: GB 1622224.2 Examiner: Dr Thomas Martin

3 I claim a radio frequency modulator combining in a single direct output summation three carrier waves of equal frequency comprising a first carrier wave with amplitude modulation mi, a second carrier wave with nominally +π/2 (alternatively -π/2) radians phase offset from the first with amplitude modulation mqand a third unmodulated carrier wave with nominally +5π/4 (alternatively -5π/4) radians phase offset from the first and where mi and mqare modulating waveforms.

4 I claim a radio frequency modulator combining in a single direct output summation four carrier waves of nominally equal amplitude, frequency and phase comprising a first carrier wave with phase modulation of nominally (π/2 - arcsin(mj)) radians, a second carrier wave with phase modulation of nominally (3π/2 + arcsin(mi)) radians, a third carrier wave with phase modulation of nominally arcsin(mq) radians and a fourth carrier wave with phase modulation of nominally (π - arcsin(mq)) radians and where mi and mqare modulating waveforms.

5 I claim a radio frequency modulator combining in a single direct output summation two carrier waves of nominally equal amplitude, frequency and phase comprising a first carrier wave with phase modulation of nominally (π/2 + arctan(mq/mi) - arcsinf/2/(111/+ m,2))) radians and a second carrier wave with phase modulation of nominally (3π/2 + arctan(mq/mi) + arcsinCZ/fm,2 + mq2))) radians and where mi and mqare modulating waveforms.

6 I claim a modulator according to claim 3, 4 or 5 wherein said modulating waveforms, mi and in,, are related by a relative phase shift of nominally π/2 radians.

7 I claim a radio frequency modulator combining in a single direct output summation four carrier waves of equal frequency comprising a first carrier wave with amplitude modulation mi, a second carrier wave with nominally π/2 (alternatively -π/2) radians phase offset from the first with amplitude modulation mq, a third carrier wave with nominally π radians phase offset from the first with amplitude modulation mia and a fourth carrier wave with nominally π radians phase offset from the second with amplitude modulation m,|a and where mi and mqand miaand mqaare modulating waveforms.

8 I claim a modulator according to claim 7 wherein said modulating waveforms, mi and mq, applied to said first and second carrier waves, are related by a relative phase shift of nominally π/2 radians and wherein said modulating waveforms, miaand 111,,,,, applied to said third and fourth carrier waves, are related by a relative phase shift of nominally π/2 radians and wherein miaand mqamay be dependent on or independent of mi and mq.

9 I claim a modulator according to claim 3 or 7 wherein amplitudes of said carrier waves are arranged such as to suppress output when said carrier waves are unmodulated.

10 I claim a quadrature amplitude modulation (QAM) radio frequency transmitter according to claim 3,4,5 or 7.

11 I claim a single sideband (SSB) radio frequency transmitter according to claim 6 or 8.

12 I claim a radio frequency transmitter using a modulator according to any preceding claim wherein the transmitter output is directly from a filtered output of said modulator.

13 I claim a modulator according to any preceding claim wherein one or more of said carrier waves are balanced switching square waveforms.

14 I claim a modulator according to claim 13 wherein each said carrier wave is paired within said summation with an auxiliary balanced switching pulse waveform arranged to facilitate suppression of third and fifth harmonics of each said carrier wave.

Ammendmetnt to the claims have been filed as follows

20 11 17

Claims:

1. I claim a radio frequency modulator consisting of one or more sets of two linked pairs of bilateral switch devices where a first pair of each of the linked pairs is driven by a primary switching waveform of nominal spectral components (8/(mr))sin(mi:/8)(l-cos(mi))cos(mi:/4)cos(mi/8) where n is the harmonic number of the fundamental radio frequency of the primary waveform; and where a second pair of each of the linked pairs is driven by an auxiliary switching waveform of nominal spectral components (8/(im))sin(nW8)( l-cos(im))sin(nW4)sin(ira/8): and where phase offset and amplitude or phase modulation of each set of linked pairs is applied equally to the first and second pair of the linked pairs; and where the amplitude of the modulated output from the second pair of each of the linked pairs is scaled to nominally a factor of tan(W8) times the amplitude of the modulated output from the first pair of each of the linked pairs; and where the modulated outputs from all pairs are combined by a vector summation to produce a composite modulated output.

2. I claim a radio frequency modulator according to claim 1 consisting of four sets of linked pairs of bilateral switch devices and wherein the fundamental radio frequency of the said primary and auxiliary waveforms driving each of the four sets of linked pairs is equal with relative phase offsets of 0, π/2, π and 3π/2 radians respectively and with amplitude modulation 1+mi, 1+mq, 1-mi and 1-mq applied respectively to the four sets of linked pairs.