GB2459478A

GB2459478A - Telecommunication System with simplified Receiver

Info

Publication number: GB2459478A
Application number: GB0807430A
Authority: GB
Inventors: Clyde Witchard
Original assignee: Audium Semiconductor Ltd
Current assignee: Audium Semiconductor Ltd
Priority date: 2008-04-23
Filing date: 2008-04-23
Publication date: 2009-10-28
Also published as: GB0807430D0

Abstract

In a communication system, a signal is transmitted from a transmitter to a receiver, comprising a carrier having data modulated thereon, and the transmitted signal is received in the receiver, by means of a local synchronization signal. The transmitted signal also includes a plurality of phase probe signals having a known property namely that, when detected in the absence of noise and with a local synchronization signal having a desired phase, the phase probe signals have zero amplitude in at least one phase. The phase probe signals are received, by means of said local synchronization signal, and the receiver determines a mean amplitude of the received phase probe signals in said at least one phase, and adjusts a phase of the local synchronization signal based on said determined mean amplitude.

Description

TELECOMMUNICATION SYSTEM WITH SIMPLIFIED RECEIVER

Telecommunication receivers, for receiving digital data, frequently employ circuits or systems for synchronizing the local carrier signal and symbol clock to the transmitter.

The circuits or systems that carry out this processing are often computationally demanding, leading to implementations that are power-hungry, large and costly.

Furthermore, the processing is usually carried out in the digital domain, leading to the need for one or more analogue-to-digital converters (ADCs) preceding the digital processing stage.

Other telecommunication systems seek to reduce some of the receiver complexity by employing non-coherent modulation schemes, such as frequency shift keying (FSK) or differential phase shift keying (DPSK). These obviate the need for recovery of the transmitter's carrier phase, but they do so to the detriment of other system characteristics. For example, FSK is spectrally inefficient, leading to longer packet lengths in packet-based systems, and hence greater power consumption in the radio frequency (RF) circuitry. DPSK can be more spectrally efficient than FSK, but is less robust to noise and interference than non-differential techniques.

Digital synchronization systems, for carrier or symbol clock, frequently comprise a sub-system for estimating a phase or frequency error, and a further sub-system to correct the error. Both systems may be complex, but typically the error estimator is the more computationally demanding of the two.

According to a first main aspect of the present invention, there is provided a method of operation of a communication system, the method comprising: transmitting a signal from a transmitter to a receiver, the signal comprising a carrier having data modulated thereon; and receiving the transmitted signal in the receiver, by means of a local synchronization signal; and the method further comprising: transmitting in said signal a plurality of phase probe signals having a known property namely that, when detected in the absence of noise and with a local synchronization signal having a desired phase, the phase probe signals have zero amplitude in at least one phase; receiving said phase probe signals in the receiver, by means of said local synchronization signal; determining a mean amplitude of the received phase probe signals in said at least one phase; and adjusting a phase of said local synchronization signal based on said determined mean amplitude.

According to other aspects of the invention, there are provided transmitter and receiver devices that operate in accordance with the method.

The invention thus provides a novel telecommunication system that massively simplifies the receiver at one end of the communication link (the remote station, RS), commensurately reducing its power-consumption, size and cost. In some embodiments, the system is able to avoid completely the use of ADCs. Instead, a simple analogue thresholder is used. To estimate carrier or symbol phase offsets, a small counter is employed, rather than the relatively complex estimation algorithms that are frequently seen. Furthermore, any phase correction calculated from this estimate may be applied in the analogue domain, rather than the digital domain, further reducing the total circuitry, and hence power consumption, of the RS. In some embodiments, the system can offer much higher spectral efficiency than FSK, allowing receiver power savings in the RF circuitry, and it avoids the robustness problems of DPSK.

The application of primary interest is the wireless transmission of audio data to battery-powered loudspeakers or earphones, in which extended battery operating lifetime, and hence low power consumption, is desirable. In this application, a base station (BS) resides in the audio source, and one RS resides in each loudspeaker or earphone unit of the audio system. Clearly, however, the communication system presented here has utility in many other power-, cost-or size-sensitive data telecommunication applications.

To show how the present invention may be put into effect reference will now be made, by way of example, to the accompanying drawings, in which:-Figure 1 is a block schematic diagram of a telecommunication system in accordance with an aspect of the present invention.

Figure 2 illustrates a quadrature constellation map, for explanation of an aspect of the present invention.

Figure 3 illustrates a symbol sequence, for explanation of a further aspect of the present invention.

Figure 4 illustrates a quadrature constellation map, for explanation of a further aspect of the present invention.

Figure 1 illustrates a telecommunication system 10 in accordance with an embodiment of the present invention. A base station (BS) 20 communicates with a remote station (RS) 30 by radio. The BS 20 includes a transmitter 22. Likewise, the RS 30 includes a receiver 34. The RS 30 can for example be a battery powered audio device, that receives signals from the transmitter 22 and plays back sound to a listener.

The signalling between the BS transmitter 22 and RS receiver 34 is based on single carrier modulation. Although the invention is described herein with reference to a wireless communication system 10, it will be apparent that the invention can also be implemented in a wired communication system, in particular where data is encoded by modulating a carrier signal, and especially where data is encoded in the phase of a sinusoidal single carrier signal. We start by considering quadrature phase shift keying (QPSK).

The BS transmitter 22 receives the payload data, and passes this to a probe insert block 222, in which probe signals are inserted, as will be described in more detail below. In-phase and quadrature components I and Q of the resulting signal are passed to a pulse shaping filter 226, which is a conventional telecommunication element, and so is not described in detail herein.

The I and Q components of the pulse shaped digital signal are passed to respective digital-to-analogue converters (DAC) 230, 232, and then to baseband filters 234, 236.

The filtered signals are passed to the transmitter radio frequency (RF) circuitry 238, where they are upconverted to the intended RE frequency for transmission by means of an antenna 28.

As already noted, the RS receiver 34 has very minimal circuitry. It must synchronize both carrier and symbol phase to the QPSK signal received from the BS transmitter 22, and decode that signal. Since no analogue-to-digital converter (ADC) is present in the RS receiver 34, this presents a significant challenge. We now describe a solution in detail.

The antenna 38 at the RS receives the radio signal transmitted from the BS 20, and the received signal is fed to receiver RF circuitry 42, which amplifies and frequency-shifts (mixes) the signal down to complex analogue baseband. A down-mix local oscillator 41 generates a sinusoidal signal (which may be real-only or quadrature, depending on the system architecture). The sinusoidal signal is passed to the mixer in RE circuitry 42. As will be described in more detail below, the frequency of the down-mix local oscillator 41 can be adjusted, and the output of the RF circuitry 42 has an accordingly adjusted carrier phase.

The complex output from receiver RE circuitry 42 is applied to analogue baseband filters 44, 46, for the in-phase and quadrature components I and Q respectively. These filters are low-pass, with a nominal cut-off frequency at half the symbol rate. They are identical, within manufacturing tolerances, to each other. They form, to a good approximation, matched filters for the purpose of optimal symbol detection in noise, and they also provide frequency selectivity, extracting the wanted frequency channel from unwanted frequencies. For pulse shaping purposes, the cascade response of the filters in BS transmitter 22, feeding into filters 44 and 46, is chosen to be significantly free of inter-symbol interference.

The outputs of baseband filters 44, 46 are fed to thresholders 48 and 50, respectively, which output a high binary logic value when their input signal is greater than zero; otherwise, they output a low binary logic value. The outputs of the thresholders 48 and are fed into registers 52 and 54, respectively, which sample the thresholder outputs on the active (e.g. rising) edge of a clock signal. The clock signal is simply a regular square wave at the intended symbol frequency, generated by a symbol clock 43. As will be described in more detail below, the frequency of the symbol clock 43 can be adjusted, and the output of the registers 52 and 54 have an accordingly adjusted symbol phase.

In the system so far outlined, we have no carrier or symbol synchronization within the RS receiver 34, and data communication would not be possible. Therefore, methods are provided to form measures of the synchronization errors in the carrier phase and symbol phase (which we refer to generically as "waveform" synchronization errors) of the signal received at the RS receiver 34, and to correct these synchronization errors.

As already noted, one issue for measurements at the RS receiver 34 is that no ADC is present, only the binary thresholders 48 and 50, sampled at symbol rate by the registers 52 and 54. To obtain useful measurements from the outputs of the registers 52 and 54, some special symbols are inserted into the main data-bearing symbol stream in the probe insert block 222 of the BS transmitter 22. These symbols constitute what are termed here phase probes, and two types of phase probe are employed in this illustrated embodiment, namely carrier phase probes, used to estimate carrier phase offsets; and symbol phase probes, used to estimate symbol phase offsets.

A carrier phase probe is a symbol inserted in the transmitted signal, to allow the receiver to form a measure of carrier phase synchronization errors. Specifically, in this embodiment, the carrier phase probe comprises a special symbol that is transmitted by the BS transmitter 22 at the normal symbol timing phase, but, rather than using one of the four conventional QPSK constellation points for its complex amplitude, the transmitted symbol ampJitude is placed exactly on the one of the decision boundaries between the normal QPSK constellation points. Figure 2 shows how carrier phase probe symbols compare to standard QPSK symbols.

Thus, in a QPSK system, data is transmitted by modulating the carrier signal with a signal of known magnitude a, where the data is carried by the phase of the signal during any particular symbol period. In particular, the phase can be equal to 45°, 135°, 225° or 315°. The I and Q components therefore have the same magnitudes (i.e. aI"i2), and the data is carried by the signs of the I and Q components.

In this illustrated embodiment of the invention, the possible carrier phase probes also have the known magnitude a, but the phase can be equal to 00, 900, 180° or 270°. In each case, one or other of the I and Q components has zero magnitude.

Carrier phase probes are inserted periodically into the main data-bearing symbol stream, according to some predetermined insertion pattern, by the carrier/symbol phase probe insert unit 222.

The same insertion pattern is output by a probe timing control unit 60 in the probe measurement block 40, and fed to the enable input of an accumulator 62.

For the purposes of initial illustration, we will now assume that every carrier phase probe uses the complex amplitude 80 shown in Figure 2. In this restricted case, the outputs of a probe quadrant control unit 64 are fixed such that a multiplexer 66 always selects the output of the register 52, and the line from the probe quadrant control unit 64 to an XOR gate 68 is zero. Thus, in this restricted case, the I component of the sampled symbols are fed uninverted to the accumulator 62. The accumulator 62 accumulates a signed value according to the output of the XOR gate 68, such that it decrements by one for a zero from the XOR gate 68, and increments by one for a one from the XOR gate 68; but it increments or decrements only when its enable input is active, i.e. only for phase probe symbols. The accumulation is performed across a predetermined number of carrier phase probe symbols, termed the accumulation period, and the probe timing control unit 60 asserts reset to zero the accumulator 62 at the start of each such period.

The probe timing control unit 60 also outputs a data valid line 70 to designate when symbols are main payload data, rather than phase probe symbols.

To aid understanding of how carrier phase probes are used to estimate carrier phase error, first consider the telecommunication system of Figure 1 to be in a state such that there is no carrier or symbol phase error at the RS receiver 34. In this state, at the sampling instant for a received carrier phase probe, the output of the baseband filter 44 would ideally be zero (reflecting the complex amplitude 80 in Figure 2), but in reality it will be either slightly positive or negative, due to additive zero-mean noise, from the communication channel, thermal noise in the receiver analogue circuitry, etc. Therefore, the output of the register 52 for carrier phase probes will be random, with equal probability of a low or high output, and the accumulator 62 will accumulate a value close to zero over the accumulation period.

However, if we now consider a small carrier phase error at the RS receiver 34, the output of the baseband filter 44 will no longer be zero-mean at the sampling instants for received carrier phase probes. In turn, the output of the register 52 will now have a greater probability of either a low or a high output, so the accumulator 62 will accumulate a value whose sign and magnitude reflect the sign and magnitude of the carrier phase error. In effect, this scheme uses the naturally occurring noise of the channel and receiver circuitry to dither the input to the one-bit quantizer (the thresholder 48), and then averages (with the accumulator 62) to give a value dependent on the input. Beyond a certain magnitude of carrier phase error, there may be insufficient noise to dither the error value, i.e. the output of thresholder 48 will be high for all the carrier phase probes, or low for all the carrier phase probes. This is effectively a form of "clipping" of the measure, and is still of use for steering control loops, or similar systems, so as to correct the error, as discussed below. Thus, we have a system and method that gives a measure of carrier phase error.

The measure from the accumulator 62 is fed to a switch 45, that directs any carrier phase error measures to an offset correction controller 47. The offset correction controller 47 drives a frequency control input of the down-mix local oscillator 41, and completes a control loop that corrects the carrier phase error. The offset correction controller 47 is a conventional element of a carrier phase recovery control loop, as frequently seen in telecommunication receivers and familiar to those skilled in the art, so it is not described further herein. For example, it may be a linear filter.

In the scheme shown, the frequency control input of the down-mix local oscillator 41 is typically analogue, so the offset correction controller 47 must contain a digital-to-analogue converter (DAC) to translate between the digital and analogue domains.

(Again, the DAC may be a conventional element, so we do not describe it in detail.) In many conventional receivers, as already noted, an ADC is used in place of the thresholders 48 and 50 and the registers 52 and 54. In such systems, the carrier phase correction is typically done in the digital domain, often requiring high-speed digital multipliers for the mixer and a digital numerically controlled oscillator to produce a sinusoid signal of the desired frequency. These digital circuits are frequently power-hungry. In the RS receiver 34, we have avoided such circuitry. Since a down-mix oscillator is usually present in a conventional receiver, we have added nothing to the power budget by having the carrier phase corrected here.

We have considered the case where every carrier phase probe uses the complex amplitude 80 shown in Figure 2. However, regular insertion of a fixed amplitude pulse leads to a signal frequency spectrum with greater energy around DC. It is often desirable, for a number of technical or regulatory compliance reasons, to have a flat signal frequency spectrum. To flatten the spectrum, the probe insert unit 222 can alternate the carrier phase probe transmitted between the probe values 80, 82, 84 and 86 shown in Figure 2, according to a pseudo-random pattern. The probe quadrant control unit 64 must then use the same pseudo-random pattern to select either the I value or the Q value, by controlling which of the outputs of the registers 52, 54 is passed by the multiplexer 66, and to determine whether or not this output value should be inverted by the XOR gate 68, by controlling the binary value supplied to the second input of the XOR gate.

A somewhat similar method can also be used to achieve symbol phase synchronization at the RS receiver 34. The method uses symbol phase probes, rather than carrier phase probes.

A symbol phase probe is a signal inserted in the transmitted signal, to allow the receiver to form a measure of symbol phase synchronization errors. Specifically, in this embodiment, the symbol phase probe comprises two special symbols that are inserted by the probe insert block 222, and transmitted by the BS transmitter 22. Figure 3 shows one possible arrangement. Symbols are usually transmitted at symbol timing points t0, t1, ... etc. The two symbols 90, 92 forming the symbol phase probe are adjacent, with one being the inverse of the other, but they are not transmitted at a normal symbol timing phase. Rather, they are transmitted either side of a normal symbol timing point t4, by an equal interval; for example, in Figure 3 the two symbols 90 and 92 are placed half a symbol period either side of the normal symbol timing point t4.

The principle of operation, and the options for implementation, are very similar to the carrier phase probe method just described, so the discussion is not repeated in full. In addition to the carrier phase probes, the probe measurement unit 40 may also be used to measure the symbol phase probes. Alternatively, a second probe measurement unit may be added, allowing carrier and symbol phase probes to be interleaved, and simultaneously accumulated.

The symbol phase probe method is based on the fact that the transmitted symbol phase probe signal crosses zero at point t4 in Figure 3. This corresponds to the ideal sampling instant for the symbol phase probe at the RS receiver 34. So, with no symbol phase error at the RS receiver 34, the output of the baseband filter 52 would ideally be zero, but in practice it will be zero-mean noise (from the communication channel, analogue circuitry etc.) that is averaged to a value close to zero by the accumulator 62.

If the symbol phase is perturbed one way or the other, we effectively move up or down the curve 94 either side of the point t4 in Figure 3. Again, the noise effectively dithers the value on the curve, so, even after the one-bit quantization of the thresholder 52, a measure of the value (and hence the symbol phase offset) is averaged by the accumulator 62. Beyond a certain magnitude of symbol phase error, likewise, there may be insufficient noise to dither the error value and "clipping" will occur.

It will be noted than an equivalent alternative would be to transmit the two symbols forming the symbol phase probe at normal transmit timing points, for example t3 and t4, or t4 and t5, and to sample the received signal at a time midway between these two transmit timing points.

Similarly, the measure is fed to the switch 45, that directs any symbol phase error measures to an offset correction controller 49. The offset correction controller 49 drives a frequency control input of the symbol clock 43, and completes a control loop that corrects the symbol phase error. The offset correction controller 49 is a conventional element of a symbol phase recovery control loop, and familiar to those skilled in the art, so it is not described further herein.

As with the carrier recovery system, it may be possible to perform the symbol phase correction in the analogue domain, for example if the symbol clock 43 is a voltage controlled crystal oscillator (VCXO). Again, this can significantly reduce power consumption compared to a conventional digital receiver chain, which would typically employ a relatively large and complex digital re-sampler.

Since the symbol phase probes are not transmitted on the normal symbol timing points, ISI from the probes can affect adjoining data symbols. Therefore, it may be desirable to leave a short "guard" period before and after each symbol phase probe, containing no data symbols, to let the probe lSl settle to a low level. For example, in Figure 3, no data symbols are transmitted at the two normal symbol timing points t2 and t3 immediately preceding the symbol phase probe or at the two normal symbol timing points t5 and t6 immediately following the symbol phase probe.

Advantage may be gained from synchronizing the mix-down oscillator, or oscillators, in the receiver RF circuitry 42, and the clock signal driving the registers 52 and 54, by deriving them from the same source oscillator (not shown) in the RS receiver 34 for example. Provided the carrier and symbol clocks in the BS transmitter 22 are also derived from a common oscillator at the BS end of the link, this allows just one type of phase probe, either carrier or symbol, to be used. The RS receiver 34 can use the measured changes in one to infer the other, since the ratio between the frequencies of the RS mix-down oscillator, or oscillators, and the clock signal driving the registers 52 and 54 is fixed and known. In particular, it is advantageous to use just the carrier phase probes, and associated measurements, for both the carrier and symbol phase tracking. In typical telecommunication systems, the carrier frequency is much higher than the symbol frequency, so carrier phase offset estimates infer very high accuracy symbol phase offset estimates. Also, each carrier phase probe takes only one downlink symbol, whereas each symbol phase probe takes several downlink symbol periods, including the lSI guard periods.

One issue with the carrier and symbol phase probe method described above is that no measure is made of the amount of noise added by the communication channel, analogue circuitry etc. As the level of noise decreases, the sensitivity of the reported measures increases (but the range of errors measurable, before "clipping", reduces).

This is an issue for many control systems, as it effectively changes the gain of the error measure, which can affect the settling time, damping factor, or other properties of the control system. Therefore, it may be desirable to derive not just an average of the dithered and quantised offset, but also a measure of the noise magnitude. The system can then use this to adjust the gain of the error measure fed to the control system, mitigating these problems.

One technique for deriving the noise measure is to use a method we term double phase probing. We explain this method using carrier phase probes, but it is also readily extendible to symbol phase probes, as will be seen, Figure 4 shows the complex amplitudes of the carrier phase probes for this method. Rather than transmitting probe signals exactly on the decision boundaries between the standard constellation points, as in Figure 2, the probes are placed a small amount either side of, and equidistant to, the decision boundaries. Probes 100, 102, 104 and 106 in Figure 4 are referred to as positive probes, as their phase angle has a positive offset from the decision boundary; and similarly, probes 110, 112, 114 and 116 are referred to as negative probes.

To handle double phase probes, the system of Figure 1 is modified, as follows. A second probe measurement unit is added, in addition to the probe measurement unit 40. One probe measurement unit measures the positive probes, call the measure m; the other measures the negative probes, call the measure m. An equal number of positive and negative probes are inserted by the probe insert unit 222 and transmitted by the BS transmitter 22. The accumulation periods for both positive and negative probes are chosen to be equal and coincident. In normal operation, it is seen that mp> m, due to the phase offsets added to the positive and negative probes shown in Figure 4.

The RS receiver 34 performs the addition m + m, yielding a single measure of phase offset, suitable to be passed into the offset correction controller 47, as before. It can be seen that, due to our formulation of the probe complex amplitudes in Figure 4, the carrier phase offset added to the positive phase probes cancels the carrier phase offset subtracted from the negative phase probes in this addition.

Additionally, the RS receiver 34 performs the subtraction m -m, to give a measure that decreases with increasing noise amplitude. The noise measure mp -m is then used to adjust the gain of the phase offset measure m + m entering the offset correction controller 47, such that, the lower the value of mp -m, the greater the gain applied.

The telecommunication system described so far has used QPSK. However, it can be seen that the system could alternatively be configured to use a lower-order modulation scheme, such as binary phase shift keying (BPSK), by removing the Q quadrature arm of the RS receiver 34. On-off keying (00K) is another possibility, with symbol detection at either the thresholder 52 or 54. Another option is to use a higher-order modulation scheme, such as quadrature amplitude modulation (QAM) or m phase shift keying (m-PSK). In the case of QAM, the simple binary thresholders 52 and 54 must be replaced with multi-level quantizers. These may be ADCs, for example. Advantage is gained, compared to existing QAM receivers, in that lower resolution ADCs may be used, with as few quantization levels as the number of discrete amplitude signalling levels on I and Q. This allows reduction in the size, power-consumption and cost of the ADCs.

Another possibility is to employ offset quadrature modulation, such as offset QPSK (OQPSK) or offset QAM (OQAM). In these schemes, the Q component of the transmitted signal is delayed by half a symbol, yielding a lower peak-to-average-power-ratio (PAPR) signal, which may be advantageous to the transmitter RF circuitry 238.

This may be accommodated at the RS receiver 34 simply by delaying the symbol clock to the thresholder 54 in the quadrature detection arm by half a symbol.

It will be readily apparent to a person skilled in the art that other constellations and modulation schemes may be employed, embodying the same essential inventive ideas.

The telecommunication system of Figure 1 is shown as having just one BS and one RS. However, it can be seen that it is straightforward to extend the ideas presented so that one BS communicates with multiple RSs. Conventional time division multiple access (TDMA) or frequency division multiple access (FDMA) techniques may be used to accommodate signalling that shares a common communication medium between the multiple units. Similarly, the system may be extended to allow multiple OSs to use the communication medium.

The telecommunication system of Figure 1 shows just the sub-systems necessary to enable communication with the simplified RS receiver 34 presented. Clearly, other sub-systems may be added to the BS 20 and RS 30, such as forward error correction (FEC), or a "back channel" to support RS-to-BS communication, which could be used to support automatic repeat request (ARQ), for example.

There is thus described a telecommunication system with simplified receiver, which nevertheless allows for accurate synchronization of the receiver with the transmitted signals.

Claims

CLAIMS1. A method of operation of a communication system, the method comprising: transmitting a signal from a transmitter to a receiver, the signal comprising a carrier having data modulated thereon; and receiving the transmitted signal in the receiver, by means of a local synchronization signal; and the method further comprising: transmitting in said signal a plurality of phase probe signals having a known property namely that, when detected in the absence of noise and with a local synchronization signal having a desired phase, the phase probe signals have zero amplitude in at least one phase; receiving said phase probe signals in the receiver, by means of said local synchronization signal; determining a mean amplitude of the received phase probe signals in said at least one phase; and adjusting a phase of said local synchronization signal based on said determined mean amplitude.
2. A method as claimed in claim 1, wherein the step of determining the mean amplitude of the received phase probe signals in said at least one phase comprises determining whether the mean amplitude of the received phase probe signals in said at least one phase is positive or negative.
3. A method as claimed in claim 1, wherein the local synchronization signal is a local oscillator signal, used to downconvert the received transmitted signal.
4. A method as claimed in claim 3, wherein the phase probe signals comprise a plurality of signals having zero mean amplitude of an in-phase component, and/or a plurality of signals having zero mean amplitude of a quadrature component, and wherein the step of determining the mean amplitude of the received phase probe signals in said at least one phase comprises determining the mean amplitude in the receiver of the signals transmitted with zero mean amplitude.
5. A method as claimed in claim 4, wherein each phase probe symbol comprises a symbol on a decision boundary between two constellation points of the signal transmitted from the transmitter to the receiver.
6. A method as claimed in claim 4 or 5, comprising estimating a waveform synchronization error by accumulating a value for the in-phase or quadrature component of a plurality of carrier phase probe symbols received in the receiver.
7. A method as claimed in claim 3, wherein the phase probe symbols comprise positive carrier phase probe symbols having a positive rotation relative to a decision boundary between two constellation points of the signal transmitted from the transmitter to the receiver, and negative carrier phase probe symbols having a negative rotation relative to said decision boundary.
8. A method as claimed in claim 7, comprising accumulating a positive accumulation value for the in-phase or quadrature component of a plurality of positive carrier phase probe symbols received in the receiver, and separately accumulating a negative accumulation value for the in-phase or quadrature component of a plurality of negative carrier phase probe symbols received in the receiver.
9. A method as claimed in claim 8, comprising estimating the error in the carrier phase of the received signal by summing the positive accumulation and the negative accumulation.
10. A method as claimed in claim 8 or 9, comprising estimating any noise added to the signal transmitted from the transmitter to the receiver by subtracting the positive accumulation and the negative accumulation.
11. A method as claimed in claim 10, further comprising controlling a gain of the adjustment of the phase of said local synchronization signal, based on the estimated noise.
12. A method as claimed in claim 1, wherein the local synchronization signal is a local clock signal, used to determine a sampling point at which a value of the received signals is detected.
13. A method as claimed in claim 12, wherein each phase probe signal comprises first and second symbols, the second symbol being the inverse of the first symbol, the first and second symbols being transmitted equal time intervals before a normal symbol timing point and after a normal symbol timing point respectively.
14. A method as claimed in claim 13, wherein said equal time intervals are equal to one half of a time period between normal symbol timing points.
15. A method as claimed in claim 13 or 14, comprising estimating the waveform synchronization error by accumulating a value for the phase probe signal received in the receiver and sampled at the normal symbol timing point.
16. A method as claimed in claim 12, wherein each phase probe signal comprises first and second symbols, the second symbol being the inverse of the first symbol, the first and second symbols being transmitted at respective normal symbol timing points, the method comprising estimating the waveform synchronization error by accumulating a value for the symbol phase probe signal received in the receiver and sampled at a timing point intermediate between said respective normal symbol timing points.
17. A method as claimed in claim 13 or claim 16, comprising inserting a guard interval, by transmitting no data symbol at at least one normal symbol timing point preceding and following the points at which said first and second symbols are transmitted.
18. A method as claimed in claim 1, wherein local oscillator and local clock signals of the receiver are synchronized to a common clock, the method further comprising transmitting a carrier phase probe signal to allow correction of the local oscillator and a symbol phase probe signal to allow correction of the local clock from the transmitter to the receiver during an initialization period, and thereafter ceasing transmission of one of said probe signals.
19. A method as claimed in claim 18, comprising ceasing transmission of one of said probe signals when an estimated waveform synchronization error has fallen to a predetermined value.
20. A transmitter, for use in a communication system, the transmitter comprising: means for transmitting a signal to a receiver, the signal comprising a carrier having data modulated thereon; and further comprising: means for transmitting in said signal a plurality of phase probe signals having a known property namely that, when detected in the absence of noise and with a local synchronization signal having a desired phase, the phase probe signals have zero amplitude in at least one phase.
21. A transmitter as claimed in claim 20, wherein the phase probe signals comprise a plurality of signals having zero mean amplitude of an in-phase component, and/or a plurality of signals having zero mean amplitude of a quadrature component.
22. A transmitter as claimed in claim 21, wherein each phase probe symbol comprises a symbol on a decision boundary between two constellation points of the signal transmitted from the transmitter to the receiver.
23. A transmitter as claimed in claim 21, wherein the phase probe symbols comprise positive carrier phase probe symbols having a positive rotation relative to a decision boundary between two constellation points of the signal transmitted from the transmitter to the receiver, and negative carrier phase probe symbols having a negative rotation relative to said decision boundary.
24. A transmitter as claimed in claim 20, wherein each phase probe signal comprises first and second symbols, the second symbol being the inverse of the first symbol, the first and second symbols being transmitted equal time intervals before a normal symbol timing point and after a normal symbol timing point respectively.
25. A transmitter as claimed in claim 24, wherein said equal time intervals are equal to one half of a time period between normal symbol timing points.
26. A transmitter as claimed in claim 20, wherein each phase probe signal comprises first and second symbols, the second symbol being the inverse of the first symbol, the first and second symbols being transmitted at respective normal symbol timing points, the method comprising estimating the waveform synchronization error by accumulating a value for the symbol phase probe signal received in the receiver and sampled at a timing point intermediate between said respective normal symbol timing points.
27. A transmitter as claimed in claim 24 or claim 26, further comprising means for inserting a guard interval, by transmitting no data symbol at at least one normal symbol timing point preceding and following the points at which said first and second symbols are transmitted.
28. A receiver, for use in a communication system, the receiver comprising: means for receiving a transmitted signal, by means of a local synchronization signal, said transmitted signal comprising a carrier having data modulated thereon; and further comprising: means for receiving phase probe signals in the receiver, by means of said local synchronization signal, said phase probe signals having a known property namely that, when detected in the absence of noise and with a local synchronization signal having a desired phase, the phase probe signals have zero amplitude in at least one phase; means for determining a mean amplitude of the received phase probe signals in said at least one phase; and means for adjusting a phase of said local synchronization signal based on said determined mean amplitude.
29. A receiver as claimed in claim 28, wherein the means for determining the mean amplitude of the received phase probe signals in said at least one phase comprises means for determining whether the mean amplitude of the received phase probe signals in said at least one phase is positive or negative.
30. A receiver as claimed in claim 28, wherein the local synchronization signal is a local oscillator signal, used to downconvert the received transmitted signal.
31. A receiver as claimed in claim 28, wherein the phase probe signals comprise a plurality of signals having zero mean amplitude of an in-phase component, and/or a plurality of signals having zero mean amplitude of a quadrature component, and the means for determining the mean amplitude of the received phase probe signals in said at least one phase comprises means for determining the mean amplitude in the receiver of the signals transmitted with zero mean amplitude.
32. A receiver as claimed in claim 31, comprising means for estimating a waveform synchronization error by accumulating a value for the in-phase or quadrature component of a plurality of carrier phase probe symbols received in the receiver.
33. A receiver as claimed in claim 32, comprising means for accumulating a positive accumulation value for the in-phase or quadrature component of a plurality of positive carrier phase probe symbols received in the receiver, and separately accumulating a negative accumulation value for the in-phase or quadrature component of a plurality of negative carrier phase probe symbols received in the receiver.
34. A receiver as claimed in claim 33, comprising means for estimating the error in the carrier phase of the received signal by summing the positive accumulation and the negative accumulation.
35. A receiver as claimed in claim 33 or 34, comprising means for estimating any noise added to the signal transmitted from the transmitter to the receiver by subtracting the positive accumulation and the negative accumulation.
36. A receiver as claimed in claim 35, comprising means for controlling a gain of the adjustment of the phase of said local synchronization signal, based on the estimated noise.
37. A receiver as claimed in claim 28, wherein the local synchronization signal is a local clock signal, used to determine a sampling point at which a value of the received signals is detected.
38. A receiver as claimed in claim 37, comprising means for estimating the waveform synchronization error by accumulating a value for the phase probe signal received in the receiver and sampled at the normal symbol timing point.
39. A receiver as claimed in claim 38, wherein each phase probe signal comprises first and second symbols, the second symbol being the inverse of the first symbol, the first and second symbols being transmitted at respective normal symbol timing points, the receiver comprising means for estimating the waveform synchronization error by accumulating a value for the symbol phase probe signal received in the receiver and sampled at a tlmhg point IntermedIate between said respective normal symbol tining point
40. An audio device, comprising a receiver as claimed in any of claims 28 to 39.