GB2382233A

GB2382233A - Hybrid filter

Info

Publication number: GB2382233A
Application number: GB0119298A
Authority: GB
Inventors: Keith Rushworth Williams
Original assignee: Koninklijke Philips Electronics NV
Current assignee: Koninklijke Philips NV
Priority date: 2001-08-08
Filing date: 2001-08-08
Publication date: 2003-05-21
Also published as: GB0119298D0

Abstract

A hybrid filter comprises an array of resonators (78 to 90) arranged such that at least the outermost resonators (78, 90) are edge coupled to the next to outermost resonators (82, 90) and the inner resonators (82 to 90) are end coupled. The hybrid filter has particular, but not exclusive, application as an image reject filter in mm-wave frequencies receivers and transmitters.

Description

DESCRIPTION HYBRID FILTER The present invention relates to a hybrid filter and has particular, but not exclusive, application to hybrid microstrip edge-end coupled line filters. The hybrid filter is usable for image rejection in mm-wave frequencies receivers and transmitters.

Many different types of planar coupled line bandpass filters are widely used in high frequency equipment. These designs are mainly based on stripline or microstrip transmission line technology and details of the various designs together with design methods can be found in"Microwave Filters, Impedance-Matching Networks, and Coupling Structures"by Matthaei, Young and Jones (Artech House 1980, ISBN 0-89006-099-1).

US Patent Specification 6,067, 461 discloses some of these filter types including an edge coupled filter, an interdigital filter, and an end (or gap) coupled filter. Other known types of filter are the hairpin filter and the combine filter. Figures 1 to 5 of the accompanying drawings respectively show diagrammatic representations of a half wave edge coupled filter (Figure 1), an interdigital filter (Figure 2), a hairpin filter (Figure 3), a combine filter (Figure 4) and an end (or gap) coupled filter (Figure 5). These types of filters can be constructed on a wide variety of substrate materials ranging from air (suspended stripline), PTFE based materials, alumina, GaAs and very high dielectric materials such as barium titanite and related compounds. High temperature superconducting materials are also used to obtain very low loss performance.

Referring to Figure 1, the edge coupled filter 10 comprises a substrate 12 which has a conductive ground plane on the entire extent of an underside surface (not shown) and an array of spatially separated, substantially parallel electrically conductive elements forming resonators 14,16, 18,20, 22,24 and 26 on a planar, top surface. Adjacent conductors partially overlap each other

lengthwise to form coupled sections in which electrical coupling is effected through the edges of the resonators. The widths S of the coupled sections vary from being the greatest for the innermost coupled section (s) to the smallest for the outermost coupled sections. In the case of the innermost coupled sections, the width S becomes comparable to its length L. At high frequencies, that is frequencies greater than 20 GHz, this width/length relationship leads to uncertainty in the level of coupling and spurious coupling between adjacent coupled sections.

The interdigital filter 28 shown in Figure 2 has a different arrangement of resonators to that shown in Figure 1. The illustrated interdigital filter has five parallel arranged, equally spaced, rectilinear resonators 30,32, 34,36 and 38 on the top surface of the substrate 12. Direct in/out coupling 40 and 42 are made at right angles to the outer resonators 30,38, respectively. Opposite ends of adjacent resonators are grounded. In practice the provision of satisfactory grounding of the resonators 30 to 38 has been found to be difficult at high frequencies.

The embodiment of the hairpin filter 44 shown in Figure 3 comprises five alternately arranged folded or U-shaped resonators 46, 48,50, 52 and 54.

The limbs of the folded resonators are substantially equally spaced and parallel. Direct in/out coupling 40,42 are made at right angles to the outer limbs of the folded resonators 46,54. At high frequencies the relatively large coupling gaps are problematic because of the short resonator length. It is difficult to provide adequate spacing between the two limbs of the folded resonators 46 to 54 to give the required coupling.

The combine filter 56 shown in Figure 4 comprises substantially the same layout of resonators 30,32, 34,36, 38 as in Figure 2 and in the interests of brevity only the differences will be described. A corresponding end of each resonator is connected directly to ground whilst the other ends of these resonators 30 to 38 are coupled to ground by way of respective capacitors C1 to C5. In practice the potential problems of the combine filter are the provision of a satisfactory grounding and the accurate realisation of the capacitors C1 to C5.

Finally, the end (or gap) coupled filter 60 shown in Figure 5 comprises a substrate 12 having an electrically conductive ground plane (not shown) on its underside and a linear array of five elongate resonators 64,66, 68,70, 72 together with input and output connections 62,74 on the top side of the substrate. The innermost resonator 68 has the greatest length and the lengths of the other resonators decreases towards the ends of the linear array.

Adjacent ends of pairs of resonators 64 to 72 and of input and output connections 62,74 and the resonators 64 and 72 are separated by gaps G2, G3, G4, G5, G1 and G6, respectively. The widths of the gaps varies outwardly from the resonator 68 with the gaps G3, G4 having the greatest width and the gaps G 1, G6 being the smallest. A limitation of the gap coupled filter is the need for a high degree of coupling for the gaps G1 and G6 at the input and output of the filter. This results in very small gaps being required which are very difficult or impossible to realise in practice because of the limitations of the processes used to fabricate the filters.

An object of the present invention is to provide hybrid filters for use at mm-wave frequencies which have a predictable performance and are capable of being manufactured with current technology.

According to one aspect of the present invention there is provided a hybrid filter comprising edge coupled outer resonators and at least two end coupled inner resonators.

The present invention also provides a hybrid filter comprising a planar array of a plurality of resonators, said resonators being arranged so that at least 2 inner resonators of the array are end coupled and the remainder of the resonators are edge coupled.

The present invention further provides a hybrid microstrip filter comprising a dielectric substrate having a plurality of resonators on a surface of the substrate, said resonators being arranged so that outer resonators are edge coupled and inner resonators are end coupled.

The present invention is based on a realisation that in an arrangement of seemingly incompatible elements edge coupled resonators have a

frequency response which repeats at odd harmonics of the passband and the end or gap coupled resonators have a response which repeats at all harmonics of the passband.

The combination of edge and gap coupling enables desired attributes to be achieved in that where high levels of coupling are required, edge coupled resonators are used and where lower levels of coupling are necessary, gap coupling is used.

The use of edge coupling for the outer section of the filter results in coupling values in the range 5 to 15 dB being achievable with coupler dimensions which can be readily achieved in manufacture. The gap coupling used in the internal part of the filter allows coupling values in the range 15 to 30 dB to be achieved at frequencies greater than 20 GHz, again with manufacturable dimensions. This range of available coupling values allows most narrowband (less than 10% bandwidth) filter requirements to be met with , this design.

Other beneficial features of this type of filter for high frequency applications include the fact that edge coupled sections are separated from each other by a relatively large distance thereby minimising spurious coupling between these sections. The electric field in the areas around the gaps is contained in a small area and is well separated from adjacent gaps, again minimising spurious coupling.

The physical width of the filter is small. In order to achieve good isolation performance, it is necessary to enclose the filter with walls and a cover, which form a waveguide below cut-off. As frequencies increase, spacing between the walls enclosing the filter must be reduced to ensure the waveguide formed by these walls remains below cut-off. The proposed filter is well suited to this requirement. Many of those described with reference to Figures 1 to 5 of the accompanying drawings either cannot fulfil this requirement or must be folded or otherwise distorted to be made sufficiently narrow.

According to a second aspect of the present invention there is provided a mm-wave frequencies transmitter including an image rejection filter

comprising at least one hybrid filter made in accordance with the first aspect of the present invention.

According to a third aspect of the present invention there is provided a mm-wave frequencies receiver including an image rejection filter comprising at least one hybrid filter made in accordance with the first aspect of the present invention.

According to a fourth aspect of the present invention there is provided a monolithic circuit comprising at least two circuit stages coupled in series, at least one of the stages comprising a hybrid filter made in accordance with the first aspect of the present invention.

The present invention will now be explained and described, by way of example, with reference to the accompanying drawings, wherein: Figures 1 to 5 respectively show known types of edge-coupled filter, interdigital filter, hairpin filter, combine filter and gap-coupled filter, Figure 6 shows a first embodiment of a hybrid filter made in accordance with the present invention, Figure 7 is a flow chart giving one embodiment of the design processes in making a hybrid filter made in accordance with the present invention, Figure 8 are graphs of the measured and predicted performance results for a specimen filter, Figure 9 shows a second embodiment of a hybrid filter made in accordance with the present invention having a plurality of edge coupled sections, Figure 10 shows a third embodiment of a hybrid filter made in accordance with the present invention having folded filter resonators, Figure 11 shows a variant of the third embodiment of a hybrid filter made in accordance with the present invention, Figure 12 illustrates how gap coupled resonators may be modified to increase the coupling across a gap,

Figure 13 is a block schematic diagram of the front end of a mm-wave frequency receiver including at least one hybrid filter made in accordance with the present invention, and Figure 14 is a block schematic diagram of the output stage of a mmwave frequency transmitter including at least one hybrid filter made in accordance with the present invention.

In the drawings the same reference numerals have been used to illustrate corresponding features.

As Figures 1 to 5 have been described in detail in the preamble they will not be described again.

Referring to Figure 6, the filter shown is a mm-wave frequencies hybrid bandpass filter 76 comprising a substrate 12 of a suitable material including air (suspended stripline), PTFE based materials, alumina, quartz, GaAs and very high dielectric materials such as barium titanite and related compounds.

Additionally high temperature superconducting materials may be used to obtain very low loss performance. On the underside of the substrate 12 is provided an electrically conductive layer (not shown) which forms the ground plane. On the top surface of the substrate 12 there is provided an array of resonators formed of elongate electrically conductive elements 78,80, 82,84, 86,88 and 90. The conductive elements can be provided by any suitable metallisation technologies such as thin film, thick film, copper and so forth and transmission line options (microstrip, stripline and so on) could also be used to realise the illustrated filter.

The inner resonators 82 to 90 are disposed in line end-to-end with gaps G1, G2, G3, G4, respectively, between the resonators 82,84 ; 84,86 ; 86,88 and 88,90 to provide gap (or end) coupling. The inner gaps G2, G3 are nominally of the same width and larger than the gaps G1, G4 which are nominally of the same width. The resonators 78,80 are disposed to one side of, and co-extend partially with, the resonators 82, 90, respectively. The coextensive portions of the resonators 78,82 and 80,90 are separated by gaps EG1 and EG2, respectively. These coextensive portions provide edge coupling

for the outer section of the filter and enable high levels of coupling to be provided. Edge coupling values in the range 5 to 15 dB can be achieved with manufacturable dimensions. The gaps G1 to G4 provide gap coupling in the inner part of the filter and allow coupling values in the range 15 to 30dB to be achieved at frequencies greater than 20 GHz, again with manufacturable dimensions. This range of coupling values enables most narrowband (less than 10% bandwidth) filter requirements to be met with the architecture shown in Figure 6.

Although no explicit design synthesis technique for the hybrid filter shown in Figure 6 exists, the flow chart shown in Figure 7 indicates a method of designing the filter making use of the design techniques for an edgecoupled filter.

Referring to Figure 7, block 92 relates to synthesising a design for an edge coupled filter to the required specification using the techniques described in the above-mentioned"Microwave Filters, Impedance-Matching Networks, and Coupling Structures" or similar. Block 94 relates to analysing the resulting design using high frequency CAD simulation software and optimising the dimensions of the filter to achieve the required performance. Block 96 relates to calculating the coupling value and phase length of the inner coupled line sections of the resulting filter using appropriate coupled line and open circuit models available in the literature. Block 98 relates to calculating the gap dimensions necessary to achieve the equivalent levels of coupling found in block 96 using the appropriate gap model from the literature, and to calculate the phase shift across the gap. Block 100 relates to, for each gap, calculating the lengths of transmission line needed on either side of the respective gaps found in block 98 using standard models for the appropriate transmission medium, to achieve an overall phase length for the gap and transmission lines equal to the phase lengths calculated in block 96. Block 102 relates to completing the determination of all the dimensions of the hybrid filter and analysing the new filter using any high frequency CAD simulator. If necessary, optimisation of the new design can be undertaken by the CAD software to correct for minor errors resulting from the process above. Block 104 relates to

a final refinement in which optionally the filter can also be analysed using 2D or 3D electromagnetic simulation to obtain a more accurate prediction of performance, since the models used in commercial CAD simulators tend to be less accurate at high frequencies.

It should be noted that the calculations required in blocks 96,98 and 100 need only be performed at the filter passband centre frequency in order to determine the required hybrid filter dimensions.

After completion of the design, block 106 relates to fabricating the filter on the appropriate substrate material using standard fabrication techniques. In view of the high frequencies and hence dimensional accuracy required for parts of the filter, the optimum manufacturing technology is considered to be thin film processing in conjunction with a hard (ceramic, quartz or similar) substrate.

Figure 8 shows the measured and predicted performance results of a bandpass filter of a type shown in Figure 6. The design was targeted at frequencies around 40 GHz to demonstrate the validity of the filter well into the mm-wave frequency range. In Figure 8 the curves 110 and 112 are respectively the measured and predicted results for frequency (GHz) versus insertion loss (dB) and the curves 114 and 116 are respectively the measured and predicted results for frequency (GHz) versus return loss (dB).

The results demonstrated in Fig 8 were achieved with a microstrip filter similar to that shown in Figure 6 having the following characteristics

Substrate material 99% alumina (AI203) Substrate thickness 0.254mm (0.010") Substrate metallisation seed layer NiCr, resistivity 500/square Substrate metallisation conductor layer Au, 0. 3/im sputtered Au, 2-3, um plated Substrate size 8.0 mm x 2.7mm Cover size 8.0 mm x 3. 0mm x 2.5 mm (LxWxH) Pattern dimensions

All conductor tracks were 260Jim wide, corresponding to 500 characteristic impedance.

The filter is symmetrical about the centre of the pattern and so dimensions are given for one half of the filter working inwards from the left hand side of Figure 6.

The edge coupled line section is 708Jim long with a coupling gap of 52Jim.

The length of track from the end of the edge coupled section to the first gap coupling is 566/im, and the length of the first gap is 120jim.

The length of the track from the first gap to the second gap is 1266jim and the length of the second gap is 158, um.

Finally, the total length of the central track is 1268Jim.

In order to make up the total length for the filter of 8. 0mm, connection tracks of 500 impedance are included.

For testing, the alumina substrate was mounted into a test circuit consisting of 500 microstrip input and output lines fabricated on a Taconic TLY 5 substrate with 0.254mm thickness, Y2 oz. (17 Jim) copper cladding on the top face and 1.5 mm thick aluminium cladding on the back face. Connection between the Taconic and alumina tracks was made using 25 Jim diameter gold wire. A total of five wires were bonded in parallel at each Taconic to alumina junction to minimise inductance.

The complete test circuit was fitted into an Argumens adjustable test fixture with Anritsu coaxial K connectors and measurements were made on an Anritsu 37377A vector network analyser calibrated with an Anritsu 3652K calibration kit.

Measurement of a 500 through line on this system indicated a loss for the test fixture and Taconic input and output lines of approximately 1.5dB and Figure 8 does not include any correction for this loss. If this loss is deducted from the measured performance shown in Figure 8, then the passband insertion loss of the alumina part of the filter would be approximately 2dB rather than the 3.5dB seen in Figure 8.

Figure 9 shows a second embodiment of a filter 120 made in accordance with the present invention. This embodiment differs from that shown in Figure 6 in that edge coupling is used for both the outermost couplers, that is resonators 122, 124 and 132,134, and the next to outermost couplers, that is resonators 124,126 and 130,132. gap coupling is used between a central resonator 128 and the resonators 126 and 130. The width of the outer edge coupler is less than that of the next to outermost coupler.

As a generality, depending on the number of sections in the filter, there could be more edge coupled sections used provided that as a minimum at least one of the coupling is made using a gap structure. This approach would permit higher levels of coupling in the internal sections of the filter 120 and could be used to increase the bandwidth available with this type of filter to more than 20%.

In the third embodiment of the invention shown in Figure 10, filter resonators 142,144, 146, 148, 150 and 152 have been folded to reduce the length of the filter at the expense of greater width. Gap coupling exists between adjacent pairs of these resonators. Resonators 138,140 and 154,156 are edge coupled and a gap coupling exists between the resonators 140,142 and 152,154, respectively. The use of folded resonators is useful at lower frequencies where the width of the filter is less important but the length given by the layout in Figure 6 would be excessive.

Figure 11 shows a variant of the third embodiment in which input and output connections 158,160 of the filter 136 are on the side edges of the substrate 12 instead of the top edge as shown in Figure 10.

Further variants of the filter layouts shown in Figures 10 and 11 are possible with different combinations of input and output positions on all edges of the substrate 12.

In all the illustrations of the filter made in accordance with the present invention the widths of all the transmission lines making up the filters and feedlines are identical. This is not a requirement of the filter operation, and different linewidths for all the sections of the filter may be used. Usually, the input and output feedlines connecting to the filter have a width determined by

the characteristic impedance of the system in which the filter is being used, usually 50 ohms.

Since the filter is a passive reciprocal network, conventionally, the elements of the filter are usually made symmetrical about the centre line of the filter, and this is the case with the filters shown in Figures 6,9, 10, 11. This is not a fixed requirement and filters could be made without symmetry.

In order to allow for a greater degree of coupling across a gap to be achieved the width of the transmission lines in the vicinity of the coupling gaps is increased by enlarging the ends 162 of the resonators R as shown in Fig 12.

This option may not be used in the filters made in accordance with the present invention, since if higher coupling levels were required then these would be achieved with edge coupled sections. However if for any reason it was necessary to achieve higher levels of coupling with a gap, then the same technique could be applied to the new filter.

Referring to Figure 13, the front end of the receiver comprises an antenna 170 coupled to a low noise amplifier 172 which amplifies the mm-wave frequency signals received at the antenna. A bandpass hybrid filter 174 of the type shown in Figures 6 and 9 to 12 passes the wanted frequency band but rejects the image frequencies. The signals in the wanted frequency band are applied to a mixer 176 in which they are frequency down-converted using a local oscillator signal to produce a lower frequency signal which is demodulated in a demodulator 178.

The basic local oscillator signal is generated by an oscillator 180 and the final local oscillator signal is produced by frequency multiplication in a frequency multiplier 182. The output from the frequency multiplier 182 comprises the wanted local oscillator signal plus unwanted harmonics of the local oscillator frequency. A hybrid filter of the type shown in Figures 6 and 9 to 12 suppresses these unwanted harmonics and the filtered signal is supplied as the local oscillator to the mixer 176.

Typically the individual stages of the receiver front end are fabricated as individual monlithic circuits but the low noise amplifier 172, the hybrid filter 174 and the mixer 176 can be fabricated as a single monolithic circuit 177 on any suitable substrate, for example GaAs.

The output stage of the mm-wave frequency transmitter shown in Figure 14 is substantially the converse of the receiver shown in Figure 13 and where appropriate the corresponding stages have been referenced using the same reference numerals. A signal from a modulator is frequency up-converted to its final output frequency in a mixer 176 using a locally generated master oscillator signal. This master oscillator signal is produced by an oscillator 180 whose output frequency is multiplied in a frequency multiplier 182 after which unwanted harmonics are suppressed in a hybrid filter 184 of the type shown in Figures 6 and 9 to 12.

The frequency up-converted signal is filtered in another hybrid filter 174 of the type shown in Figures 6 and 9 to 12 in order to reject image and other spurious signals. The output of the hybrid filter is supplied to a power amplifier 188 for amplification prior to being applied to an antenna 170.

Optionally in the receiver and the transmitter more than one hybrid filter may be used in order to increase signal rejection.

In the present specification and claims the word"a"or"an"preceding an element does not exclude the presence of a plurality of such elements. Further, the word"comprising"does not exclude the presence of other elements or steps than those listed.

From reading the present disclosure, other modifications will be apparent to persons skilled in the art. Such modifications may involve other features which are already known in the design, manufacture and use of hybrid filters and component parts therefor and which may be used instead of or in addition to features already described herein. Although claims have been formulated in this application to particular combinations of features, it should be understood that the scope of the disclosure of the present application also includes any novel feature or any novel combination of features disclosed herein either explicitly or implicitly or any generalisation thereof, whether or not it relates to the same invention as presently claimed in any claim and whether or not it mitigates any or all of the same technical problems as does the present invention. The applicants hereby give notice that new claims may be formulated to such features and/or combinations of such features during the

prosecution of the present application or of any further application derived therefrom.

Claims

1. A hybrid filter comprising edge coupled outer resonators and at least two end coupled inner resonators.

2. A hybrid filter comprising a planar array of a plurality of resonators, said resonators being arranged so that at least 2 inner resonators of the array are end coupled and the remainder of the resonators are edge coupled.

3. A hybrid microstrip filter comprising a dielectric substrate having a plurality of resonators on a surface of the substrate, said resonators being arranged so that outer resonators are edge coupled and inner resonators are end coupled.

4. A filter as claimed in claim 1, 2 or 3, characterised in that a plurality of the resonators are edge coupled and in that the width (s) of the edge coupled sections increases inwardly from the outer coupled section.

5. A filter as claimed in any one of claims 1 to 4, characterised in that the lengths of the inner resonators are greater than that of the outer resonators.

6. A filter as claimed in any one of claims 1 to 6, characterised in that end coupling gaps between resonators decreases from the innermost resonator (s) outwards.

7. A filter as claimed in any one of claims 1 to 6, characterised in that the resonators are symmetrically arranged.

8. A filter as claimed in any one of claims 1 to 7, characterised in that said resonators are of substantially rectilinear shape.

9. A filter as claimed in any one of claims 1 to 7, characterised in that at least the end coupled resonators are non-rectilinear.

10. A filter as claimed in any one of claims 1 to 9, characterised by conductive in/out connections to the resonators.

11. A filter as claimed in any one of claims 1 to 10, characterised in that at least the ends of the end coupled resonators are enlarged to increase the degree of coupling.

12. A filter as claimed in any one of claims 1 to 11, characterised in that the widths of the resonators being the same.

13. A mm-wave frequencies transmitter including an image rejection filter comprising at least one hybrid filter as claimed in any one of claims 1 to 12.

14. A mm-wave frequencies receiver including an image rejection filter comprising at least one hybrid filter as claimed in any one of claims 1 to 12.

15. A monolithic circuit comprising at least two circuit stages coupled in series, at least one of the stages comprising a hybrid filter as claimed in any one of claims 1 to 12.

16. A hybrid filter constructed and arranged to operate substantially as hereinbefore described with reference to and as shown in the accompanying drawings.

17. A mm-wave frequencies transmitter constructed and arranged to operate substantially as hereinbefore described with reference to and as shown in the accompanying drawings.

18. A mm-wave frequencies receiver constructed and arranged to operate substantially as hereinbefore described with reference to and as shown in the accompanying drawings.