GB1584641A - Frequency-modulated data communications receivers - Google Patents

Abstract

This device is suitable for use under propagation conditions severely affected by interference, in particular in mobile stations, in which propagation delay differences occur at the receive location due to the alternate routing differences of the reflected carrier waves. The information losses caused by phase and amplitude distortion are thereby automatically identified according to their cause in two arrangements which complement one another, i.e. in an equaliser for dynamic cancellation and an equaliser for static cancellation. The equaliser for dynamic cancellation has a frequency discriminator (3) with a device connected downstream for noise peak detection (DE) and the equaliser for static cancellation (SE) has an amplitude demodulator (9) with a polarity inverter (11) connected downstream and controlled by a polarity integrator (12). Depending on the type of interference detected, one or the other equaliser is through-connected on the output side via a changeover switch (13) to the common output. <IMAGE>

Description

(54)IMPROVEMENTS IN OR RELATING TO FREQUENCY-MODULATED DATA COMMUNICATIONS RECEIVERS (71) We, SIEMENS AKTIENGESELLSCHAFT, a German Company, of Berlin and Munich, Federal Republic of Germany, do hereby declare the invention, for which we pray that a patent may be granted to us, and the method by which it is to be performed, to be particularly described in and by the following statement: The invention relates to frequencymodulated data communications receivers, for use in systems where digital signals are impressed in the form of a frequency modulation upon a carrier and transmitted over a propagation path subject to reflection, and in particular to receivers for use in mobile stations, long-distance traffic links and scatter systems.

In digital communications transmissions systems, heavily disturbed propagation conditions, such as multipath propagation, cause the operating range to vary in an approximately inversely proportional manner to the magnitude of the bit rate that is to be transmitted. The critical situation which determines the range is represented by total information extinction, where, as a result of the differences in transmit time produced by the path-length differences of reflected carrier waves, the modulation signals arrive in phase opposition at the location of reception, and therefore mutually extinguish one another. In a wide range preceding this critical situation, partial information lossess occur due to transit time distortions and amplitude distortions which lead to very high fault rates in the transmission.

One object of the present invention is to effect a considerable improvement in the transmission quality for the last-mentioned situation, i.e. in effect to achieve an improvement in the range of digital communications systems exhibiting frequency modulation, in particular between mobile stations and under constantly changing propagation conditions.

The invention consists in a frequency modulated data communications receiver, for use in a system where digital signals are impressed onto a carrier for propagation, in which receiver means are provided to automatically prevent any information loss caused by phase or amplitude distortion in two mutually supplementary arrangements, one in the form of a dynamic distortion corrector comprising a frequency discriminator followed by means for the recognition of interference peaks together with sample and hold circuit operable to lop any said interference peaks, and the other supplementary arrangement in the form of a static distortion corrector comprising an amplitude demodulator connected in parallel with said frequency discriminator arrangement, and the outputs of said demodulator and said discriminator being connected to respective inputs of a change-over switching circuit which is controlled by an amplitude modulation-analysis device and which acts when the amplitude modulation level exceeds a given value to switch the amplitude demodulator to a common output which is otherwise fed by the frequency discriminator, the output of the AMdemodulator being followed by a polarisation inverter which operates under the control of a polarity integrator to reverse the AM-demodulation product in dependence on the magnitude of the FM-demodulation product, in the sense to give polarity-correct AM-demodulation.

The invention will now be described with reference to the drawings, in which: Figure 1 is a first explanatory graph; Figure 2 is a set of explanatory waveforms: Figure 3 is a further set of explanatory waveforms; Figure 4 is an explanatory block schematic diagram; Figure 5 is a more detailed explanatory block schematic diagram; Figure 6 is a block schematic diagram of the relevant parts of one exemplary embodiment; Figure 7 is an explanatory detail; and Figures 8 to 11 are further explanatory waveforms.

Figure 12 is a simplified block schematic circuit diagram giving details of one exemplary construction of an dynamic distortion corrector for use in the embodiment of the invention described with reference to Figure 6; Figure 13 is a set of explanatory waveform diagrams relating to the circuit shown in Fig ure-12; Figure 14 is a simplified block schematic circuit diagram of one exemplary construction of a static distortion corrector for use in the embodiment of the invention described with reference to Figure 6, together with its associated units; Figure 15 is a set of explanatory waveform diagrams relating to Figure 14; Figure 16 is a simplified block schematic circuit diagram of a preferred input circuit for the static distortion corrector of Figure 6 or Figure 14; and Figure 17 is a set of explanatory waveform diagrams relating to Figure 16.

In radio systems, in dependence upon the topographic factors, multipath wave propagations may occur, and particularly with mobile systems, transmission of digital, frequency modulated data employing omnidirectional antennae, this may lead to serious reception disturbances under certain circumstances. In this case the successive wave fronts emitted from the transmitting antenna each arrive at the receiving antenna via different paths with different transit times, as a result of reflections from mutually different directions. As a result of the vectorial addition of the received wave fronts of any transmitted wave as they arrive at the location of reception, the antenna voltage exhibits amplitude and phase fluctuations because thc response is dependent both upon frequency and location (minima and maxima).The resulting distortions and energy reductions (minima) mean that the consequential energy distribution causes a loss of ability to read out digital reception signals at many frequencies in many locations.

In order to provide a fundamental explanation of the distortions caused by multipath propagation, it is expedient to firstly consider fixed location points for the transmitter and the receiver. This allows any location dependent energy variations to be disregarded, and only consider the frcquency-dependent energy distribution.

As soon as the differences in transit times of the wave fronts of a direct beam Ud and a by-pass beam Uu received at the reception location have a magnitude of an order approaching that of the bit duration (approximately At =0.1 to 0.7 x tbit), the frequency spacing between the minima of the distribution characteristic becomes so small that the energy of the reception signal may fluctuate almost arbitrarily within the modulation range with the modulation speed and in dependence upon the radio frequency oo and the depth of the respective minima.

A predominant result of these energy fluctuations, which are produced by the vectorial addition of the oncoming signals and which can be eliminated again in an amplitude limiter of the receiver system prior to demodulation, is that there are rapid phase changes in the resultant signal, which inevitably occur during the vectorial addition. Of course, it is not possible to suppress these rapid phase changes by means of the amplitude limiter, and consequently they produce a bitsynchronous interference modulation at the output of the provided FM-demodulator.

The extent of this interference modulation can be a multiple of the significant modulation, and thus prevents read-out of the useful modulation.

The maximum phase change rate of the resultant vector occurs in the minima of the distribution characteristic and is the greater, the deeper a minimum is. In the critical situation, with selective total extinction, it can be of any arbitrary value.

As will now be explained, in dependence upon whether the minimum is located within that deviation range defined by the two angular frequencies, or is located outside of the latter, two interference situations Case A or Case B occur, which exhibit characteristic differences: Case A:- Minimum lying outside the deviation range When the distribution characteristic minimum lies outside the deviation range, but in the vicinity of one of the two angular frequencies, the reception energy will be relatively low at this angular frequency, whereas the receiption energy at the second angular frequency must inevitably now be higher, since it lies closer to the next maximum. As a result of this ratio, prior to the limiter, the reception signal exhibits a clearly defined, bit- synchronous amplitude modulation, the polarity of which is either identical to or inverted to the original modulation signal, in dependence upon the postion of the minimum. The limitation, which normally takes place prior to frequency modulation, suppresses this amplitude modulation, and consequently it is not effective at the output of the demodulator. On the other hand, the phase change which occurs in the vicinity of the minimum in the event of a signal change, and which is manifest as a heavy signal distortion at the output of the demodulator, is effective.

An extreme critical situation for this operating behaviour pattern is achieved as soon as the energy undershoots the selfgenerated noise in the receiver at one of the angular frequencies. This is frequently the case when a radio system is operated in the vicinity of its critical sensitivity, or the minimum lies directly on the angular frequency and is very low (selective total extinction). As a result of the negative signal-tonoise ratio at one of the angular frequencies, instead of all the logic symbols corresponding to this angular frequency (zeros or ones), merely noise occurs at the output of the limiter and demodulator combination. The signal which has been demodulated by the FM-demodulator has thus become unserviceable. However, even in this case the reception signal exhibits a bit-synchronous amplitude modulation prior to the limiter stage.

The duration of the state of noise at the demodulator output in each case corresponds to the character sequence of the modulation data flow. As, during a character which lasts for longer than one bit and which has the same content (zero or one), there is no change in the receiving frequency, and therefore the same frequency arrives at the location of reception via all by-pass routes, consequently this state is retained unchanged until the next character change, and this situation is referred to as "static". Signal losses due to this hypothetical postulate, are referred to hereinafter as "static extinctions.".

Case B:- Minimum lying inside the deviation range The state at which the minimum lies at the angular frequency is defined as static extinction. This definition also applies when the extinction point lies within the deviation range, but is only close to the angular frequency, since the deviation alteration speed of the soft keying which is usually employed2 for reasons of frequency economy (cos -transition) is very low in the vicinity of the angular frequencies. However, as soon as the minimum noticeably approaches the centre of the deviation range, the conditions change as follows: (i) The rate of phase change in the minimum becomes very high. The resultant instantaneous frequency displacement at the limiter output and demodulator output likewise becomes very considerable, and reaches a multiple of the useful deviation range.The duration of the displacement is dependent upon the modulation speed and the relative depth of the minimum. As this correlationship means that the duration of the displacement must always be shorter than the bit duration, the displacement within a modulation character (bit) is manifest as a peak whose size and definition are dependent upon the depth of the minimum.

More than one peak can occur within one character, which is a normal situation with a low modulation index, namely a maximum of 2 peaks of opposite polarity per bit.

However, the distortion peaks do not inevitably occur within each individual bit, but only in the event of a character change, as it is only in this situation that the deviation range is passed through. For this reason these distortions of the demodulated output signal are referred to as "dynamic distortions".

(ii) As soon as a detected minimum noticeably approaches the middle frequency, the definition of the bit-synchronous amplitude modulation prior to the limiter is lost.

In a receiver constructed in accordance with the invention, it is now proposed to avoid the loss of the possibility of reading out the reception signal, which occurs as a result of static extinction, by employing the bitsynchronous amplitude modulation which occurs prior to the limiter in an arrangement suitable for this purpose, forming a static distortion corrector, and further to avoid the loss of the read-out facilities which occurs as a result of the dynamic extinction, by blanking the peaks in an arrangement which is suitable for this purpose, using a dynamic distortion corrector, both of which devices will be described in the following.

Figure 1 illustrates schematically three individual, clearly defined situations I to III.

Firstly we shall discuss situation I, in which the resultant vector of the receiption signal Ures passes through a minimum at the middle radio frequency fin, and is therefore approximately equal in magnitude at the two angular deviation frequenciesfo andfi. As soon as the instantaneous frequency f approaches the middle frequency fin, in addition to the retrogression of the resultant amplitude Ures, there is inevitably an associated phase jump in the roation (preys. This relatively short phase jump, which occurs within the modulation spectrum, must inevitably manifest itself as an instantaneous frequency displacement (dcp/dt) or corresponding transit time distortion (dsso/df), and signifies an interference function which is superimposed upon the original modulation function and which occurs with every digital character change. A digital character change in each case passes through the entire deviation range.

The waveforms of Figure 2 illustrate a comparison between the orginally existing digital data flow, the first row (a) showing binary frequency modulation with associated data flow, and the interference function, with modulation signal in row (b), which occurs on the occurence of the interference pattern illustrated in row (c) in accordance with (I) in Figure 1. It can be seen that the deviation peaks which occur in the frequency demodulator far exceed the voltage values of the so-called angular frequencies fo and fl. and thus exceed the maximum deviation consi dered tolerable. However, the illustration of the situation for case (I) in Fig. 1 shows that the character can still readily be read out, fundamentally because interrogation points in each case occur in the centre of a bit.

The conditions take a more serious turn if the middle frequency fm is altered, as shown for example by the value fm', which is the situation (II) shown in Fig. 1, which is synonymous with a low by-pass change ( < 2) in comparison to situation (I). Now the extinction, and the amplitude minimum occur at the angular frequencyf1,, and as it is in this state that the modulation function exhibits a reversal point (relatively low phase change speed), no decisive interference functions occur in this case. However, it is far more serious that as a result of the noticeable reduction in the reception voltage, the signal-to-noise ratio reduces, and in many cases even becomes negative, (i.e. undershots the minimum reception level).This results in an immediate loss of the possibility of reading out all the digital characters of the "one" state and a resultant intermediate fault rate at the output of the FM demodulator becomes very high.

However, in terms of amplitude the character can fundamentally be read out, because with all the "zero" state data characters (fo' in Fig. 1) the reception level is definitely higher than atfm'.

The situations shown in (I) and (II) in Fig.

1 represent two basic types of distortion, and in the following where these relate to situation (I) dealing with the extinction between the angular frequencies and consequently distortions only in the even of a character change this is referred to as "dynamic extinction". Where distortions relate to situation (II) in Fig. 1, dcaling with the extinction at the angular frequency and thus loss of the possibility of reading out one of the two digital states, which conditions lasts until the next character change, this is referred to as "static extinction".

By its naturc, static extinction can occur only when one of the two angular frequencies is located substantially at the extinction point. On the other hand, dynamic extinction occurs as soon as the point of extinction lies between the two angular frequenciesfo and f,. As a result, dynamic extinction and static extinction constantly merge with one another, as a result of the change in the posi tionofthe spectrum minimum.

A relatively non-problematic situation in multipath propagation occurs during operation at the middle frequencyfm directly at the addition point, e.g. in the situation (III) of Fig. l. Here neither noticeable amplitude distortions nor transit time distortions occur within the angular frequencies f"" and f,".

The frequency-modulated signal is practi cally undistorted in this case.

The conditions shown in Fig. 1 apply to fixed transmitter and receiver relationships, and represent frequency dependent amplitude and phase distributions. Generally speaking it can be said that the conditions remain constant at any one frequency for the duration of a conversation, provided that there are only fixed-location reflectors, and no mobile reflectors such as aircraft participating in the propagation, which must be expected in most cases. In the case of mobile operation, in addition to the frequency distribtion of the amplitude and phase characteristics, there is also a noticeable spatial distribution of these parameters with movement in space. The spatial distribution is directly related to the wavelength of the radio frequency.Therefore, in a critical situation the distance between the two minima corresponds to half the wavelength, (e.g. atf - 300 MHz = X/2 = 0.5m), the antenna of a vehicle moving at 36 km/hour = 10 m/sec, will accordingly pass through 20 minima per second. In order to obtain a plausible picture of the distortion consequencies, it is expedient when considering the relationships shown in Fig. 1 to replace the frequency axis by a time axis, and to displace the modulation bandy0 andf,, as illustrated in (I) towards the right at such a speed that the times required to pass through an amplitude and phase wave amount to 1/20 sec., and 20 such waves are passed through at a uniform speed per second.It follows from the above recognitions that in mobile operation, the cases of static and dynamic extinction represented by sita- tions (I) and (II) in Fig. t, and likewise situation (III), in which no FM-distortion occurs, all merge with one another in a rapid succession, corresponding to the passage of the spatial distribution, and repeat in a given speed-dependent period.

Now a process will be described by which it can be substantially ensured that digital characters can be recognised in each of the situations (I) to (III), assuming that a distortion correction process is to be provided which is not only economical but in particular is technically in a position to instantaneously recognise and therefore to compensate for the configurations of the propagation mechanisms automatically and exclusively at the location of the section, during the actual course of the normal information transmission. The advantage of such an arrangement is obvious, because as a result of the system control. it is no longer necessary to interrupt the data flow by any test transmission, since these are not necessary. This dispenses with the need for the provision of measures for test transmission at the transmitter, the instantaneous recognition of the effects of the propagation situation being carried out only at the receiver. Firstly, Fig. l, situation (I) will be considered in explanation of the base-band-frequency correction facilities relating to dynamic extinction. As defined, the extinction point then lies between the angular frequencies fo and fl. The effects of the extinction point can be seen from the waveform shown in Fig. 3a, which clearly indicates that in this case the interference function represents a frequency jump which occurs only in the event of a character change.This frequency jump occurs periodically, with Ol-sequences, and considerably impedes the analysis of individual bits, since it changes their energy content and thus produces a shift relative to longer zero state or one state sequences. This shift is independent of whether integrating or band-limiting means are employed for the further signal analysis and regeneration.

In order to avoid those undesired energy components in the demodulated signal which are produced by the phase jumps, it is possible to employ a blanking method, as illustrated in Fig. 4, which shows a limit value switch GS, which is actuated whenever a specific limit value, thus e.g. the normal deviation range value of fo or fl is overshot.

The normal, frequency-modulated signal is present at the input of the limit value switch GS, and a blanked signal is present at the output thereof. As a result of the blanking, a reduction to zero occurs where a large signal peak previously existed as can be seen by reference to Fig. 3b. In this way, although the peak is avoided, the energy component withdrawn from the individual bit is generally too great, and does not exclude signal analysis errors. A better possibility consists in a circuit of the type shown in Fig. 5, wherein, on the over-shooting of the above mentioned peak value, this value is stored in a sample and hold circuit SH, and is substituted for the duration of the overshooting of the limit value into the gap formed during the blank ing process.As the limit value switch exhibits a low response delay, the specimen of the reception signal which is to be stored is fed to the sample 1 and 2 hold circuit via a delay line At. During this time a change-over switch US is then switched to the sample 1 and 2 hold circuit, and consequently is no longer present at the direct signal input.

The result of this method of operation is illustrated in the waveform shown in Fig. 3c.

Thus dynamic extinction correction would initially appear to be a satisfactory solution.

However, this method fails in the case of static extinction (extinction of the angular frequency), since in this case no deviation range peak occurs.

Before the possibility of correcting the sta tic extinction is discussed, the characteristics of the substitution method outside the static and dynamic extinction range will be explained. The standard situation in this respect is the situation (III) shown in Fig. 1, whose demodulated FM-signal possesses no peaks which occur merely with dynamic extinction. Thus any limit value switch is not actuated, and in this case the unchanged, directly switched through input signal is present at the output of the corrector whilst using the substitution method.

This relatively simple arrangement itself facilitates automatic distortion correction, which adapts to the corresponding, instantaneous operating state in the range of the dynamic extinction, and outside the static extinction simultaneously and without time delay.

In order that dynamic extinction may be handled, it is necessary to observe the following conditions, namely that as soon as extinction occurs at the angular frequency during frequency modulation, and thus the minimum reception-level is undershot at this frequency, all distortion correcting processes based on FM-distortion correction fail. Previous conclusions regarding FM-distortion correction were based on the assumption that on account of the amplitude limitation prior to the frequency demodulation only the phase distortion is of interest.However if one considers the amplitude response prior to the limiter in the case of static extinction, such as occurs in situation (it), it can be seen that whenever the frequency fo' is reached, the intermediate frequency voltage reaches the maximum value, whereas in the case of extinction when the frequencyfl' is reached, it attains the minimum value. Consequently, an obviously analysable amplitude modulation which corresponds to the digital character sequence occurs in the intermediate signal prior to the limiter. In other words this means that whenever static extinction occurs in accordance with the frequency modulation, the amplitude modulation of the unlimited intermediate frequency signal is the most marked.

However, the occurrence of a correct amplitude modulation does not give any information regarding its anlaysis, as a serious difficulty is presented by the fact that the values of the intermediate frequency voltage can fluctuate by approximately 80 dB, i.e the analysable amplitude modulation is sufficiently high with a high IF-voltage, but is very low with a low IF-voltage. In fact, it is for low intermediate frequency voltages that analysis is most desirable. This disadvantage can be overcome by providing a negatively logarithmic amplifier having a high dynamic range in a parallel arm to the frequency demodulator, together with a seriesconnected amplitude limiter. The latter is followed by an AM-demodulator whose output emits a peak-to-peak voltage waveform which corresponds to the logarithm of the degree of modulation, which itself is inde pendent of the absolute reception level.

Another problem is that with static extinction (extinction at one of the angular frequencies), two different states inevitably exist, namely : (i) Extinction at the angular frequency fo, which corresponds to the digital "zero" state.

In this case the amplitude modulation is in phase with the digital character sequence as agreed, (ii) Extinction at the angular frequency f" which corresponds to the digital "one" state.

The amplitude modulation is then opposed in phase to the digital character sequency.

Thus, when necessary, a suitable criterion must be made available for the correct analysis of the amplitude modulation.

If the two described analyses of amplitude modulation and frequency modulation are carried out in a common related manner, a maximum degree of adaption speed, simplicity and economy are provided in the distortion correction of propagation disturbances.

Practical mcasurements have completely confirmed this recognition.

An exemplary embodiment, which is schematically illustrated by the block circuit diagram shown in Fig. 6 will now be described.

Fig. 6 shows a block schematic circuit diagram of the overall arrangement, consisting of an IF and demodulation section ZD, a dynamic distortion corrector DE, a static distortion correcter SE, and a data analysis unit DA.

The IF and demodulation section ZD forms part of a conventional receiver, and is shown only in schematic form. An IF input signal ZF is fed via an IF filter 1 to a limiter 2, and is demodulated in a FM-demodulator 3.

Between the filter I and the limiter 2, there is additionally connected a feed branch to a selective-IF-output connected to the static distortion corrector SE.

The output of the FM-demodulator 3 leads to the dynamic distortion corrector DE.

The dynamic distortion corrector DE receives the output signal from the FMdemodulator 3, which is fed to a switch 5, and in the evenh of disturbance-free FMreception the switch 5 is in its rest position so that the incoming signal is fed directly to a switch 13 to a data regenerator 15. As soon as any dynamic distortions occur with a minimum approximately in the middle of the deviation range, as shown for example by state (I) in Figure 1, the resultant peaks which occur cause a limit value switch 4 to be actuated, and switches the switch 5 into a second position. At the same time a switching command is fed to a sample and hold circuit 7, which receives a slightly delayed demodulator signal via a transit time element 6.At the instant of the response of the limit value switch 4, there is applied to the sample-and-hold circuit 7 a delayed signal whose instantaneous value corresponds in first approximation to that of the demodulated signal prior to any overshooting ofthe limit value. For the duration of any limit value overshooting, this instantaneous value is stored by the sample-hold circuit 7, and substituted in the data path via the switch 5.

This measure ensures that the energy content of the original bit is obtained, and that it can be read out in the regenerator 15.

The mode of operation of the static distortion correcter is as follows. In the event of static extinction, the IF signal which is fed out prior to the limiter 2 exhibits a bitsynchronous amplitude modulation. This AM signal is fed via logarithmic amplifier 8 to AM demodulator 9. The logarithmic amplifier 8 ensures that the amplitude of the data signal which appears at the output of the demodulator 9 is substantially independent of the reception field strength. This AM output signal passes from the output of the AM demodulator 9 to an AM limiter 10 and an inverter 11 to a switch 13, which initially remains in its basic position connecting the output of the switch 5.The polarity of the demodulated AM signal at the output of the demodulator 9 and the limiter 10 is either equal to or inverse to the demodulated FM signal at the output of the switch 5, in dependence upon whether one or the other of the two angular frequencies, as defined, have been extinguished. To provide the nonambiguous conditions necessary for this purpose, that component of the demodulated FM signal which can be read out is compared with the demodulated AM signal in a polarity integrator circuit, 12 and is inverted in the inverter 11, if necessary. The polarity integrator circuit 12 includes a coincidence circuit, in which, in dependence upon the relative equality or inversion of FM/AM, a correspondingly integrated decision value is emitted.

At this point the function of the inverter 11 and of the polarity integrator circuit 12 will be briefly described. As already mentioned, the phase of the AM function may be incorrect by 180 , regardless of whether the angular frequency is f,, corresponding to the "one" state, or the angular frequencyfo corresponding to the "zero" state at the extinction point.

The particular angular frequency occurring at the extinction point is unable to provide any sensible information in the FM demodulator. The other angular frequency, which does not occur at the extinction point, enables a completely clear statement, as whenever it occurs in the digital character sequence. there is a high level receiver input voltage and thus a high instantaneous value of the AM function. Figure 7 shows a coincidence circuit KS and polarity integrator IR, keyed on at all instants of high AM voltage so that a positive or negative integration voltage occurs, in dependence upon the polarity position of the AM signal relative to the FM signal. If the result is negative the inverter 11 is reversed, so that the AM function which is fed to the switch 13 obtains the correct polarity.

In an AM decision unit 14, which is accommodated in the data analysis unit DA, it is possible to automatically check whether a serviceable, bit-synchronous AM signal, and thus, with a certain degree of probability, no serviceable FM exists. If this is the case, which can be the case virtually only with static extinction, the AM decision unit 14 connects the switch 13 to the inverter 11, and the regenerator 15 is fed with the data obtained from the AM signal.

The above devices cooperate in the following manner.

The IF signal ZF, which has been distorted in accordance with the relevant propagation situation and may possess a level from -92 to -10 dBm, firstly passes through the IF filter 1 (B - 16 kHz) and then passes through a separating amplifier. With a level from in each case -82 toO dBm (1 mW), it simultaneously reaches the limiter 2 and the dynamics compressor 8, in order to be demodulated either in the FM demodulator 3 or the AM democulator 9. Now a signal which is proportional to the corresponding intended or actual disturbance deviation range is available at the output of the FM demodulator 3, and a signal which is proportional to the AM modulation degree available at the output of the AM demodulator 9.

In the case of a pure FM signal, no AM signal appears at the output of the AM demodulator 9, an AM decision unit 14 supplies a logic output signal "zero", and the FM-AM switch 13 remains in its rest position FM.

In this way the FM output signal which in this case is pure (free of disturbance function) of the FM demodulator 3 can pass directly via the substitution switch 5, which occupies the rest position, the FM-AM switch 13 and a baseband filter to the regenerator 15. This signal flow corresponds exactly to the conventional signal flow of an optimised FM-receiver.

However, pure FM occurs only rarely, namely when one single propagation path is provided. A comparable situation occurs, as already explained, in the case of multipath propagation, when the location of the radio frequency fm is at the maximum of the amplitude characteristic, as shown for Fig. 1, situation (III). Under this assumption, Fig. 8 illustrates in an upper line an oscillogram of the FM data flow and the related AMfunction at the output of the AM detector 9 in the lower line.

If the position of the spectrum is now changed, e.g. due to a change in the radio frequency, then as shown in Fig. 9, this results in a corresponding AM change. In this state, the AM signal in the lower line, although increased relative to Figure 8, it is not yet sufficient to actuate the AM decision unit, and neither would this be necessary, since the FM signal can still be satisfactorily read out.

If the spectrum is displaced further towards the zero point, as shown in Fig. 10 the angular frequency actually reaches the minimum, so that the FM signal can no longer be read out positively due to static extinction, whilst the AM signal is now completely formed. The AM decision unit 14 has already brought the FM-AM switch 13 into its AM position. The AM signal present at the output of the AM demodulator 9 is gated into the signal path via the AM limiter 10 and the inverter 11 due to the operation of the switch 13, with an amplitude corresponding to the FM signal.

If the spectrum is now further displaced, so that symmetry is achieved between the angular frequencies around the extinction point (Fig. 11) the AM output disappears, and the FM output exhibits an interference function since this is a dynamic extinction.

The AM decision unit 14 will now have reset the switch 13 to its starting position FM.

The interference function prevailing at the output of the FM demodulator 3 overshoots the limit value switch 4, which in the sample-and-hold circuit 7 leads the instantaneous value which is to be substituted from the delay line 6 to the substitution switch 5, which is simultaneously switched downwards by the limit value switch 4 for the duration of the overshooting, and thus substitutes the analogue value stored in the sample-andhold circuit.

In this way the regenerator 15 is offered an interference-function-freed signal at the output of the base-band limiting filter in respect of all the situations discussed.

The above described arrangement is able to automatically compenstate errors with a transit time displacement of At = 1/2 bits on the by-pass and with a maximum extinction depth of 22 dB.

If short noise or impulse disturbances occur additionally during the FM anaylsis, these may become manifest as short peaks in the modulation text. The dynamic distortion correcter automatically recognises and eliminates such peaks, and thus functions as an intereference blanking device.

Figure 12 shows details of an exemplary embodiment of the dynamic distortion corrector DE already described with reference to Figure 6, and Figure 13 is a set of waveform diagrams illustrating the flow of individual pulse processes.

The demodulated FM signal is applied to an input FM of the circuit, a typical pulse of which is shown in row a of Figure 13. In Figure 12, the circuit points at which the signals shown, in the respective rows of Figure 13 have been identified by the respective row letter enclosed in a circle. The signal FM is passed via a delay line 20 (line 6 in Figure 6) to a switch 25 in a limit value switch GS (stage 4 in Figure 6), which is controlled by a monostable state 23 in the sample-hold circuit SH. The FM-signal is passed to a unipolar limiter circuit 21 in the limit value switch GS. The limiter circuit is composed of a double voltage comparator whose positive threshold can be varied by means of an adjusting potentiometer 21a and the negative threshold of which can be varied by means of a potentiometer 21b.The response thresholds of the limiter 21 are set via the potentiometers 21a and 21b in such manner that the latter reacts to every positive and negative change (multi-path distortion, noise) which exceeds the useful level, as already described with reference to Figure 6.

For such time as this threshold is exceeded, the voltage comparator 21 emits an output signal which is connected via an OR gate 22 to the monostable stage 23 and is also fed to a logic AND element 23a. The drive signal, i.e.

the output signal of the OR gate 22, represented in row c of Figure 13, supplies a rectangular pulse for the length of time during which the positive or negative threshold is exceeded. The monostable stage 23 is set in such manner that from the rising flank of this signal it produces a narrow control pulse (rowd in Figure 13) for the switch 25 which is contained in the sample-hold circuit SH (block 7 in Figure 6). Thus the switch takes a sample from the output signal of the FMdemodulator, which signal is represented in row b of Figure 13 after it has been passed via the delay element 20, and a capacitor C is charged to the value of this sample via an amplifier 26.time 7 or 26.

The delay time T of the delay element 20 is short relative to the bit duration, but is contrived to be such that shortly before the threshold is excecded, from the signal b there is taken a sample which, at the maximum, corresponds to the amplitude value of the particular undistorted character information.

A monostable stage 24 triggers with the negative flank of the signal of the logic-AND element 23a and, an the OR gate 24a.

extends the substitution time by T. The output signal from the gate 24a controls a switch 32 which is closed during undistorted operation. In this case the FM-signal which can easily be read out passes via the delay element 20 and the switch 32 to an intermediate amplifier 33, and from there to the output E.

As shown in the block circuit diagram in Figure 6, the FM-AM change-over switch 13 therein is connected to this output. The output signal from gate 24a also passes via an inverter 31 to control a switch 30 which is open during undistorted operation. However, with any overshootings of a set level at the output of the OR gate 24a, a pulse occurs as indicated in row e of Figure 13, which closes the switch 30 and opens the switch 32.

The monostable stage 24 extends the substitution period, which has been shortened by the duration of the sample, by the delay time 7 of the delay line 20. The output end, comprising the switches, has been enclosed within a block S1, which essentially corresponds to the switch 5 of Figure 6.

In parallel to the capacitor C is arranged a switch 27 which is controlled by any pulse from an OR gate 29 in such manner that it opens during the time of the pulse and thus does not change the charge state of the capacitor. The OR gate 29 receives an opening pulse from the monostable stage 23 during the sample phase and receives an opening pulse from the OR gate 24a during the hold phase. That is to say that for the period of time following the pulse from rowg, the control signal of the switch 27, the charge stored in the capacitor C can pass via an amplifier 28 and the closed switch 30 to the FM-output.

During the remainder of the time the capacitor C is short-circuited, however, by the closed switch 27, i.e. is discharged. This ensures that no unchecked charges can pass from the capacitor C to the switch 30 and from there to the output. The switch 30 (control signals does not switch the substitution value to the output until the sample phase has been concluded.

Figure 14 shows details of the static distortion corrector for static operation which corresponds to the following parts of Figure 6, polarity integrator PI corresponds to block 12; a polarity inverter I corresponds to block 11; an AM limiter AMB corresponds to the AM-limiter 10; whilst elements 52 and 50 form the AM-decision device 14 for the data analysis unit DA forming the output of Figure 6.

In connection with that block circuit diagram it has been pointed out that the IF-signal firstly passes through a logarithmic amplifier 8 of known type and is then demodulated in a AM-demodulator 9. The output signal of this demodulator 9 is applied to an input AM of a clamping circuit 57. The latter serves to separate the mean d.c. voltage value which is determined by the field strength of the received signal. In the simplest circumstances, as indicated in the circuit, it consists of a series capacitor and a clamping diode in a shunt arm. Then the signal is fed to a lowpass filter 56, whose cut-off frequency is substantially equal to the maximum modulation frequency. Then the signal passes firstly to an AM-limiter 54, whose threshold can be set by means of a potentiometer 55.The potentiometer 55 is set in such manner that in pure FM-analysis states (Figure 1, situations I and II) the AM-ripple which occurs in multi-path reception as a result of the amplitude response is unable to activate the AManalysis. This limiter 54, which is designed as comparator, serves to limit the AM- signals and thus to convert them into an item of digital information. The signal then passes as AM-data flow into the block of the polarity integrator PI and into the AM-inverter I.

As already mentioned, in the case of an analysable AM-data flow, the IF-level is inevitably a certain amount highsr at one of the two angular frequencies than at the other. Accordingly the character polarity of the higher IF-level must also be able to be read out following the FM-demodulator.

Only this signal allows a reliable indication of the character polarity, i.e. a logic "zero" or a logic "one". In dependence upon the particular angular frequency at which the triggering occurs, the polarity assignment between the AM-signal and the FM- data flow can be identical or opposed. Thus the polarity integrator is supplied not only with the AM-signal from 54 but, for comparison, with the FMsignal, fed from input FM, via a low-pass filter 35. This FM-signal is obtained from the output E of the dynamic distortion corrector shown in Fig. 12. The low-pass filter 35 has a cut-off frequency which corresponds approximately to the maximum modulation frequency. Its output is followed by a limiter 36.

When a switch 37 is closed, this FM-signal then passes to an integrator element 38 formed by a series resistor R and a shunt capacitor C'. The switch 37 is closed only when the AM is so high that the threshold set by 55 is exceeded. In other words, during any undistorted angular frequency, the capacitor C' in the element 38 is charged, and retains this charge during any distorted angular frequency. A switch 39 is arranged in parallel to the capacitor C' of the element 38. This switch is controlled via a pulse shaper 40 and a monostable stage 41. From the rising flank of the AM data signal the monostable stage 41 produces a pulse which is extremely short in comparison with the bit duration. During this pulse the capacitor C' in the element 38 is fully discharged. The switch 39 is follows by a low-pass filter 42 and a pulse shaper 43.

The low-pass filter is to suppress the discharge switching peaks in order to ensure the necessary clearly defined polarity information at the output of the pulse shaper 43.

The processes are represented in Fig. 15, and will now be described in detail with reference to those waveforms.

Row a of Fig. 15 represents a distorted FM-signal which, during the period I which corresponds to one bit, contains a clearly defined statement, and during the period II can no longer be read out for one bit. During the period I the discriminator output voltage can be positive or negative in dependence upon the character polarity (logic "1" or logic "0"). This is indicated by a circle with a broken line for minus quantities. In the following period of time III (corresponding to a plurality of bits) the signal can be either positive or negative, but can be read out following the frequency demodulator. Row b represents the associated AM-data signal of the AM-demodulator, which appears as a digital output signal (row c) at the output of the comparator 54. This signal controls the monostable stage 41 and the switch 37.Row d illustrates the curve of the capacitor voltage ANC' for a case in which a positive polarity of the FM-signal existed. Row e illustrates the same curve for the case of a negative polarisation. On account of the initial condition (charge 0) for a short length of time equal to or less than one bit, in the event of a rise in the AM-signal via the monostable stage 41 the capacitor charge must be shortcircuited by the switch 39. This is effected by the signal shown in row f, Fig. 15. Thus in dependence upon like polarity or inverse polarity of AM-data relative to the FM-data, at the output of the pulse shaper 43 there appears a clearly defined item of digital adjusting information which, via the polarity inverter I, inserts the now analysable AMdata flow from comparator 54 into the FM-data flow in accordance with the direction of the character polarity.

In the simplest circumstances the polarity inverter I consists of an EXCLUSIVE-ORgate 51 as shown. This ensures that the input supplied by the comparator 54 always appears in accord with the FM-data at the output of 51. This signal is then fed via a switch 47 to the actual data output, i.e. to a data regenerator 48 (15 in Fig. 6). To ensure that the AM-analysis is carried out only with a sufficient signal/noise ratio and a reliable statement from the AM-limiter and polarity integrator, two retriggerable monostable stages 44 and 50 are provided. The monostable stage 44 is controlled from the output of the pulse shaper 43 and releases the AMswitch-over when a reliable statement has been supplied by the polarity integrator for a specific length of time. The monostable stage 50 is controlled via an AM-decision device AME.The latter consists of a comparator 52 having a threshold which can be adjusted via a potentiometer 53, and is driven from the AM-output of the low-pass filter 56. Here the same principles apply as for the potentiometer 55 in the comparator 54. When the AM level at the decision device 52 has exceeded the threshold for a specific length of time which is considerably longer than the bit duration, the monostable stage 50 releases the AM-analysis.

The actual change-over switch between AM and FM consists of switching paths 45 and 47, and of a polarity inverter 46, and is only switched over to AM-analysis when both the clearly defined AM-statements from the monostable stages 44 and 50 are available in a logic-AND gate 49. The time constant of the monostable stages 50 and 44 is fundamentally dependent upon the speed of change of the propagation medium and the associated automatic switch-over speed of the analysis states.

The description of Fig. 1 has been based on the assumption that transmitter and receiver are stationary, so that the energy distribution of the received signal level is fundamentally dependent upon the frequencies employed.

With predetermined radio frequencies, a movement of the minimum out of the frequency range or into the frequency range can be achieved by local modifications of the reflectors or fluctuations of the reflection and diffraction phenomena during the course of multi-path reception (ionospheric and tropospheric scatter reception). Generally speaking these modifications exhibit relatively low speeds.

If either or both the transmitter and receiver carry out movements during the operation. as is the case with mobile stations during travel, the received signal level obeys not only the frequency-wise energy distribution, but in addition the associated locationdependent energy distribution, of which the local spacing of the minima is directly proportional to the radio-frequency wavelength being used. In other words. during mobile operation, under the influence of long bypasses. and with stationary reflectors, the relevant degree of distortion changes in dependence upon location with the relative speed of transmitting and receiving vehicle, and in dependence upon the radio frequency wavelength being employed.For example when a radio frequency of 300 MHz is employed, with a half wavelength of 0.5 M, then at a speed of 10 m/s, (36 km/h) a mobile station travels through 20 minima per second. Fig. l shows the extent of the distortions if its frequency axis is replaced by a time axis and the modulation band shown in situation I is considered to be moveably displaced towards the right between the frequencies FO and Fl at a speed which is such that the times required to pass through an amplitude and phase wave amounts to 1/20 sec.. and 20 such waves can be passed through in each second at a uniform speed. The cases I, II and III. which have been represented in Fig. 1 thus merge into one another in a rapid sequence in accordance with the passing through of the spatial distribution, and are repeated with corresponding periods.

The substitution speed of the dynamic distortion corrector DE shown in Fig. 6 is dependent only upon the response-and transit time of the integrated modules employed therein. Thus the dynamic distortion correction is considerably more rapid than the maximum expected path change between transmitter and receiver.

Conditions are different in the case of the static distortion corrector SE. From the IFsignal which has been withdrawn prior to the limiter, and has been logarithmically evaluated and rectified, in the static distortion corrector SE, the bisynchronous a.c. voltage necessary for obtaining the AM-data is separated from the d.c. voltage corresponding to the mean field strength at the output of the amplitude demodulator 9 via a capacitor. If during mobile operation, the mean field strength changes periodically, the chargeand discharge time constant of the capacitor adulterates the value of the signal voltage occurring at the output of the amplitude demodulator if these time constants are no longer negligibly low in comparison to the speed of change of the mean field strength.

This adulteration of the a.c. voltage considerably impairs the analysis of the AM-data.

The a.c. voltage distortion which occurs with a conventional a.c. voltage separation with a capacitor coupling in the case of rapid changes of the mean field strength impairs not only the accurate operation of the AM-decision device 14 in Fig.6 and thus the correctly timed switch-over of the changeover switch 13, but also results in an asymmetrical pulse duty factor of the bit flow at the output of the AM-demodulator 9, as a result of which the integration result in the polarity integrator obtains a very large stray range. This can seriously restrict analysis of the data received via amplitude modulation.

Expediently the connection path from a second sampling circuit to a subtractor contains a low-pass filter in order to thus smooth the course of the change of the equal value proportional to the mean field strength, in a fashion which is more favourable for the function of the overall circuit.

The control signal for the pulse train supply to the second sampling circuit in dependence upon the change in the amplitude course of the output signal from a first sampling circuit is advantageously obtained in that the control input of the switch is connected to the output of the first sampling circuit via a differentiator, possibly in series with a pulse shaper stage.

The a.c. voltage separating circuit shown in Figure 16 replaces a capacitor coupling at the output end of the amplitude demodulator 9 of the static distortion corrector SE in Figure 6, or the units 56 and 57 of the circuit diagram shown in Figure 14. Two sampling circuits 116 and 117 replace the clamping circuit 57 and a low-pass filter 119 replaces the filter 56.

The sampling circuits 116 and 117 are supplied with the demodulated signal, and are both controlled by a pulse train T which is derived, at the input end, from the incoming signal, and which is applied to the sampling circuit 116 directly, and to the sampling circuit 117 indirectly via a switch 122. At the output end of the a.c. voltage separating circuit shown in Figure 16 consists of a subtractor 118 whose one input is supplied directly with the output signal from the sampling circuit 116, whereas the output signal from the sampling circuit 117 is fed to the other input of the subtractor 118 via the low-pass filter 119. The control input of the switch 122 is connected to the output of the sampling circuit 116 via a series arrangement of a differentiator 120 and a pulse shaper stage 121.

The voltage curves, plotted against time T, in the diagrams a to fin Fig. 17 represent the voltage curves at the correspondingly referenced points of the a.c. voltage separating circuit shown in Fig. 16. Diagram a shows the input end demodulated signal which represents a data flow fluctuating between the voltage values U1 and U2. In the centre of a bit, this input signal is in each case sampled pulse-wise by a pulse train having pulses of amplitude UT in a period of time which is short relative to the bit length. This initially applies only to the sampling circuit 116, which is directly controlled by the pulse train.

Accordingly the regenerated input signal appears with a symmetrical pulse duty factor at the output of the sampling circuit 116 in the form of a rectangular pulse train superimposed upon a d.c. voltage. This rectangular pulse train is differentiated in the differentiator 120, and having passed through the pulse shaper stage 121, is fed to the control input of the switch 122. The circuit which serves to derive the control signal for the switch 122 from the putput signal of the sampling circuit 116 is contrived to be such that only the rising flanks of the rectangular pulse train in diagram c reverse the switch from the open to the closed state. This results in the fact that the sampling circuit 117 only stores a value from the input-end, demodulated signal, which possesses a maximum value corresponding to the voltage value U1.As a result, at the output of the sampling circuit 117 there occurs a d.c. voltage which is represented in diagram e and has the value Ul. This d.c. voltage is proportional to the relevant mean value of the field strength of the received, original signal, and thus supplies a reference value for the amplitude modulation of the AM-modulated signal. The cut-off frequency of the low-pass filter 119 is contrived in accordance with the maximum radio frequency employed (location-dependent spacing of the attenuation maxima) and the maximum occurring relative speed between transmitting and receiving vehicle. This ensures that the maximum speed of change of the d.c. voltage at the output of the sampling circuit 117 is still transmitted in full via the low-pass filter, whereas more rapid changes caused by any disturbances are suppressed.At the output of the subtractor 118, the voltage curve represented in diagram f is obtained from the difference between the voltage values U1 and U2 in diagram a.

WHAT WE CLAIM IS : 1. A frequency modulated data communications receiver, for use in a system where digital signals are impressed onto a carrier for propagation, in which receiver means are provided to automatically prevent any information loss caused by phase or amplitude distortion in two mutually supplementary arrangements, one in the form of a dynamic distortion corrector comprising a frequency discriminator followed by means for the recognition of interference peaks together with a sample-and-hold circuit operable to lop any said interference peaks, and the other supplementary arrangement in the form of a static distortion corrector comprising an amplitude demodulator connected in parallel with said frequency discriminator arrangement, and the outputs of said demodulator and said discriminator being connected to respective inputs of a changeover switching circuit which is controlled by an amplitude modulation-analysis device, and which acts when the amplitude modulation level exceeds a given value to switch the amplitude demodulator to a common output which is otherwise fed by the frequency discriminator, the output of the AMdemodulator being followed by a polarisation inverter which operates under the control of a polarity integrator to reverse the AM-demodulation product, in dependence on the magnitude of the FM-demodulation product, in the sense to give polarity-correct AM-demodulation.

2. A receiver as claimed in Claim 1, in which a common IF input filter is provided, whose output is connected to the FM discriminator by a limiter, and to the AM demodulator by a dynamics compressor.

3. A receiver as claimed in Claim 2, in which said dynamics compressor is an amplifier having a negative logarithmic amplitude characteristic.

4. A receiver as claimed in any preceding Claim, in which said means for the recognition of interference peaks consists of a limit value switch which is operated in the event of any interference peak exceeding a given value to disconnect the signal path from the FM-demodulator to the output of said dynamic distortion correcter for the duration of the interference peak and connects said output to said sample-and-hold circuit.

**WARNING** end of DESC field may overlap start of CLMS **.

Claims (14)

**WARNING** start of CLMS field may overlap end of DESC **. the filter 56. The sampling circuits 116 and 117 are supplied with the demodulated signal, and are both controlled by a pulse train T which is derived, at the input end, from the incoming signal, and which is applied to the sampling circuit 116 directly, and to the sampling circuit 117 indirectly via a switch 122. At the output end of the a.c. voltage separating circuit shown in Figure 16 consists of a subtractor 118 whose one input is supplied directly with the output signal from the sampling circuit 116, whereas the output signal from the sampling circuit 117 is fed to the other input of the subtractor 118 via the low-pass filter 119. The control input of the switch 122 is connected to the output of the sampling circuit 116 via a series arrangement of a differentiator 120 and a pulse shaper stage 121. The voltage curves, plotted against time T, in the diagrams a to fin Fig. 17 represent the voltage curves at the correspondingly referenced points of the a.c. voltage separating circuit shown in Fig. 16. Diagram a shows the input end demodulated signal which represents a data flow fluctuating between the voltage values U1 and U2. In the centre of a bit, this input signal is in each case sampled pulse-wise by a pulse train having pulses of amplitude UT in a period of time which is short relative to the bit length. This initially applies only to the sampling circuit 116, which is directly controlled by the pulse train. Accordingly the regenerated input signal appears with a symmetrical pulse duty factor at the output of the sampling circuit 116 in the form of a rectangular pulse train superimposed upon a d.c. voltage. This rectangular pulse train is differentiated in the differentiator 120, and having passed through the pulse shaper stage 121, is fed to the control input of the switch 122. The circuit which serves to derive the control signal for the switch 122 from the putput signal of the sampling circuit 116 is contrived to be such that only the rising flanks of the rectangular pulse train in diagram c reverse the switch from the open to the closed state. This results in the fact that the sampling circuit 117 only stores a value from the input-end, demodulated signal, which possesses a maximum value corresponding to the voltage value U1.As a result, at the output of the sampling circuit 117 there occurs a d.c. voltage which is represented in diagram e and has the value Ul. This d.c. voltage is proportional to the relevant mean value of the field strength of the received, original signal, and thus supplies a reference value for the amplitude modulation of the AM-modulated signal. The cut-off frequency of the low-pass filter 119 is contrived in accordance with the maximum radio frequency employed (location-dependent spacing of the attenuation maxima) and the maximum occurring relative speed between transmitting and receiving vehicle. This ensures that the maximum speed of change of the d.c. voltage at the output of the sampling circuit 117 is still transmitted in full via the low-pass filter, whereas more rapid changes caused by any disturbances are suppressed.At the output of the subtractor 118, the voltage curve represented in diagram f is obtained from the difference between the voltage values U1 and U2 in diagram a. WHAT WE CLAIM IS :

1. A frequency modulated data communications receiver, for use in a system where digital signals are impressed onto a carrier for propagation, in which receiver means are provided to automatically prevent any information loss caused by phase or amplitude distortion in two mutually supplementary arrangements, one in the form of a dynamic distortion corrector comprising a frequency discriminator followed by means for the recognition of interference peaks together with a sample-and-hold circuit operable to lop any said interference peaks, and the other supplementary arrangement in the form of a static distortion corrector comprising an amplitude demodulator connected in parallel with said frequency discriminator arrangement, and the outputs of said demodulator and said discriminator being connected to respective inputs of a changeover switching circuit which is controlled by an amplitude modulation-analysis device, and which acts when the amplitude modulation level exceeds a given value to switch the amplitude demodulator to a common output which is otherwise fed by the frequency discriminator, the output of the AMdemodulator being followed by a polarisation inverter which operates under the control of a polarity integrator to reverse the AM-demodulation product, in dependence on the magnitude of the FM-demodulation product, in the sense to give polarity-correct AM-demodulation.

2. A receiver as claimed in Claim 1, in which a common IF input filter is provided, whose output is connected to the FM discriminator by a limiter, and to the AM demodulator by a dynamics compressor.

3. A receiver as claimed in Claim 2, in which said dynamics compressor is an amplifier having a negative logarithmic amplitude characteristic.

4. A receiver as claimed in any preceding Claim, in which said means for the recognition of interference peaks consists of a limit value switch which is operated in the event of any interference peak exceeding a given value to disconnect the signal path from the FM-demodulator to the output of said dynamic distortion correcter for the duration of the interference peak and connects said output to said sample-and-hold circuit.

5. A receiver as claimed in any preceding

Claim, in which said polarity integrator controls a coincidence circuit and the polarisation inverter to give the correct polarity of the AM-signal.

6. A receiver as claimed in any preceding Claim, in which an AM-limiter is connected between the output of said AM demodulator and said polarisation inverter.

7. A receiver as claimed in any preceding Claim, in which said means for interference peak recognition comprises a double voltage comparator provided with means for setting its positive and negative thresholds, said comparator being followed by an OR-gate whose output signal in the event of any overshooting of a threshold acts to temporarily control a switch which acts prior to the instant of overshooting acts to transmit the instantaneous value of a delayed input signal, whose delay time is short relative to the bit duration, to a capacitor whose charge is maintained until a delay period after any undershooting of the threshold, said capacitor being connected via a component of a changeover switch circuit to the output of the dynamic distortion corrector, limit value control means being provided which act on the occurrence of any switching pulse for a sample-and-hold switch and opens a short-circuiting switch for discharging the capacitor, and which closes a first pole of the changeover switch circuit, whereas it opens another part thereof which normally con nects the delayed signal to the said corrector output, and maintains this switching state until any overshooting of the threshold terminates.

8. A receiver as claimed in Claim 7, in which means are provided to feed the signal from the AM-demodulator to a clamping circuit which serves to separate the mean d.c.

voltage, and then via a low-pass filter to an AM-limitcr whose output is fed to one input of a polarity inverter that is in the form of an EXCLUSIVE OR-gate, and also to a switch which closes when any AM-signal of adequate magnitude appears, and also switches through the substitute FM-signal to a capacitor from where this signal is fed out via a low-pass filter and a pulse shaper. and also fed to the other input of said EXCLUSIVE OR-gate, the output signal from the AMlimiter being fed to a pulse shaper circuit which forms a pulse from each rising flank of this signal, which pulse is very short in comparison to the bit duration and acts to close a switch which discharges the capacitor, the output signal from the polarity inverter being passed via a switching path of the AM-FM changc-over switch which is closed in the case of legible AM signals being received.

9. A receiver as claimed in Claim 8. in which the output of the pulse shaper is connected with a retriggerable monostable stage, whose output leads to an input of a logic AND-gate, whilst the AM-signal output of the low-pass filter is fed via a threshold circuit and a retriggerable monostable stage to the other input of said gate, until a clear statement is obtained concerning the availability of readily analysable AM by said retriggerable monostable stage, when the AM-FM change-over switch is switched to the AM position, whereas in the normal situation, with reception of legible FM, the switching path of the AM-FM change-over switch is closed and the switching path is opened and the FM from the dynamic corrector output is connected to the input of the data regenerator.

10. A receiver as claimed in Claim 1, in which the output of the AM-demodulator contains an a.c. voltage separating circuit whose input possesses a first sampling circuit and second sampling circuit, which are mutually in parallel, the first being controlled directly and the second indirectly via a switch by pulses derived from the incoming signal, which separating circuit has at the output end, a subtractor whose two inputs are connected to the two outputs of the sampling circuits, and that the switch for the pulse train supply to the second sampling circuit is actuated in dependence upon the change in the amplitude course of the output signal from the first sampling circuit.

11. A receiver as claimed in Claim 10, in which a low-pass filter is connected between said second sampling circuit and said subtractor.

12. A receiver as claimed in Claim 10 or Claim 11, in which the control input is converted via a differentiator to the output of the first sample circuit.

13. A receiver as claimed in Claim 12, in which a pulse shaper is connected in series with said differentiator.

14. A frequency modulated data communications receiver substantially as described with reference to Figure 6, or as modified with reference to any one of Figures 12, 14 or 16.