FR2709554A1 - Method and device involving pulse Doppler radars with frequency sweep - Google Patents

Abstract

A Doppler radar with pulsed transmission and with a linearly modulated carrier frequency. On reception, switching (11, 12) is made to occur in synchronism with the transmitted pulses before making the frequency change (81) which compensates for the frequency sweep. This spreads out the spectrum of the stray signals related to the switching. With advantage, on transmission, the signals applied to the conversion stage (5), placed between the linearly variable frequency generator (4) and the modulator-transmitter (3), pass through a circuit breaker (21) in the generator (4) circuit and another (22) in the local oscillator (52) circuit, these circuit breakers being controlled at the repetition frequency as are the previous circuit breakers and the modulator-transmitter.

Description

Method and device for pulse Doppler radars with frequency scanning.

The invention relates to pulse emission and frequency scanning Doppler radars.

Frequency sweeping opposes the emission of fixed frequency pulses. The scanning consists in modulating the transmission carrier frequency, modulation which is of predetermined type. Very often, this is a linear modulation, that is to say that the transmission carrier frequency varies linearly as a function of time. between a maximum value and a minimum value, with a rapid return from the minimum value to the maximum value. The law of variation is therefore jagged.

The above is true in the short term, that is to say over a time interval at least equal to the time taken to completely filter the useful signals (final analysis filter ,.

Over a longer period, we can naturally proceed in "frequency agility", provided that the resulting frequency variations are slow compared to the time parameters which define the basic operation of the radar.

As regards more particularly pulse Doppler radars, the frequency of recurrence of the transmitted pulses must be fairly high, for example of the order of a few hundred kHz. These radars also have an important form factor, that is to say that the duration of the emission pulse is a significant fraction of the recurrence period, practically 25% of this for example. It is also necessary to avoid any passage of the transmission pulse in the reception channels. This supposes, in the latter, a switching, carried out synchronously of the transmission, to attenuate as much as possible the response of the reception channel during each transmission pulse. These switches are made using electronic switches. These produce each time the receiver is closed during the transmission time a spurious signal, which is repeated at the frequency of recurrence. The spectrum of these parasitic signals therefore comprises lines which are distributed as multiples of the frequency of recurrence.

A conventional method for eliminating these spurious signals consists in filtering all the corresponding lines, or at least those which appear in the useful reception band. It is clear that useful signals will be eliminated by the same filters. The resulting disadvantage is all the more serious the higher the frequency of recurrence and the form factor of the transmission pulses.

The present invention provides a more advantageous solution to the problem which consists in attenuating the spurious switching signals, by losing as little as possible of the useful signal.

To this end, the invention first of all proposes a Doppler radar method with frequency scanning transmission pulses, preferably with linear frequency modulation, in which, on reception, the synchronous switching of the pulses transmission is carried out before the frequency change which compensates for the frequency sweep, so as to spread the spectrum of the parasitic signals linked to said switching.

In practice, as will be seen below, the switching can be carried out at several levels simultaneously in the reception channels. The main part of the attenuation of the transmission pulses should then occur upstream of the above-mentioned compensating frequency change.

According to another aspect of the method of the invention, there is also provided, on transmission, in addition to the conventional switching exerted on the power amplifier, a switching acting on the upstream signals whose frequency is different from that of 'program.

The invention also relates to a pulsed and frequency sweeping Doppler radar, the general structure of which comprises an antenna with a duplexer, a modulator-transmitter coupled to the transmission side of the duplexer, a generator of scanning frequencies, conversion means for apply to the modulator-transmitter a frequency derived from that of the sweep generator, and a reception and frequency change chain, coupled to the reception side of the duplexer, this chain comprising a frequency change capable of compensating for the frequency sweep, thus than switching to the frequency of recurrence of the transmission pulses.

In the Doppler radar according to the invention, this latter switching is carried out before the frequency changes which compensate for the frequency sweeps, so as to spread the spectrum of the parasitic signals linked to said switching.

More particularly, the radar reception chain comprises a first frequency change stage, followed by the switching stage, then a second frequency change stage, the local signal of which is derived from the output of the frequency generator. sweep.

In the preferred version of the radar according to the invention, the medium frequency, that is to say the intermediate frequency available at the output of the first frequency change stage, is chosen to be relatively high, taking into account the frequency radar transmission. It is currently considered that this medium frequency must be at least of the order of one hundred MHz.

In a particular embodiment of the present invention, the output of the second frequency change stage is connected to a filter with a passband lower than the value of the frequency of recurrence of the transmission pulses. This filter selects the useful band for the amplification and processing circuits located downstream.

In a first particular embodiment, the switching stage comprises a broadband amplifier, a switching member downstream thereof, and preferably another switching member upstream of the broadband amplifier.

In another particular embodiment, the switching stage comprises a high-pass filter upstream of the broadband amplifier, and the other switching member is then placed at the level of the first frequency change stage, in the connection of this at its local signal source.

In the same way as for the method, the radar according to the invention may comprise, on transmission, conversion means consisting of a frequency change stage followed by bandpass filtering, with switching members. on the two inputs of the frequency change stage coming respectively from its local signal source and from the balayaqe frequency generator. These other switching operations make it possible, at times other than when the transmission pulse is present, to attenuate the signals upstream of the transmission channel whose frequency is different from the transmission carrier frequency.

Other characteristics and advantages of the invention will appear on examining the detailed description below, as well as the appended drawings, in which - FIG. 1 illustrates the electrical circuit diagram of a first embodiment of 'a radar according to the present invention - Figure 2 illustrates the electrical circuit diagram of a variant of the radar according to Figure 1 - Figure 3 illustrates a possible embodiment of the frequency generator 4 used in Figures 1 and 2 - Figures 4 and 5 illustrate two different detailed embodiments of the processing circuits 30 appearing in FIGS. 1 and 2 - FIG. 6 illustrates the general shape of a transmission carrier frequency with linear sawtooth modulation, as well as a corresponding received signal: and - Figures 7 to 9 illustrate the spectral characteristics of signals taken at different points of the radar according to the invention.

Figures 1 and 2 have common parts which will be described first.

An antenna 1 is coupled to a duplexer which is here a circulator 2. The transmission side of the circulator 2 is coupled to a modulator-transmitter 3, which plays the role of power amplifier for the transmission.

The transmission is carried out with frequency sweep, that is to say that the transmission carrier frequency is variable according to a known law as a function of time. The most common type of modulation is sawtooth linear frequency modulation, as illustrated by the SE curve in Figure 6. The frequency decreases, with a slope of constant variation as a function of time, between a maximum value
F and a minimum value F, t with a rapid return of max min
F to F The same figure 6 shows a signal received SR, which
min max. is shifted with respect to the transmitted signal SE by a time Tp equal to the time of return movement of the electromagnetic waves between the antenna and the target. The comparison, at a given instant, of the frequency transmitted and the frequency received provides an indication of the time Tp, and thereby the distance of the target from the radar. As they involve a comparison of frequencies, these radars are commonly called Doppler radars with frequency modulation ",
The modulator-transmitter 3 does not operate continuously, but in a controlled manner by a niche signal having a repetition rate or frequency of recurrence FR Conventionally, the signal FR is applied to the modulator-transmitter 3, to control its operation by pulses . In Doppler radars of the aforementioned type, the value of FR is quite high, for example of the order of 200 kHz. Per tailor, the form factor of the niche signal FR, as well as that of the transmission pulses, is quite large, that is to say that the duration of the transmission pulse is a significant fraction of the period of recurrence (equal for example to 25% of it)
We will now describe the means used on transmission to generate the carrier frequency.

The basic source is a linearly variable frequency generator 4. The frequency which it delivers is applied to a frequency conversion device 5, comprising a mixer stage 51, followed by a bandpass filter 53. The mixer 51 receives as heterodyne a frequency F1 supplied by a local oscillator 52. The frequency F1 is fixed (at least in the short term).

The bandpass filter 53 has a bandwidth at least equal to the band F min -F. Its bandwidth may
max be higher if the radar also operates in frequency agility.

The output frequency, swept available at the output of the generator 4, and the frequency F1 supplied by the local oscillator 52, are chosen such that their sum or their difference produces the signal SE of FIG. 6. In practice, the value of F1 is of the order of a hundred MHz at least.

Although the only command of the modulator-transmitter 3 by the repetition frequency FR is sufficient in certain applications, the invention also provides, as a matter of preference, that switches 21 and 22 are respectively placed on the two inputs of the frequency change stage 51. These switches are both controlled at the recurrence frequency, to be open in the absence of the transmission pulses.

A particular embodiment of the generator 4 is illustrated in FIG. 3. I1 comprises a voltage-controlled oscillator, or VCO 42, which receives the output of a voltage ramp generator 44 through a summing junction 43.

The output of oscillator 42 constitutes the output of qénerateur 4. It is also applied to the input of a delay line 45, the output of which goes to a first input of a mixer 47, while the second input of this directly receives the output of the oscillator 42. The output of the exchanger 47 is applied to a phase comparator 48, which moreover receives a stable reference frequency, and the output of the phase comparator 48 is applied to a loop filter 49, the output of which comes as the second input of junction 43. The reference frequency Fref is provided for example by a very stable quartz oscillator.

The loop filter 49 is preferably an integrating filter with phase advance.

Those skilled in the art will understand that the device illustrated in FIG. 3 makes it possible to obtain a linearly variable frequency very precisely, of the form illustrated in the graph attached to FIG. 3. If we note T the transit time in the delay line, at the desired slope for the frequency variation, we will take Fref =
The output of the linearly variable frequency generator 4 is also applied to a frequency transposition device 40, which can operate under the control of a transposed frequency control device 41, illustrated in dashed lines.

We will now focus on the reception part of the radar, which is not the same in Figures 1 and 2
In FIG. 1, the output side of the circiilator 2 is applied to a first mixer 61, which receives from a local oscillator 62 a fixed heterodyne frequency F2 (at least in the short term).

Preferably, the value of F2 is chosen such that the output of the mixer 61 remains situated in a fairly high frequency band, for example of the order of a hundred MHz.

This medium frequency is applied to a switching stage, which here comprises a broadband amplifier 65, followed by an orqane switch 11. and preferably preceded by another switching member 12. The switching members li and 12 are actuated at the frequency of recurrence FR
Those skilled in the art will understand that the switches involved in the present invention are of the solid state type, that is, they do not involve any mechanical movement. The practical implementation of the switches may differ according to the frequency ranges concerned, it being observed that the PIN diode switches are most of the time suitable.

By way of example, the attenuation achieved by the switch 11, when it is open, will be 50 dB. The attenuation of switch 12 will be of the order of 40 dB.

The output of this switching stage, that is to say here that of the switch 11, is applied to a second frequency change stage, which receives as local signal the output of the frequency transposition device 40. This transposition is established in such a way that the output signal from the second mixer 81 is of fairly low frequency, for example of the order of ten or so MHz.

The output of the second mixer 81 is applied to a bandpass filter 82, the bandwidth of which is normally less than the value of the recurrence frequency FR. The output of the filter 82 is applied to an amplifier 83, followed by an analog or digital processing device 30.

FIG. 4 illustrates an example of the structure of the device 30 in the case of analog processing. The output of the amplifier 83 is applied to a detector 31, followed by an extractor assembly consisting of a comparator 72 and a threshold generator device 33. The output of the extractor or comparator 32 is applied to a device analog processing 34. This will work on the arrival times of the echoes selected taking into account threshold 33.

FIG. 5 illustrates an exemplary embodiment of the device 30 in the case of digital processing. The output of the amplifier 83 is applied to a mixer 35 receiving as a local signal the output of an oscillator 36 with frequency F3, close to the central frequency of the filter 82. The output of the mixer 35 is applied to a blocker sampler 37 , followed by an analog-to-digital convertixsevr 38, then by a digital processing device 39, which can carry out processing of the same kind as in analog, but in a more sophisticated manner.

An explanation will now be given of the operation of the devices according to the present invention, using the spectral curves of FIGS. 7 to 9.

The curve of FIG. 7, denoted S1, illustrates the spectral shape of the signal emitted, taken for example at point A which is the emission input of the circulator 2. The spectral analysis is assumed to be made over a time greater than the frequency modulation period existing in the carrier. We then obtain a continuous spectrum between the frequency F min and the frequency F
max
FIG. 8 illustrates the spectral shape of different components of the signal present at point B, which is the signal input received from the second mixer 81. Here we find the spectral band S1 of the transmission signal. This spectral band is contained inside the bung bandwidth BP1 of the first frequency change stage 61. (It will be observed that FIG. 8 and FIG. 9 do not take into account the frequency changes made, to facilitate their comparison, between them and with figure 7).

The switches such as 11 and 12, actuated at the frequency of recurrence, produce parasitic lines R on the harmonics of these. We know that the amplitude of the lines tends to decrease as the order of the harmonics increases. as shown by the sloping curve CP in FIG. 8. On this one, the illustration of the partial lines R has been limited to those contained in the bandwidth BP1, and it also takes account of the attenuation due to the flanks of these, as shown by the examination of the first line R on the left.

It will be observed in passing that the bandwidth BP1 attached to the first mixer stage 61 (which includes a band filter not shown) is quite considerable. In this sense, one of the components of the present invention consists in choosing a first frequency change stage which provides a medium frequency of high value, with a large bandwidth, whereas the previous tendency was rather to make the compensation. frequency sweep from the first frequency change stage, in order to limit the bandwidth.

Reference will now be made to FIG. 9 which illustrates the alleu re of the signal at point C, at the output of the second mixer 81. This second mixer is the one which compensates for the frequency sweep on transmission.

Because, according to the present invention, no switching at the frequency F R is carried out downstream of the second mixer 81, no interference is created at this level; as for the parasites previously created, the lines R of FIG. 8, the compensation of the frequency sweep has the effect of transforming them into a quasi-continuous spectrum SR, corresponding practically to a white noise, which can be easily rendered lower level thermal noise, especially if the medium frequency is chosen high enough.

Conversely, the spectrum S1 has been transformed into useful signal pulses, which appear periodically on multiples of the frequency of recurrence FR The dashed line
BP2 represents the bandwidth of filter 82, which selects by filtering one of the signals SU, to use it for further processing.

Those skilled in the art know that an application of DopSiler pulse and frequency scanning radars consists in detecting objects, whether mobile or not, and in measuring their distance. This detection includes an acquisition phase followed by a tracking phase. The bandwidth BP2 can be made more or less wide, as well as moved, during the different phases of this detection. This can be obtained either by acting on the filter 82 itself, or, preferably, by modifying consequently the frequency transposition carried out by the circuit 80. This is the reason why the circuit 41 illustrated in dashed lines realizes a frequency control transposes. preferably, the circuit 40 is arranged from a phase loop placed around the oscillator 42 of FIG. 3. This phase loop is controlled by the circuit 41, depending on whether one is in the state of standby, awaiting the acquisition of a mobile object, or on the contrary in pursuit of this mobile object.

We will now return briefly to the structure of the switching stage incorporated in the reception channel in FIG. 1. Although a single switch 11 may be sufficient in certain cases, it is preferable to also provide a second switch 12 upstream of amplifier 65.

The reason is that the amplifier 65 is broadband, therefore vulnerable within certain limits to saturation.

Given the high level of the signal emitted, a first attenuator 12 is provided upstream of the amplifier 65, the attenuation of which is chosen to be just sufficient to avoid saturation of the amplifier 65. Downstream of this same amplifier, the switch It achieves the resre of the attenuation required to eliminate almost completely any influence of the transmission pulses <9 on the reception channels.

FIG. 2 illustrates in this respect a variant of the device of FIG. 1. There we find the switch 11 downstream of the broadband amplifier 65. On the other hand, upstream of the latter, there is only a high-pass filter 64 connected to the output of the mixer 61. Another switch 13 is then preferably installed between the local oscillator 62 and the mixer 61. The operation of this variant can be explained as follows: we will all remember First of all, that the medium frequency is of high value, the filter 64 can therefore cut all the frequencies below the useful band of the signal at medium frequency. This considerably relieves the bandwidth amplifier 65, and allows, without too many insertion losses linked to the various filterings, to be satisfied with the switch 11 downstream. On the other hand, it is of course necessary to interrupt the application of the frequency F2 to the mixer 61 by the switch 13, to allow the correct setting in frequency of the filter 64 ut of the amplifier 65 with respect to the frequency of program.

Of course, the present invention is not limited to the embodiments described. We can still provide ci'au very variants. One of the simplest consists in constructing from a single source the frequencies F1 and F2, said source possibly being a Gunn effect oscillator.

The method and the radar device according to the present invention can be applied in different fields, which use the frequency bands I, J, or even millimeters.

The functions exercised by the processing device placed at 30 in FIGS. 1 and 2 can include, for example, a standby filter bank, and a tracking monitor.

Claims (10)

Claims. 1. Doppler radar method with frequency scanning transmission pulses, characterized in that, on reception, the synchronous switching of the transmission pulses is carried out before the frequency change which compensates for the frequency scanning, so spreading the spectrum of the parasitic signals linked to said switching. 2. Method according to claim 1, characterized in that at transmission, in addition to the switching of the power amplifier, provision is made for switching acting on the upstream signals whose frequency is different from that of transmission. 3. Pulse and frequency sweep Doppler radar, of the type comprising - an antenna (1) with a duplexer (2) - a modulator-transmitter (3) coupled to the emission side of the duplexer - a frequency generator (4) scanning - conversion means (5) for applying to the transmitting modulator a frequency derived from that of this generator (4 '- a reception and frequency change chain, coupled to the reception side of the duplexer. this chain comprising a change frequency (81) capable of compensating for the frequency sweep and a switching (11) at the frequency of recurrence of the transmission pulses characterized in that this switching (11. 12; 11,13) is carried out before the frequency change which compensates for the frequency sweep, so as to spread the spectrum of the parasites linked to said switching. 4. Radar according to claim 3, characterized in that the reception chain comprises a first stage of frequency change (61), followed by the switching stage (11, 12), then of a second stage of change of frequency (81) whose local signal is derived (40) from the output of the sweep frequency generator (4). 5. Radar according to claim 4, characterized in that the output of the second stage (81) is connected to a filter (82), of passband lower than the value of the frequency of recurrence (FR) of the pulses program. 6. Radar according to one of claims 4 and 5, characterized in that the switching stage comprises a broadband amplifier (65) and a switching member downstream thereof (11). 7. Radar according to claim 6, characterized in that the switching stage comprises another switching member (12) upstream of the broadband amplifier (65). 8. Radar according to claim 6, characterized in that the switching stage comprises a high pass filter (64) upstream of the broadband amplifier (65), and in that the first frequency change stage comprises another switching member (13), on the side of its local signal source (62) 9. Radar according to one of claims 3 to 6, characterized in that the transmission conversion means comprise a frequency change stage (51) followed by bandpass filtering (53), with organs switching (21,22) on the two inputs of the frequency change stage respectively from its local signal source (52) and the sweep frequency generator (4). 10. Radar according to one of claims 6 to 9, characterized in that the switch member or members are of the PIN diode type.