FI110965B - Method for using Frequency Modulated Frequency Modulated Continuous Wave Radar (FMICW) signal in geophysical remote sensing - Google Patents

Description

110965

A method for using a frequency modulated interrupted continuous wave radar (FMICW) signal. sex in geophysical remote sensing. The invention relates to a method according to the preamble of claim 1 for the use of a frequency modulated intermittent continuous wave radar (FMICW) signal for geophysical remote sensing.

This author describes a modulation method for improving the usability of the return beam of a wind detector radar signal.

10

The invention relates to wind profiler radar, but the principle can be used in any remote sensing device that measures volume scattering or volume reflectance profiles.

The invention relates to ice systems that perform measurements of geophysical volume, such as atmospheric, sea, lake, soil, etc. The invention can be applied to weather radar, wind profiler radar or any other system that obtains independent samples of the medium at short time intervals relative to the fading time of the detected phenomenon of said medium. The invention is particularly applicable to, but not limited to, monostatic waste systems.

20

A typical fire profiler radar uses pulsed transmission. It sends a short pulse (τ in length) to the atmosphere and then listens to the return echo of this pulse. Responses from different distances are received at different times in the receiver. If the scattering density in the atmosphere were constant, the radar return wobble would show a dependence of 1 / r2 on the distance. In practice, the scattering density decreases slowly as a function of height in the free troposphere. This results in scattering dependence on the distance 1 / r ", where n> 2. The fact that the return echo varies - strongly as a function of distance - requires a large dynamic range at the receiver.

The average return power as a function of distance from the radar can be plotted as follows:

Pr = C 1 P, 1 η (Γ) / r 2, 30 110965 2 where C includes all constants including those concerning the antenna array, Pt is the average transmitted power, T | (r) is the average scattering density as a function of distance, and r is the distance the radar. In the pulsed form, Pt is constant. The formula applies to distance 5 that is less than the time it takes to travel back and forth to the pulse interval:

Imax = Co / 2 * Pulse »10 where Co is the speed of light and TpuisSi is the time interval for transmission of two consecutive pulses.

Typically, T | (r) is a function of unknown form. In the free troposphere up to heights approaching the troposphere, T 1 (r) is usually a function of distance.

The received power equation can be used from the first distance the radar is able to receive a return echo after the transmission is cut. Hereby we can say:

Pr = C * Pt * ri (r) / r2 where rcrmax (Equation 1) 20 When looking at Equation 1, it can be understood that wind profiler radars, but also tuned radars, often detect scatterers up to a certain distance. If the receiver power Pr falls below the detectable value at a distance ro, a large increase in the function Ti (r), i.e. greater than r.2, is required to make the signal detectable. There are times when this happens, especially in weather radar (if there is rain behind distance ro), but also in wind profilers when there is a layer of higher scattering density behind distance ro.

'·' The method of improving the detection limit ro has been to use encoded pulse transmission. The coding used is typically binary phase coding. When considering the same region resolution throughout this description, the coded pulse is composed of bauds equal to the original pulses in the above description. If there are n bauds per pulse and if the pulse repetition rate is kept constant (to keep the radial velocities at the same 110965 3

In the Nyquist region, the average transmit power Pt can be increased by n times. We can therefore use equation 1 to estimate the sensitivity of the wind profiler radar after having first updated the average power transmitted.

5 The problem of the strong distance dependence of the radar echo also persists in coded transmission; In addition, there is another problem: heights below co / 2 * n * T cannot be recovered from full decoding of the pulse. This is because the beginning of the pulse has already entered the receiving antenna at the time the receiver is turned on. This results in procedures known as partial pulse decoding, which adds to the hassle and uncertainty of the detection process.

The fact that partial pulse decoding is less secure than the use of simple pulses has led to the practice of using pulse profilers in two modes: down mode and top mode. The (simple-to-15) pulsed transmission is used in the sub-mode to acquire profiles up to the height ro, and the encoded pulse transmission is used to expand the profile past the height ro.

Another technique used in some atmospheric radar radars is the frequency mode-20 continuous wave method. In this method, the transmitter and receiver are continuously turned on, and there are different antennas for transmission and reception. In these systems, the average transmit power is equal to the peak transmit power, which gives the system greater sensitivity.

A weakness of the FMCW method is the need to isolate the transmitted waveform from the receiver. If the isolation is not sufficient, the effective transmission signal and the weak reflected signal cannot be distinguished due to the finite dynamic range of the receiver.

FMCW rigid systems have proven to be too complex in practice to be used in commercial weather radar and wind profilers.

110965 4

The object of the present invention is to overcome the drawbacks of the above-described technique and to provide an entirely new type of method for the use of frequency modulated intermittent continuous wave radar (FMICW) signal in geophysical remote sensing.

The invention includes a method for modulating a signal to be transmitted, a method for receiving a Siron / reflected signal from a geophysical medium to be measured, and a feedback loop used to derive from the received signal profile a new modulation parameter time that no replacement would be detected or replaced in a predetermined manner.

The modulation consists of a rigid system of frequency sweep and pulse. Frequency sweep frequency deviation determines the regional resolution of the geophysical profiling system. Continuous frequency sweep is divided into a repetitive chain of transmit and receive sub-blocks.

15

The receiving system receives the transmitted signals scattered / reflected by the geophysical medium to be measured. The signal from the antenna is only fed to the receiving system while the transmission is off. This allows only one antenna to be used, but still has a long payload compared to pass-through or encoded pulse modulation.

The object of the invention is achieved by changing the port frequency to maximize the average signal power at a desired distance from the antenna.

The method according to the invention is particularly characterized by what is stated in the characterizing part of claim 1.

The invention offers considerable advantages over the prior art.

110965 5

The invention provides an effective means of measuring geophysical backscatter profiles with a rigid system using the same antenna aperture for both transmission and reception. The present invention does not require a separate transmission aperture for transmission and reception.

However, the invention can be applied to two radiation apertures, i.e. multistatic systems, if other factors make such a method desirable.

The invention: 10 - substantially reduces the peak power required in a given continuous profiling area of a geophysical profiling system, - eliminates the deflection of two antennas in a high useful life profiling system, 15 - reduces the dynamic range of power detection required in a receiver of a geophysical profiling system.

By the method of the invention, it is possible to generate modulation that can be parameterized to obtain the desired illumination of the geophysical medium. The invention also provides a receiving system that changes the amount of scattering as a function of distance. The feedback loop of the invention sets the modulation parameters to obtain optimal (defined by the rigid system user) performance.

In the following, the invention will be described in detail with reference to exemplary embodiments and the accompanying figures, of which: Figure 1 shows a frequency-time-frequency curve of the principle of FMICW modulation, Figure 2 shows another frequency-time-curve of the principle of FMICW modulation. power-height curve comparison in the prior art and in the present invention, Fig. 4 shows a power-height curve comparison in the prior art and in this invention, Fig. 5 shows a power-height curve comparison in the prior art and in this invention, 7 is a multi-graph representation of the signals obtained by the invention,

Fig. 8 is another multi-graph representation of the signals obtained by the invention.

15

A solution to the known problems in the prior art is to use frequency modulated interrupt continuous wave transmission (FMICW) with a feedback control system controlling the modulation parameters.

An FMICW radar typically comprises the following steps. A signal is generated whose frequency changes with the set bandwidth. This signal is ported on and off at a predetermined frequency (fg) to form on-time and off-time slots. This gated signal is transmitted to the desired destination by means of a transmitter and an antenna. The reflected or backscattered signal is received during 25 µF time slots of the gated signal, and the desired properties of the received signal are detected and collected during the off time slots of the gated signal.

The FMICW transmission is described herein using the parameters of the boundary layer profiler. Let the system have a distance resolution of 150 meters and an interesting distance range from that 150 meter 30 upwards to preferably 3000 meters and above. A linear frequency sweep of 5 ms and deviation of 1 MHz is then formed. The frequency sweep is coupled to 110965 7 transmitters and the transmitter pulses modulates the signal by generating 20 | long pulses every 20 | increments. Figure 1 illustrates this.

Figure 1 shows the transmitted signal and return stacks at three heights: 1200, 3000 S and 4200 meters. The return echo of 1200 meters is delayed by 8 and the return echo of 3000 meters is delayed by 20. The return echo of 4200 meters is delayed by 28 | is.

In the FMICW ice system, the signal is only allowed to enter the receiver when no transmission is in progress. The baseband signal is generated by mixing the received signal with the non-transmitting sections of the frequency sweep. As a result, baseband return signals from the ice system are obtained at three heights: 1200 meters, 3000 meters and 4200 meters. The return signals from these heights are shown in Figure 2.

The frequency sweep shown in Figure 1 is continuous in phase throughout the 5 ms period. In the sections shown in Figure 2, the 1.6 kHz (1200 m), 4.0 kHz (3000 m) and 5.6 kHz (4200 m) signals also show a phase continuous throughout the frequency sweep, with a 5 ms scan much shorter as the decorrelation time of the volume scatterers to be examined. Typical decorrelation times for Bragg scatterers are in the range of 0.1 to 1.0 seconds for Kirk-20 in air.

When the return signals are phase continuous, each portion of each frequency of Figure 2 forms a sine wave which is interrupted in time. The intermittent 1.6 kHz signal has time to form 8 cycles of interrupted sine waves during a 5 ms scan, a 4 kHz signal corresponds to -25 corresponding to 20 cycles, and a 5.6 kHz signal generates 28 cycles. These sine waves are summed at the receiver, resulting in a voltage containing different intermittent frequency components associated with different heights.

When digital samples are taken while the receiver is on, a 125-point saq is generated (if sampled for each RX interval). The different frequencies at this time are separated by Fourier analysis. For example, the 8th, 20th, and 28th samples in this Fou 110965 8 rier transform correspond to heights of 1200, 3000, and 4200 meters. If the receiver is, as in the preferred case, capable of separating the so-called in-phase and transverse-phase components at baseband, the corresponding Fourier transform is complex and the frequencies obtained in the given example are negative.

5 The benefits of using FMICW modulation in wind profiling become apparent if we look at the average power received by the profiler. As shown in Figure 2, the portions of the 1.6, 4.0 and 5.6 kHz signals received during measurement are respectively 0.2, 0.5 and 0.3 of the full measurement period. At altitudes less than 3000 meters, the portion of the sine wave that is usable in the Fourier transform is directly proportional to the altitude. Thus, in FMICW modulation, equation 1 can be written as:

Pr = C * Pt * r / ropt * iKrj / r2, where r <ropt (Equation 2) 15 Note that in this case, Pt is 50% of the peak transmitter power, and the system transmission payload is significantly higher than in a pulsed or coded pulsed transmission system.

It is important to note that the power received in Equation 2 is proportional to l / r rather than 20 l / r. Thus, the power transmitted by the FMICW system is better utilized for detecting long distances than in a pulsed system. This increases the detection limit of the FMICW system.

Pr = C * Pt * (2-r / ropt) * T ^ rj / r2, where r0pt <r <2 * r0pt 25 The power received from this range is sharply reduced as a function of height.

It should be noted that the Fourier process, which detects signals at different heights, is a coherent process and thus corresponds to the coherent integration used in pass-through profilers.

110965 9

Let's now look at a wind profiler whose power amplifier is capable of producing 1000 watts of peak output power. consider three types of wind profilers: one that uses uncoded pulses (typical values: 1 | pulse with 40 ps pulse cycle), one that uses coded pulses (typical values: 6 baud pulses, 1 ps baud, and 40 ps S pulse) and FMICW- profilers who are supposed to track altitudes up to 3 kilometers. Let us now consider a parameter that describes the usable power that a wind profiler can transmit as a function of height. This parameter describes the system's ability to express atmospheric phenomenon as a function of distance. The actual atmospheric scattering density and antenna considerations are removed from the parameter. The power amplifier and modulation characteristics of the system 10 are maintained in the parameter. The parameter can be written to Pt / r2 for a transit system and to Pt / (r * ropt) for the FCIMW system. The correct value Pt, which includes the utility period, must be used for each system.

In Fig. 3, curve 1 corresponds to a pass-through wind profile, curve 2 corresponds to an encoded pulsed-15 wind profile, and curve 3 corresponds to an FMICW wind profile. Curve 2 starts at 900 meters, indicating that transmitter power can only be fully utilized at this height due to pulse decoding.

3, the value printed as a function of height is the average power in the resolution cell 20 divided by the square of the distance. The value is normalized so that it matches the average power at one kilometer altitude.

Let us now look at an illustrative example. Suppose the scattering density η (τ) is a standard value at atmospheric lower kilometers and a standard value such that 100 W average power is required to detect the signal at a height of 1 km. It can be seen from Figure 3 that under these conditions a pulsed wind profiler, coded pulsed wind profiler and. ** The FMICW wind profiler detects the atmospheric signal in heights of 500, 1200 and 1650 meters, respectively. Because the user was interested in the wind profile above 150, and preferably up to 3000 meters, he sets the system to perform detection at 30 maximum average power at 3 km. In this case, the limited scattering density limits the expression to a maximum of the above-mentioned heights.

110965 10 The user now detects atmospheric return as a function of altitude. Since the atmospheric signal is absent at an altitude of more than 1650 meters, it can be concluded that the average power was not sufficient to produce a signal at the prevailing r | (r). The system of the present invention 5 sets new pulse parameters based on the detected profile. The system now makes settings to send 11 ps pulses at 11 ps intervals to maximize radar illumination at the height at which the signal is lost. As a result, the average illumination power changes to correspond to the situation in Fig. 4.

10

As can be seen in Figure 4, the FMICW system can now detect an atmospheric noise signal up to 2000 m. The usability of the wind profiler signal is thus improved. The improvement is effected at the expense of the altitude detection power of 2.1 to 4.2 kilometers, but since the signal could not be detected at these altitudes at the first 15 settings, there is no loss of signal usability compared to the first setting.

In the third stage, the system detects a signal up to 2000 m. This corresponds to a delay of 13.3. The system can now set the timing parameters to 14 ps in order to produce an illumination profile that is better suited to the height range from which the signal was detectable. Fig. 5 shows a new modulation setting by curve 5.

The FMICW profiler can now detect an atmospheric signal up to 2200 meters, which is far more than the original 1650 meters performance.

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The above example gives a rough example of how the FMICW parameter feedback can be implemented. Other methods, such as calculating the maximum achievable height from the first profile based on the T 1 (r) model, are part of the invention. The pulse parameters may not be set steplessly, but there may be a number of modes from which the system can select under the prevailing conditions. This is also part of this invention.

π 110965

One important embodiment of the invention is that the feedback is continuously switched on. In such a case, the rigid system updates the FMICW modulation parameters continuously to provide maximum information about the current geophysical situation.

The invention may be used in a form where the transmission period of the transmission is less than 50%. In this case, the illumination curve consists of three parts, one with a 1 / r dependence, another with a 1 / r dependence, and a third with a sharp dependence (approximately 10 l / r 3).

A preferred embodiment of the invention will now be described with reference to Figure 6.

Kim has now sampled one pulse, after integration over a reception period, each frequency component in the post-integration signal contains a signal portion and a noise portion. The signal portion consists of the integrated voltage over the duration of each curve shown in Fig. 6. However, the noise portion of this frequency compartment corresponds to integration over a full 20 µc period. Although the signal portion remains coherent over multiple pulses, the noise portion does not, and thus the noise has a detrimental effect on the signal in such a way that it interferes with the phase and voltage of the actual desired signal.

Let us now consider one continuous 5 ms sweep and the corresponding output from the FMICW receiver. Let there now be an object, either discrete or volume scatterer, at a distance of 1200 m, which corresponds to a 1.6 kHz signal in Figures 2 and 3. If we have a receiver 25 with a bandwidth of 1 Mhz, a matched bandwidth value for the transmitted waveform, signal at 1 MHz quadrature sampling at baseband. We do not consider the effect of the finite steepness of the transition bands of the receiver's concatenated filters or the folding effect there. Practical. in the device, taking into account these phenomena, a sampling bandwidth greater than 1 MHz could be selected. However, this does not change the situation substantially - in this case the same part of the time would be digitally integrated as in these examples. At that time, however, there would be more samples to be integrated per case.

12 110965

In Figure 7, the top result line shows the time-domain signal (simulated) of the object at 5,200 meters, plus noise. The right-hand output shows the result of using a 5000-point high-speed Fourier transform to bring out different frequency components - and simultaneously distances - from the signal. By this method, the signal-to-noise ratio of the target is 15.9 dB when riveted to a bandwidth greater than 200 Hz. It occurs at 1.6 KHz carrier band at the 8th frequency compartment, which corresponds to the 8th height compartment at 1200 meters. Signal 10 consists of 20 zeros, which corresponds to the transmission time during which the receiver is muted, followed by 20 samples of signal and noise, followed by 20 zeros, etc. (see Figure 8). The time unit in the left column of Fig. 7 is 1 ps, and the Frequency unit in the right column is 1 MHz. In this specification, it is assumed that the mute is achieved in the digital parts of the equipment, which allows for a complete null-15 of the transmission time signal. It should be noted that the 1 MHz bandwidth is greater than the 25 kHz pulse repetition rate, which causes copies of the signal to be + 25 kHz off center frequencies.

In Fig. 7, the middle output row shows the effects of using over 20 ps cycle, full reception, integration, and 125-point fast Fou-20 rier conversion to reveal components of different signal areas. The length of the Fourier transform was reduced without degradation in the signal-to-noise ratio, which is now 15.7 dB. The pulse effect is no longer visible due to the reduced Nyquist frequency at the base frequency. The integration is accomplished by summing the values from each receiving period, resulting in higher voltage amplitudes. It is noteworthy that here, the 25 signals to be examined, at 1200 meters, correspond to a rotating pointer which rotates less than 5 degrees during one reception period. Thus, it is possible to sum up all samples from the receiving period without significantly reducing the signal voltage due to the addition of rotating indicators.

The top two rows of Figures 7 and 8 can be considered as being included in the prior art. The last line of Figure 7 illustrates the effects of the integration 110965 of the present invention. In this case, a discrete Fourier transform that can also be implemented using only one of the fast Fourier transform output components. is used to display a frequency component of 1200 meters. The difference is that instead of using a full receive time integrated signal, an integrated signal is now used to match the integration time with the bidirectional delay between transmission and echo reception of said destination. In this case, only the first 8 samples are integrated during each reception period to produce the left-hand time series of the last row. The integration of the eight samples was chosen for this discrete Fourier transform (DFT) because it aligns the start of the reception-10 period with the signal scattered from the subject. This can be seen in Figure 6: only the first 8 microseconds of the reception period contain a scattered signal at a distance of 1200 meters. The amount of noise in this time slot is less than the noise in previous time slots because the last 12 samples have been omitted from integration, and the noise in these samples does not contribute to the output of the 8-point integrator.

The signal itself remains unchanged since all information from this area is used as shown in Figure 6. In this example, the signal-to-noise ratio due to the integration of the partial reception time is 19.5 dB, 3.8 dB higher than the signal-to-noise ratio. reception period integration.

The essence of the invention described herein is that a plurality of digital integrators are constructed, preferably by continuously summing and using different integration times for each region. For example, an area of 150 meters could thus be detected using single point integration, i.e., a first sample of each receiving period in conjunction with a DFT that detects a first non-zero frequency from a 125-point time interval comprising said first samples. The 300-meter range would be detected using 2-point integration, i.e., the first two samples summed, from each receiving period, and exporting the resulting 125-point time to DFT, which extracts the second non-zero frequency component from the 125-point time series. This procedure is continued up to pulse-matched altitude, in this case 3000 meters, for which a full 20 30-point integration result is used and for which the DFT determines the 20 non-zero frequency component.

14 110965

For altitudes above 3000 meters, only the remainder of the reception period is used for integration to obtain all signal information and to minimize the amount of noise corresponding to the frequency component. For example, for the 4200-meter signal shown in Figure 2-3, 5 of the last 12 samples of each 20 fis-long reception period sampled at 1 MHz are used, with the DFT displaying a 28th frequency component.

One preferred embodiment of the present invention would be to use a DSP or FPGA system to compute area-specific signals using a cumulative sum-10 register to determine the waveform transmitted at a sufficiently high frequency and multiplying the contents of said register by complex weights corresponding to the DFT coefficient for that pulse. The multiplication would result in an input to another complex summary register for each region, and the contents of those registers would be read at the end of each sweep. These values are complex voltages corresponding to each height for that sweep. The DFT registers are read and reset at the end of each sweep, and the integrator registers are reset at the beginning of each receive cycle.

It is within the scope of the invention that DFT is used in place of a digital summation of all samples during a single pulse reception period on a region-matched subset of the original 1 MHz samples. This is useful when objects are detected in areas that are within the sweep repetition period. In the exemplary case presented in this document, for example, objects hundreds of kilometers away would be suppressed if a sum integrator was used. This is because the complex voltages of the rotating pointer are summed over the time at which the pointer rotates substantially. So, for objects that are very far from the radar, * - the preferred method is to use suitable subsets of the original samples as input to the long DFT. In these cases, the summing of the digital samples would result in some loss of signal amplitude due essentially to the summing of the rotating pointer 30.

15 110965

In the example corresponding to Fig. 7, the amount of detection enhancement, 3.8 dB, is due to the fact that the target being detected remains somewhat coherent for a 5 ms scan period, while the noise is completely incoherent, i.e. has no phase correlation between samples. The noise power is thus additive during the reception period, and the preferred 5 integration methods receive 8/20 of the noise power instead of 20/20 of full integration. The factor 8/20 corresponds to 4.0 decibels, which is somewhat consistent with the simulated result.

To better illustrate the partial integration during pulse reception, FIG. 8 shows blurry prints of FIG. 7.

In Fig. 8, the data in the second row is obtained by summing 20 points of the top output signal per reception period. The data in the last line is obtained by summing 8 points per reception period, which results in less noise. A similar πιει 5 procedure is used for all heights separately.

Although the typical utility period for porting is 50%, according to the invention, the utility period can range from 10% to 90%.

20 • c • ««

Claims (10)

A method for using a frequency modulated interrupted continuous path (FMICW) radar signal, which method comprises the following steps: - a signal whose frequency is changed at a specified bandwidth (B) is generated; (fg), forming on-time intervals and off-time intervals, - the gate controlled signal is sent to a desired message by means of a transmitter and an antenna, - a reflected or scattered signal is received during the off-time interval of the gate controlled signal. , and - the desired characteristics are detected and collected from the received signal, characterized in that - the gate frequency (fg) is changed to maximize the average signal power at a desired distance from the antenna. 15 A method according to claim 1, characterized in that the useful period, which is a result of the gate operation, amounts to about 50%. Method according to Claim 1 or 2, characterized in that the gate frequency is up to about 5% of the bandwidth (B). Method according to claim 1 or 2 or 3, characterized in that the gate frequency 20 is varied within a range between 1 and 50% of the bandwidth (B). i9 110965 Method according to any of the preceding claims, characterized in that the change in gate frequency (fg) is carried out in stages. Method according to any of the preceding claims, characterized in that the change of the gate frequency (fg) is carried out in a continuous manner. 5 Method according to any of the preceding claims, characterized in that a desired portion of the off-time interval is detected and / or processed to obtain information from a desired distance and to minimize the noise from the signal related to the desired distance. Method according to claim 1, characterized in that only the beginning of the off-time interval is processed. Method according to claim 1, characterized in that only the end of the off-time interval is processed. Method according to any of the preceding claims, characterized in that the method is used for geophysical remote analysis. 15 ·