EP4128511A1

EP4128511A1 - Circuit for limiting the inrush current

Info

Publication number: EP4128511A1
Application number: EP21716972.1A
Authority: EP
Inventors: Michael Daus
Original assignee: Freifrau Von Fuerstenberg Stephanie; Freiherr Von Fuerstenberg Maximilian
Current assignee: Freifrau Von Fuerstenberg Stephanie; Freiherr Von Fuerstenberg Maximilian
Priority date: 2020-03-30
Filing date: 2021-03-25
Publication date: 2023-02-08
Also published as: WO2021197997A1; DE102020108780A1

Abstract

The invention relates to a circuit for limiting the inrush current, said circuit comprising a voltage input (1), a voltage output (2) for a load, and a rectifier (3). An input side of the rectifier (3) is connected to the voltage input (1) and an output side of the rectifier (3) is connected to the voltage output (2). An output capacitor (C9) is located between a first pole (2a) and a second pole (2b) of the voltage output (2) or the output side of the rectifier (3). The circuit comprises a control group (4), wherein the control group (4) has a control element (T3) for limiting the charging current of the output capacitor (C9). The control element (T3) has at least one control input (14) and a two-pole power path. The control group (4) comprises an actuator (5) for influencing the control element (T3), wherein the actuator (5) is connected to the control input (14). The control group (4) has a sensor element (6), wherein an output of the sensor element (6) is connected to an input of the actuator (5). An input of the sensor element (6) is associated with the output side of the rectifier (3).

Description

Circuit for inrush current limitation Description:

The invention relates to a circuit for limiting inrush current. Such circuits are already known to the person skilled in the art from practice and are often housed in power supplies. These circuits have a bridge rectifier which rectifies the charging current from an AC voltage input to a DC voltage output with an output capacitor. If the uncharged output capacitor were directly connected to the rectifier, an undesirably high current would flow to the output capacitor right at the beginning of the application of the alternating voltage.

To limit inrush currents, it is known from practice to connect a thermistor between the rectifier and the output capacitor. When the AC voltage source is applied, the thermistor is initially still cold and therefore has a high ohmic resistance. Due to the current flow through the thermistor, the thermistor heats up increasingly so that the ohmic resistance of the thermistor becomes increasingly lower. At the same time, the output capacitor draws less and less current because it carries more and more charge. As a result, the effects of the increasingly warmer thermistor and the increasingly more strongly charged output capacitor compensate each other with regard to the flow of current, so that the flow of current is evened out. This protects both the voltage network and, above all, the circuits involved and thus usually at least the power supply unit itself and the loads connected to it. The consumer is naturally particularly affected when it is a predominantly capacitive load. A disadvantage of such a circuit is that a not insignificant amount of heat is generated on the NTC thermistor and power loss is accepted. In addition, current still flows through the thermistor when it is no longer needed to limit the current, because the charging process of the output capacitor has already progressed accordingly. After all, high inrush currents still occur when the output capacitor is discharged and the thermistor is still hot. A power supply unit that is connected to a load (e.g. a mobile phone) and an alternating voltage source may serve as an example. If the power supply unit is disconnected from the voltage source after a while and then immediately reconnected to it, the output capacitor can discharge and the thermistor can be hot, so that high currents occur.

To counteract this problem, a control element is arranged parallel to the thermistor, which in this case is a voltage-controlled switch in the form of a thyristor. If the voltage at the output capacitor is sufficiently high, the thyristor short-circuits the thermistor, as proposed in DE 69623394 T2. The current then flows through the thyristor with practically no loss, while at the same time the thermistor can cool down at an early stage. This reduces both the heat losses and the probability of high inrush currents - especially in the case of connections to the voltage source in quick succession.

Nevertheless, unnecessarily high heat losses still occur due to the required heating of the thermistor. It can also easily happen, for example, that a power supply unit is left in the sun or in some other hot environment (in the vehicle, on a hot window sill, etc.). Then when the power pack is connected to the voltage source, a high switch-on ström, which under certain circumstances can lead to the immediate destruction of the circuits involved, but can in any case impair their service life.

For this reason, DE 102006014297 A1 suggests dispensing with the NTC thermistor. For this purpose, a control group is used, which is referred to as the “control unit ST” and which enables phase cutting control. The control group or control unit controls a control input of a control element in the form of a MOSFET switch S1 (= V1), so that it opens and closes at certain times according to the phase cut control. As evidenced by FIG. 2, the phase cut control permits short-term, needle-shaped current peaks which, however, do not exceed a defined maximum value. In particular the elements D1 and T1 from FIG. 4.

If the voltage drop across D1 from FIG. 4 becomes too high and exceeds the breakdown voltage, D1 becomes conductive, the voltage at the base of T1 rises, T1 then switches through and S1 or V1 blocks due to the voltage drop at the gate of V1. The switch T1 is thus an actuating element of the control group for the control element S1 or V1, whereas D1 forms a sensor element of the control group. While the output of the sensor element D1 is connected to the input of the adjusting element T1, the input of the sensor element is connected to a sensor node, which in turn is arranged between the voltage input and the control element S1.

According to FIG. 5, an evaluation unit 130 closes a switch S2 when the output capacitors C1 and C2 are fully charged in order to bypass the control group ST and the control element (switch S1). As a result, the losses of the avoided thermistor are completely eliminated. Also not applicable the risk of accidentally high inrush currents at high ambient temperatures.

In the last two decades, the legal regulations regarding the efficiency of power supply units, including with regard to climate change, have been tightened. A relevant series of these regulations is the “Energy Conservation Standards” series of regulations issued by the US Department of Energy, the most recent of which is the so-called Level VI specification. The Level VI specification came into effect in early 2016 and is one of the strictest regulations in the world. Due to the trend towards harmonization, further state and supranational regulations will follow the example of the USA. For example, the Level VI specification has already had an impact on the currently ongoing EU legislative process. In addition, manufacturers of power supply units will always be guided by the strictest regulations in order to open up the largest possible sales markets.

Depending on the nominal power and the nominal voltage of the power supply unit, the Level VI specification requires that a minimum, average degree of efficiency is exceeded when the load is connected and that the maximum no-load power is undercut. "Idle power" means that the power supply unit is connected to the voltage network, but not to a load - for example to an electronic device. According to the Level VI specification, commercially available power supplies for, for example, smartphones with a nominal power of 22.5W and a nominal voltage of 5V must have an average efficiency of> 83%. The maximum idle power ("stand-by mode") for this device class is 0.1W. In particular, the maximum idle power is already a challenge for the manufacturers of power supplies. It is also foreseeable that state and supranational Due to climate change, borrowed authorities will strive to tighten the regulations beyond the Level VI specification.

The invention is therefore based on the object of limiting inrush currents of electronic or electrical devices - in particular of power supplies - and reducing associated losses. The aim is preferably to increase the average efficiency of power supply units when the load is connected and / or to reduce the no-load power of power supply units.

This object is or these objects are achieved by a circuit for inrush current limitation, comprising a voltage input, a voltage output for a load and a rectifier, an input side of the rectifier is connected to the voltage input and an output side of the rectifier is connected to the voltage output, wherein an output capacitor is arranged between a first pole and a second pole of the voltage output or the output side of the rectifier, the circuit comprising a control group, the control group having a control element for limiting the charging current of the output capacitor, the control element having at least one control has input and a two-pole power path, the control group having an actuating element for influencing the control element, the actuating element being connected to the control input, the control group having a sensor element, an output of the sensor element is connected to an input of the control element, characterized in that an input of the sensor element is assigned to the output side of the rectifier.

The invention is initially based on the knowledge that the constant switching of the circuit breaker of the known phase cut control causes greater energy losses under full current load. The invention is also based on the knowledge that the sensor element should be arranged as close as possible to the output capacitor so that the control of the inrush currents can be designed in a particularly targeted manner. It has been found that both the measurement of the controlled variable and the setting of the manipulated variable can then take place in the vicinity and preferably in the immediate vicinity of the output capacitor and the regulation can thus be designed particularly effectively.

The invention is also based on the knowledge that almost linear courses of the charging of the output capacitor can be achieved with such circuits, so that the inrush currents are evened out in a manner that was previously unattainable. As has been found, the strong equalization is at the same time an essential basis for using a minimum of energy. With appropriate dimensioning of the components of the control group for, for example, a power supply unit of a smartphone, the power consumption of these components during the inrush current limitation is in the nanowatt range. After charging, power consumption or energy consumption can practically no longer be measured. These savings are so great that, for example, the efficiency of the corresponding network components is increased by several percentage points. This means that this power supply unit can be made compliant with the regulations simply by integrating the control group into a corresponding power supply unit. The no-load power is positively influenced to an even greater extent by the control group according to the invention, so that in particular the critical hurdle of the required, lower no-load power can be overcome more easily.

Furthermore, the equalization of the inrush currents is already advantageous because this avoids the needle-shaped currents from DE 102006014297 A1. The needle-shaped current peaks initially generate harmonic waves, which have a negative effect on some downstream elements. In particular, transformers are mentioned as examples (which are also used in switched-mode power supplies), which are more thermally stressed due to the harmonics and thus cause losses. In some cases, the needle-shaped currents can also damage or even destroy components. In addition, by avoiding needle-shaped currents, undesirable distortion reactive powers are reduced, which would burden the low-voltage networks to a significant extent due to the large number of power supply units. After all, the above-mentioned objects are achieved by the circuit according to the invention and, moreover, the disadvantages caused by needle-shaped currents are also avoided.

The term “voltage input” preferably means two individual electrical connections, each of the two connections being assigned to a pole. The voltage input can, for example, be a plug with two pins, which are designed to be connected to sockets of the low-voltage network. The voltage input or the circuit can be part of a Device. The device can in particular be a power supply unit, preferably a switched-mode power supply unit, for an electronic terminal. The electronic terminal may be a smartphone or a tablet, for example. The device can, however, also be a DC voltage converter or a frequency converter. The frequency converter can for example be designed for the operation of electric motors, so that the circuit according to the invention in the device, for example in the switched-mode power supply or the DC voltage converter or the frequency converter, can be dimensioned very differently. This means that the circuit functions in a wide variety of power ranges and the components can be scaled almost as desired without departing from the scope of the invention.

The word "voltage output" preferably means a two-pole connection within the device. For example, within a switched-mode power supply, the voltage output of the circuit can be located in front of a switching module with switches for chopping up the direct voltage. It is within the scope of the invention that the voltage output of the circuit is not identical to the output of the device. In particular, it is provided that the circuit for limiting the inrush current makes up the first section within the device. The term “voltage output” preferably relates to the output of the rectifier or the output of the circuit according to the invention. The term "first pole" preferably means the positive pole. The term "second pole" preferably means the negative pole. According to one embodiment, however, the first pole can also represent the negative pole and the second pole the positive pole. The word “connected” preferably means an indirect or direct connection. The control element is preferably a current-controlled and / or a voltage-controlled switch. The control element very preferably comprises a transistor or field effect transistor (FET) or a metal oxide semiconductor field effect transistor (MOSFET). It is preferred that the control input corresponds to the gate or the base. It is useful that the power path corresponds to the source-drain path or the collector-emitter path. The control element preferably comprises a series resistor at the control input.

According to a particularly preferred embodiment, the power path of the control element is assigned to the output side of the rectifier. It is particularly preferred that the power path is arranged between the first pole and the second pole of the voltage output or is assigned to the output side of the rectifier. The control input is preferably assigned to the input side of the rectifier. It is advantageous that the control input is connected to the input side of the rectifier or to the voltage input. It is preferred that the control element is arranged between the voltage input and the voltage output or between the input side and the output side of the rectifier.

According to a particularly preferred embodiment, the power path of the control element is connected in series with the output capacitor. The power path is preferably connected to the output capacitor, in particular directly. It is very preferred that the output capacitor is arranged together with the power path in parallel to the output side of the rectifier. Particularly preferably, the sensor element and the actuating element together with the control element form a closed control loop. The control loop is advantageously designed such that the sensor element detects a state of the power path. It is advantageous that the sensor element converts the detected state of the power path according to a sensor function and transmits a sensor signal to the actuating element. It is very preferred that the actuating element converts the sensor signal according to an actuating function and outputs an actuating signal to the control input of the control element. The state of the power path detected by the sensor element preferably allows a conclusion to be drawn about the current in the power path. It is of great advantage if the control signal determines a voltage or a current at the control input of the control element. It is very particularly preferred that the control loop is designed in such a way that the actuating signal reacts in opposite directions to the state of the control element. If, for example, the current in the power path increases, it is particularly advantageous that the actuating element outputs an actuating signal which allows the control element to reduce the current in the power path. The control loop is expediently designed in such a way that when the current or status parameter falls, the actuating element outputs an actuating signal which the control element allows the current or the status parameter to increase. The control loop is particularly preferably designed in such a way that the charging current of the output capacitor runs linearly or essentially linearly.

According to one embodiment, the sensor element and the adjusting element are components of an isolation amplifier and preferably an optocoupler. It is advantageous if the sensor element is a light-emitting diode, in particular a special light-emitting diode or laser diode. It is advantageous if the control element is a phototransistor or a photodiode. It is preferred that the optocoupler is a Darlington optocoupler or has a Darlington circuit. The isolation amplifier can also transmit the signals inductively or capacitively. In addition, it is possible for two transistors, in particular two bipolar transistors or two MOSFETs, to form the sensor element and the actuating element.

According to a particularly preferred embodiment, the control group comprises an auxiliary capacitor, the auxiliary capacitor being connected in parallel to the output capacitor. A first connection of the auxiliary capacitor is expediently connected to the first pole. A second connection of the auxiliary capacitor is preferably connected to the power path or to the sensor element. It is very preferred that the auxiliary capacitor and the output capacitor are connected to one another on the control element side via a synchronization diode. The orientation of the synchronization diode depends on whether the first pole is the positive pole or the negative pole. If the first pole is the positive pole, the diode is connected to the auxiliary capacitor on the cathode side. If, on the other hand, the first pole is the negative pole, the synchronization diode is connected to the output capacitor on the cathode side. It is advantageous that the auxiliary capacitor is connected in series with the sensor element. A limiting resistor is preferably connected in series with the auxiliary capacitor or with the limiting resistor, in order preferably to help determine the time constant of the charging of the auxiliary capacitor.

According to a particularly preferred embodiment, a sensor switch is connected to the sensor element, the sensor switch being designed to deactivate the sensor element. The term “deactivate” preferably means switching off or bridging the sensor element. It is very it is preferred that the sensor switch is a transistor and preferably a field-effect transistor and particularly preferably a MOSFET. It is preferred that a power path of the sensor switch is connected in parallel to the sensor element. Preferably, a control input of the sensor switch, in particular special via a blocking resistor, is connected to the control input of the control element. It is preferred that the control input of the sensor switch is connected to a switch capacitor and / or to a switch diode. The power path of the sensor switch is expediently connected to the switch capacitor and / or the switch diode. The sensor switch and the switch capacitor and / or the switch diode can be referred to as a sensor switching group. It is useful if the switch capacitor is arranged in parallel with the switch diode. The switch diode is preferably a breakdown diode and in particular a Zener diode or a suppressor diode. It is preferred that the switch diode and / or the switch capacitor are connected to the power path of the control element and / or to the sensor element and / or to the output capacitor. It is particularly preferred that the control group or the sensor switch or the sensor switch group is designed so that the sensor switch is opened when the output capacitor has a certain state of charge, for example at least 50% or at least 70% or at least 90% of the full charge of the output capacitor sators has reached. It is particularly advantageous if the open power path of the sensor switch short-circuits the sensor element so that the control loop is disabled.

It is particularly preferred that the control group comprises a control capacitor for providing a control voltage at the control input of the control element. It is preferred that the control capacitor with the Control input is connected. It is preferred that the control capacitor is connected to the power path or the second pole. It is advantageous that the control capacitor is arranged parallel to the "control input - second pole" path. It is particularly preferred that the control capacitor is arranged parallel to the actuating element. The control capacitor expediently has a control side and a pole side, the control side being connected to the control input and the pole side being connected to the second pole.

According to a preferred embodiment, the circuit has an output voltage sensor. The output voltage sensor preferably comprises a breakdown element, which can be designed in particular in the form of a Zener diode or a suppressor diode. The output voltage sensor is preferably arranged between the control input of the control element and the second pole. It is advantageous if the output sensor is connected in parallel to the control capacitor and / or to the actuating element. It is advantageous if the output sensor is connected on the control input side to the non-return valve and / or to the series resistor or the control element. The output voltage sensor advantageously comprises a breakdown switch, which is preferably a semiconductor component, more preferably a transistor and particularly preferably a bipolar transistor. The breakdown switch has a control input which is preferably connected to the breakdown element. The breakdown switch preferably comprises a power path which is connected at one end to the second pole and at the other end to a signal output for a power good signal. It is preferred that the output sensor or the breakdown element limit the voltage level at the control input of the control element to a maximum value. It is advantageous that the output sensor or the breakdown switch when the maximum value of the voltage level is present Control input of the control element supplies a power good signal at the signal output. The power good signal can be a start signal for subsequent circuit sections of the device. For example, the power-good signal can trigger an operation of the switching module of a switched-mode power supply for chopping up the DC voltage.

It is very preferred if the circuit has an input block. The input block advantageously comprises an input voltage sensor and / or a non-return valve. It is very preferred that the non-return valve is in the form of a diode. The non-return valve is preferably arranged between the voltage input and the control input. The non-return valve is advantageously designed so that the AC voltage present on the input side preferably leads to a charging of the control capacitor, so that the control element preferably opens the power path when a limit value of the voltage on the control capacitor is exceeded. If, for example, an N-channel MOSFET (enrichment type) is the control element and the source connection is connected to the second pole in the form of the negative pole, an increasing voltage above a threshold voltage or a threshold value is required at the control input so that the control element can The power path opens and connects the output capacitor to the second pole or the negative pole. In this case, it is useful that the non-return valve in the form of a diode is connected to the control input on the cathode side and to the voltage input on the anode side.

The input voltage sensor preferably comprises an input switch and / or an input resistor and / or an input capacitor. The input switch is preferably a transistor and more preferably a bipolar transistor. It is very preferred that a power path of the input Switch is arranged in parallel with the control capacitor. A control input of the input switch is preferably connected to the voltage input and / or to the non-return valve. It is preferred that the power path of the input switch is connected to the non-return valve. It is expedient that the input resistor and the input capacitor are connected in parallel to one another and form a timing element of the input voltage sensor. The timing element of the input voltage sensor is connected between the control input of the input switch and the second pole. It is very preferred that the input voltage sensor is designed such that a voltage drop at the voltage input opens the power path of the input switch, so that the control capacitor is discharged via the input switch.

It is preferred that the circuit has an input voltage divider, the input voltage divider being assigned to the input side of the rectifier and preferably a tap of the input voltage divider being connected to the control input of the control element. The input voltage divider or the tap is expediently arranged between the voltage input and the non-return valve. It is advantageous if the input voltage divider has two resistors, the two resistors expediently being of the same size. It is preferred that the tap is arranged between the two resistors of the voltage divider. The tap is preferably connected to the non-return valve or to the control input of the control element or to the control input of the input switch. The voltage divider is preferably designed in such a way that it protects against electric shock if the connections of the voltage input of the device or the circuit are touched. It is very preferable to have the voltage divider between the voltage input and the Rectifier is arranged. It is advantageous that the voltage divider is connected between the two poles of the voltage input.

It is possible that the circuit has a capacitor or several capacitors on the input side and / or on the output side of the rectifier for radio interference suppression. It is preferred that for radio interference suppression two capacitors are located on the input side of the rectifier and two capacitors are located on the output side of the rectifier. It is possible that a current-compensated throttle for vibration damping is arranged on the output side of the rectifier. The vibration damping of the current-compensated choke is particularly advantageous in a switched-mode power supply in which the switching module with the circuit breakers for chopping up the DC voltage generates undesired vibrations.

In order to achieve the object or objects mentioned at the outset, the invention teaches a device, in particular a power supply or switched-mode power supply, comprising a circuit according to the invention for limiting the inrush current. The power supply preferably comprises a transformer. The power supply preferably has a switching module chopping a DC voltage for the purpose of transmission by means of a transformer. The power supply unit or switched-mode power supply unit can comprise a microcontroller. The power supply or switched-mode power supply preferably has a connection for connection to a terminal.

To achieve the object or objects mentioned at the beginning, the invention teaches the use of the circuit according to the invention in a power supply unit, in particular a special switched-mode power supply unit, and / or in a frequency converter and / or in a DC voltage converter. The invention is explained in more detail below with the aid of several exemplary embodiments using several circuit diagrams and block diagrams. Show it

1 shows a circuit diagram of the circuit according to the invention in a first embodiment,

FIG. 2 shows a block diagram of the exemplary embodiment according to FIG. 1, FIG. 3 shows a block diagram of a second exemplary embodiment,

Fig. 4 is a block diagram of a third embodiment and

5 shows a circuit diagram of a fourth exemplary embodiment.

In Fig. 1, a first embodiment of the circuit according to the invention for inrush current limitation is illustrated using a detailed circuit diagram. The circuit according to FIG. 1 may, for example, be part of a power supply unit for a smartphone and corresponds to an input section of the power supply unit. Two alternating voltage connections AC together form a voltage input 1 in the form of two conventional pins for insertion into a socket. The circuit according to the invention for limiting inrush current also includes a voltage output 2 with a first pole 2a (positive pole) and a second pole 2b (negative pole). In this embodiment, for example, the voltage input 2 is followed in the reading direction by a further module of the switched-mode power supply, not shown here, and preferably a switching module for chopping up the direct voltage for the purpose of subsequent transformation. In another exemplary embodiment, the circuit according to the invention may be integrated in or connected to a frequency converter be, the frequency converter is used, for example, to supply a three-phase motor.

The first section of the circuit according to the invention according to the first exemplary embodiment comprises in the reading direction a first radio interference suppression module 10, a voltage divider 9, a rectifier 3 and a second radio interference module 11. With the exception of the voltage divider 9, the modules 10, 3, 11 are known. In the case of the rectifier 3, it may be a bridge rectifier which generates pulsating positive half-waves. These half-waves are already somewhat smoothed by capacitors C3, C4. In addition, the capacitors C3, C4 - as well as capacitors C1 and C2 and a current-compensated choke L - are used in a known manner for radio interference suppression. This radio interference suppression is particularly useful in view of the switching module, not shown in FIG. 1, for chopping up the direct voltage, in order to reduce the load on frequency bands of radio traffic (in particular radio and television transmissions). While the first radio interference suppression module 10 is connected in the reading direction between the voltage input 1 and the rectifier 3, the second radio interference suppression module 11 is located in the reading direction between the rectifier 3 and the voltage output 2.

The voltage divider 9 with resistors R1 and R2 is preferably connected in the reading direction between the voltage input 1 and the rectifier 3, the resistors R1 and R2 expediently being of the same size. A center tap is located between the two resistors R1 and R2, which represents the provision of an auxiliary voltage for the circuit according to the invention. The voltage divider 9 also has the task of a To provide protection against electric shocks if the connections of voltage input 1 are touched.

In this embodiment, the center tap of the voltage divider 9 is followed by an input block 8, a control group 4, an output voltage sensor 7, a control element T3 together with a series resistor R5 and an output capacitor C9. The input block 8 comprises a non-return valve D1 in the form of a diode and an input voltage sensor. The input voltage sensor has an input switch T1 in the form of a bipolar transistor, an input resistor R3 and an input capacitor C5. The output voltage sensor 7 has a breakdown element D2 in the form of a suppressor diode, a breakdown switch T2 in the form of a bipolar transistor and a signal output 15 for a power good signal.

The control group 4 comprises a control capacitor C6 and an isolating amplifier 5, 6 in the form of an optocoupler with an actuating element 5 comprising a phototransistor and with a sensor element 6 in the form of a light emitting diode. The control group 4 of this exemplary embodiment further comprises a sensor switching group 12 with a sensor switch T4 in the form of a MOSFET, a switch capacitor C7 and breakdown element D3 in the form of a suppressor diode. In addition, the control group 4 according to the exemplary embodiment of FIG. 1 comprises from top to bottom an auxiliary capacitor C8, a synchronization element D4 in the form of a diode, a limiting resistor R6 and a blocking resistor R4. The individual elements or components from FIG. 1 can also be represented in blocks according to FIG. 2. The operation of the circuit according to the invention according to the execution example of FIG. 1 is as follows. The non-return valve D1 only lets through the positive half waves, so that the control capacitor C6 is charged and the potential of the control input 14 rises. As soon as the potential of the control input 14 exceeds the threshold value of T3 at the gate, the power path of T3 is opened in the form of the drain-source path, whereby the charging of the output capacitor C9 in the form of an electrolytic capacitor begins. In parallel with this, a significantly smaller current flows from the second pole 2b through the sensor element 6, as a result of which the auxiliary capacitor C8 is also charged. Because of the synchronization element D4, C8 and C9 are charged synchronously.

At the beginning of the charging phase of C9, a relatively large amount of current is drawn from the output capacitor C9. As a result, the current through the sensor element 6 or through the light diode is correspondingly large, which is why a relatively large amount of light is emitted. As a result, the actuating element 5 in the form of the phototransistor receives a relatively large amount of light, as a result of which the collector-emitter path of the actuating element 5 conducts well. This has the consequence that the charging speed of the control capacitor C6 is braked from the start. The time course of this current flow is almost linear, so that in particular no current peaks arise. The potential of the control input 14 continues to rise slightly after the threshold value of the control element T3 has been reached, so that the control element T3 increasingly opens, and the non-linear curve of the charge saturation of the output capacitor C9 is compensated and an almost linear curve of the charging current is achieved. It should also be emphasized that in particular the timing element from the auxiliary capacitor C8 and the limiting resistor R6 determines the time constant of the charging of the output capacitor C9 via the sensor element 6, whereby its charging speed is practically freely selectable. The breakdown element D2 of the output voltage sensor 7 breaks through above a limit value or maximum value of the potential of the control input 14, so that the breakdown switch T2 is opened. Then the collector output of the breakdown switch T2 is open and supplies the power good signal for subsequent sections of the switched-mode power supply. They are now informed that the target voltage has been reached at voltage output 2 and can now start operating.

In parallel with the capacitors C8 and C9, the switch capacitor C7 is also charged, the time constant for this process depending on the capacitance of the switch capacitor C7 itself and also on the blocking resistor R4. This increases the voltage at the gate of the sensor switch T4 until the threshold value is also reached here. Then the sensor element 6 is short-circuited due to the open drain-source path of the sensor switch C4, so that no more current will flow through the sensor element 6 in the future. As a result, this also ensures that the control element 5 no longer receives any light from the sensor element 6, so that the control element will always remain closed in the future and the control loop from the control element T3, the sensor element 6 and the control element 5 is inactivated. This ensures that the control capacitor C6 can no longer be discharged via the actuating element 5, so that the potential of the control input 14 always corresponds to the value of the breakdown voltage of the breakdown element D2. The breakdown voltage of the breakdown element D2 is greater than the threshold value of the control element T3, so that the control element T3 is permanently open and the output capacitor C9 is permanently connected to the second pole 2b. If, for example, supply voltage fluctuations or individual shaft failures occurred in the AC mains on, the potential of the control input 14 would remain constant due to the non-return valve D1 and the breakdown element D2 at the level of the breakdown voltage of the breakdown element D2.

Only when the voltage input 1 is disconnected from the socket does the input capacitor C5 discharge through the input resistor R3, which causes the potential at the base of the input switch T1 of the voltage input sensor to drop. As a result, the control capacitor C6 is discharged, so that the potential of the control input 14 drops and the circuit according to the invention is put into the starting state. Due to the drop in the potential of the control input 14, the breakdown voltage of the breakdown element D2 is also undershot, so that the breakdown switch T2 blocks and a power good signal is no longer present. This deactivates subsequent circuit sections, for example the switching module. Due to the drop in the potential of the control input 14, the drain-source path of the control element T3 is also blocked, so that the current state of charge of the output capacitor C9 is preserved and does not have to be discharged when the AC source is reconnected, but rather has to be recharged from its current state of charge can.

In Fig. 3, a second embodiment is shown, which potentialver reverses to the first embodiment of FIG. 2 is designed. In this case, the second pole 2b is the positive pole, while the first pole 2a is the negative pole. The third embodiment of FIG. 4 largely corresponds to the first embodiment of FIG. 2, but differs in terms of the voltage source. Accordingly, the voltage input 1 of the circuit according to FIG. 4 is connected to a direct voltage source, so that the voltage divider 9 of this exemplary embodiment only has a resistor R has. The rectifier 3 of this exemplary embodiment is not designed as a bridge rectifier, but rather as a simple diode.

FIG. 5 shows a fourth exemplary embodiment which is identical to the first exemplary embodiment according to FIG. 1 with the exception of the adjusting element 5 and the sensor element 6. Thus, in FIG. 5, the sensor element 6 is implemented via a MOSFET T5, a suppressor diode D5 and a gate resistor R7. The adjusting element 5, however, is formed from a bipolar transistor D6 and two resistors R8 and R9.

As soon as the control element T3 is opened in the manner already described, the voltage-controlled transistor T5 is also opened via the rising voltage at R7. As a result, a voltage drops across the resistors R8 and R9, so that the current-controlled transistor T6 is opened, as a result of which the potential at the control input 14 drops and the charging current is limited. The suppressor diode D5 limits the gate voltage of the voltage-controlled transistor T5 to a maximum value. The voltage-controlled transistor T5 in the form of a MOSFET can, for example, also be replaced by a bipolar transistor. The bipolar transistor T6 can also be exchanged for a MOSFET. This shows that an isolating amplifier or optocoupler is not absolutely necessary for the control loop. List of reference symbols:

Claims

Patent claims:

1. A circuit for limiting the inrush current, comprising a voltage input (1), a voltage output (2) for a load and a rectifier (3), an input side of the rectifier (3) with the voltage input (1) and an output side of the rectifier ( 3) is connected to the voltage output (2), an output capacitor (C9) being arranged between a first pole (2a) and a second pole (2b) of the voltage output (2) or the output side of the rectifier (3), wherein the circuit comprises a control group (4), the control group (4) having a control element (T3) for limiting the charging current of the output capacitor (C9), the control element (T3) having at least one control input (14) and a two-pole power path, the control group (4) having an actuating element (5) for influencing the control element (T3), the actuating element (5) being connected to the control input (14), the control group (4) being a sensor element ent (6), an output of the sensor element (6) being connected to an input of the adjusting element (5), which means that an input of the sensor element (6) is assigned to the output side of the rectifier (3).

2. A circuit according to claim 1, wherein the power path of the control element (T3) is assigned to the output side of the rectifier (3).

3. A circuit according to claim 1 or 2, wherein the control element (T3), the sensor element (6) and the adjusting element (5) form a closed control loop.

4. Circuit according to one of claims 1 to 3, wherein the control group (4) comprises an auxiliary capacitor (C8), the auxiliary capacitor (C8) being connected in parallel to the output capacitor (C9).

5. Circuit according to one of claims 1 to 4, wherein a sensor switch (T4) is connected to the sensor element (6), the sensor switch (T4) being formed to deactivate the sensor element (6).

6. Circuit according to one of claims 1 to 5, wherein the control group (4) comprises a control capacitor (C6) for providing a control voltage at the control input of the control element (T3).

7. Circuit according to one of claims 1 to 6, wherein the circuit has an output voltage sensor (7), wherein the output voltage sensor (7) preferably comprises a breakdown element (D2).

8. Circuit according to one of claims 1 to 7, wherein the circuit comprises an input block (8), the input block (8) advantageously having an input voltage sensor (T1, R3, C5) and / or a non-return valve (D1).

9. Circuit according to one of claims 1 to 8, wherein the circuit has an input voltage divider (9), wherein the input voltage divider (9) is assigned to the input side of the rectifier (3) and preferably one Tap of the input voltage divider (9) is connected to the control input of the control element (T3).

10. Circuit according to one of claims 1 to 9, wherein the power path of the control element (T3) is connected in series with the output capacitor (C9).

11. Circuit according to one of claims 1 to 10, wherein the sensor element (6) and the adjusting element (5) are components of an isolation amplifier and preferably an optocoupler.

12. Circuit according to one of claims 1 to 11, wherein the control element (T3) comprises a transistor, preferably a field effect transistor and particularly preferably a metal oxide semiconductor field effect transistor.

13. Circuit according to one of claims 1 to 12, wherein the circuit has one or more capacitors (C1, C2, C3, C4) on the input side and / or the output side of the rectifier (3) for radio interference suppression and / or wherein on the output side of the rectifier (3) has a current-compensated choke (L) for damping vibrations.

14. Device, in particular power supply unit or switched-mode power supply unit, comprising a circuit for limiting the inrush current according to one of claims 1 to 13.

15. Use of the circuit according to one of claims 1 to 13 in a power supply unit and / or in a frequency converter and / or in a DC voltage converter.