EP3915186A1 - Dc-dc converter with bridge circuit for voltage-free switching, and associated method - Google Patents

Abstract

The invention relates to a bridge circuit (101) having: a first (A) and a second (C) high-side switch; a first (B) and a second (D) low-side switch; a transformer (T1) having a primary coil (T1_A) and a secondary coil (T1_B); a coil (T3_B); a current impressing device (T3_A); wherein: the first high-side switch (A) and the first low-side switch (B) are connected in series at a first bridge connection (105) in order to form a first bridge branch (107); the second high-side switch (C) and the second low-side switch (D) are connected in series at a second bridge connection (106) in order to form a second bridge branch (108); the first (107) and second (108) bridge branches are connected in parallel at a first (102) and a second (103) input connection; the secondary coil (T1_B) has a first (110a) and a second (110b) output connection; the primary coil (T1_A) and the coil (T3_B) are connected in series in order to connect the first bridge connection to the second bridge connection; and the current impressing device (T3_A) is configured to impress a predefined current into the coil (T3_B) at a predefined point in time.

Description

DC-DC CONVERTER WITH BRIDGE CIRCUIT FOR POWERless SWITCHING AND RELATED METHOD

Field of the Invention

The invention relates to the technical field of bridge circuits. In particular, the present invention relates to a bridge circuit, a DC-DC converter with the bridge circuit, a method for operating a bridge circuit and a programming element.

Background of the Invention

In order to operate bridge circuits effectively, the aim is to switch them as far as possible when there is no voltage. This type of switching is called ZVS (Zero Voltage Switching) switching. In order to keep the circuits simple, the challenge with the ZVS is to optimally set switching times and not to use complex additional circuits, such as measuring circuits, which give feedback to the control system for the switches.

In particular when operating a bridge circuit in a high-voltage DC network, as is the case, for example, in the high-voltage circuit

If electric cars are used, high losses can occur due to the high voltages used if the switches are not switched at the right moment.

It may be considered an object of the present invention to enable effective ZVS switching. Summary of the invention

Accordingly, a bridge circuit, a DC / DC converter with the

Bridge circuit, a method for operating a bridge circuit and a program element specified.

The object of the invention is characterized by the features of the independent

Claims specified. Exemplary embodiments and further aspects of the invention are specified by the dependent claims and the following description.

According to one aspect of the invention, a bridge circuit is provided. The bridge circuit has a first and a second high-side switch, a first and a second low-side switch, a transformer with a primary coil and a secondary coil, a coil and a current injection device. In this bridge circuit, the first high-side switch and the first low-side switch are connected to a first bridge connection in a series circuit in order to form a first bridge branch. In addition, the second high-side switch and the second low-side switch are connected to a second bridge connection in a series connection in order to form a second bridge branch.

The first and second bridge branches are each connected to a first and a second input connection in a parallel circuit, the secondary coil having a first and a second output connection. The primary coil and the coil or inductor are connected in series to the first

To connect the bridge connection with the second bridge connection. The

Current impressing device is set up to impress a predetermined current into the coil at a predetermined point in time.

According to a further aspect of the present invention, a DC / DC converter (direct current / direct current converter) with the bridge circuit according to the invention is described. According to yet another aspect of the present invention, a method for operating a bridge circuit is specified, the method comprising operating the switches of the bridge circuit in such a way that a predetermined circuit is operated

Point in time a predetermined current is impressed into the coil by the current impressing device.

According to another aspect of the present invention, a program element is described, comprising a program code, which is set up, when it is executed by a processor, to carry out the method for operating a bridge circuit.

According to yet another aspect of the present invention, a

computer-readable storage medium provided on which a program code

is stored which, when executed by a processor, carries out the method for operating a bridge circuit.

A floppy disc, hard disk, USB (Universal Serial Bus) storage device, RAM (Random Access Memory), ROM (Read Only Memory) or EPROM (Erasable Programmable Read Only Memory) may be used as a computer-readable storage medium. An ASIC (application-specific integrated circuit) or an FPGA (field-programmable gate array) can also be used as storage medium, as well as SSD (solid-state drive) technology or a flash-based storage medium. A web server or a cloud can also be used as a storage medium. As a

Computer-readable storage medium may also be viewed as a communications network, such as the Internet, which may allow program code to be downloaded. It can be a radio-based network technology and / or one

wired network technology can be used.

The use of a current injection device can ensure that existing energy is withdrawn from a switch of the bridge circuit in order to switch the switch in a state that is as currentless as possible. In particular, the

Current injection device ensure that when the energy is withdrawn, the switch when Switching is supported. This impressed current can also enable a live node to be discharged quickly and thus favor ZVS switching. For example, the output capacitance or parasitic capacitance of the first high-side switch is discharged and the output capacitance or parasitic capacitance of the first low-side switch is charged and the first bridge connection moves from an upper potential to a lower potential or ground potential, which then causes ZVS switching can be achieved for the first low-side switch.

According to another aspect of the present invention

Current injection device a further coil, which in combination with the coil form a second transformer or an additional transformer.

According to a further aspect of the present invention, a coupling between the further coil and the coil is low. In other words, the coupling between the further coil and the coil is less than the coupling between the primary coil and the secondary coil of the transformer. For example, a coupling of the further coil and the coil has a lower magnetic coupling factor than the coupling between the primary coil and the secondary coil of the transformer.

This low coupling factor of the ZVS transformer may allow a current to be injected into the coil, but not load the circuit with a high voltage. In other words, the low coupling factor of the ZVS transformer can provide a large leakage inductance of the ZVS transformer, which allows magnetic energy to be stored, but which has only a minor influence on the output performance of an inverter. If, contrary to the low coupling factor, the ZVS transformer, which has the coil and the current shaping device, had a high or good coupling factor, the remaining inductance would not be sufficient to summarize or withdraw the energy which is required for ZVS conditions. The high leakage inductance or the leakage inductance of the ZVS transformer is used to generate or impress a current that is required to achieve ZVS (Zero Voltage Switching). The magnetic coupling factor between the further coil and the coil may have a value of approximately 0.9 with a maximum possible value of 1. Coupling factors of typical power transformers may be in the range of 0.995. The main transformer may also have a magnetic coupling factor of approximately 0.995 and thus be significantly larger than the magnetic coupling factor of the ZVS

Additional transformer, which is about 0.9.

According to another aspect of the present invention, the bridge circuit further has a high-side capacitor and a low-side capacitor. The high-side capacitor and the low-side capacitor are connected in series at a third bridge connection to form a third bridge branch, the third bridge branch being connected to the first and second input connections and the further coil being the third bridge connection connects to at least one of the first bridge connector and the second bridge connector. The two capacitors, the high-side capacitor and the low-side capacitor, keep the ZVS transformer at a medium voltage potential. The magnetic core of the ZVS transformer is balanced and the first high-side switch and the first low-side switch can be controlled with a symmetrical switching pattern.

According to yet another aspect of the present invention, the

Bridge circuit on a synchronous rectifier. The synchronous rectifier is connected to the first and second output terminals.

In contrast to diodes, the synchronous rectifier can be actively controlled. The control can be designed such that the synchronous rectifier is short-circuited for a predeterminable duration during a free-running phase of the bridge circuit or the phase-shifted full bridge. By short-circuiting the synchronous rectifier during the free- _{running phase of} the bridge circuit, the current in the ZVS additional transformer can be increased, in particular an additional current can be impressed in a coil of the ZVS additional transformer T _zvs . This additional current can be used to switch the ZVS and / or ZCS to the respective one To enable switching phase belonging to the high-side switch and / or low-side switch by this switch making a transition from one switching state to the other, essentially without a voltage being present across this switch.

According to yet another aspect of the present invention, the

Bridge circuit a control device which is connected to each of the first and second high switches and low switches. The control device is set up to operate the switches in such a way that the predetermined current is impressed into the coil by the current impressing device at the predetermined point in time. Secondary switches may also be used for impressing, for example switches of a secondary rectifier and / or the synchronous rectifier. This switching of the switches may take place during a freewheeling phase.

The control device may, for example, be set up in such a way that it operates switches on the secondary side such that the predetermined current is impressed into the coil. The switch or switches on the secondary side may be switches of a rectifier on the secondary side and / or a synchronous rectifier. The secondary switch and / or the plurality of secondary switches may be implemented using MOSFET components. The level of the impressed current may be able to be determined indirectly through the time period for which the one, the two and / or the plurality of secondary switches are switched simultaneously and thus the one, the two and / or the plurality of secondary coils are short-circuited. This short-circuiting of the secondary coil and / or the plurality of secondary coils may take place during one freewheeling phase of one of the high-side switches and / or the low-side switches.

According to yet another aspect of the present invention, the control device is further configured to operate the and / or the high-side switch and / or low-side switch in such a way that the predetermined current is impressed into the coil when the predetermined one Time the current through the coil is below a predetermined threshold. Brief description of the figures

Further exemplary embodiments of the present invention are described below with reference to the figures.

1 shows a block diagram of a DC-DC converter with a bridge circuit according to an exemplary embodiment of the present invention.

Fig. 2 shows a block diagram of a DC-DC converter with a bridge circuit and two main transformers according to an exemplary embodiment! of the present invention.

3 shows a block diagram of a DC-DC converter with a bridge circuit and a main transformer with center tap according to an example

Embodiment of the present invention.

4a shows diagrams of various signal profiles of a PSFB without using the additional transformer according to an exemplary embodiment of the present invention.

4b shows diagrams of different signal profiles of a PSFB using the additional transformer according to an exemplary embodiment of the present invention.

5a shows a section from diagram 4a according to an example

Embodiment of the present invention.

5b shows a section from diagram 4b according to an example

Embodiment of the present invention. Detailed description of exemplary embodiments

The representations in the figures are schematic and not to scale. In the following description of FIGS. 1 to 5b, the same reference numbers are used for identical or corresponding elements.

In this text, the terms "capacitor" and "capacitance" as well as "coil" or "choke" and "inductance" may be used interchangeably and, unless stated otherwise, should not be interpreted restrictively.

The term "high-side" may refer to a connection to a live potential. The term "low-side" may refer to a connection with a reference potential.

1 shows a block diagram of a DC-DC converter 100 with a bridge circuit 101 according to an exemplary embodiment of the present invention. With this switching arrangement, good switching conditions for the switching devices or switches A, B, C, D of a phase-shifted full bridge (PSFB) 101 or one can

Bridge circuit 101 can be achieved with phase-shifted switching behavior. These good switching conditions may be achieved if an essentially voltage-free switching of the respectively active switches A, B, C, D can be established. The voltage-free switching, ZVS (Zero Voltage Switching) or zero voltage switching makes it possible to avoid high switching losses, which can arise in particular when switching high voltages due to parasitic elements in the switches A, B, C, D, since energy is stored in these parasitic elements may be against which the switches A, B, C, D must be worked against. As an alternative or in addition to the ZVS switching, currentless switching can also be achieved (ZCS, Zero Current Switching).

A bridge circuit 101 can be used, for example, in a DC-DC converter 100 to convert an input voltage V _in to an output voltage V _out . V _in and V _out are direct voltages. On the way from the input to the output, the input DC voltage Vin from the

Bridge circuit 101 converted to an alternating current (AC) and converted back to the output DC voltage by means of rectification. Particularly in applications that are used in an OBC unit (on-board charging unit) of an electric or hybrid vehicle, it may be necessary to convert very high voltages V _n (HV) into conventional on-board voltages V _out of approximately 12 V, for example can be used to operate a radio. The voltages Vin are provided, for example, by the DC intermediate circuit of the electric vehicle. Alternatively, the voltage Vin can also come from the on-board component of a charging device.

The DC-DC converter 100 can be used instead of an alternator of a vehicle for providing the on-board voltage 12V. In an example, the 12V on-board voltage is not generated directly by mechanical work, but rather by the DC-DC converter 100 converting the high voltage (HV) of an HV battery (DC voltage, DC) into the 12V on-board voltage of an EV (electrical vechicle) or PHEV (plug - converts into hybrid electric vechicie). The HV is in a load circuit or intermediate circuit

Energy supply system of a vehicle available. The energy extracted from the HV circuit is used to charge a 12V on-board supply battery on which the 12V

Consumers are connected. If the 12V battery were not constantly recharged from the HV circuit via the DC-DC converter, the connected consumers would discharge the 12V battery, similar to when using a mechanical one

Power supply the alternator would fail.

The OBC unit (not shown in FIG. 1) that Vin supplies is used to charge the DC link's HV battery. The voltage of the HV battery can be Vin = 400V or 800V. The OBC unit itself obtains its energy, for example, from an AC power supply (also not shown in FIG. 1), the so-called main, for example via an AC or three-phase connection. Therefore, the voltages Vin of the HV direct voltages (DC) can be in the range of 400V - 800V or in a range less than 800V. The bridge circuit 101 or the

Bridge circuit 101 is configured in such a way that it can deal with voltages of corresponding size and fluctuation range, The voltage V _in is supplied to the bridge circuit 101 via a first input terminal 102 and a second input terminal 103rd The first input connection 102 may be referred to as a high-side connection 102 and the second input connection 103 may be referred to as a low-side connection 103. These input connections 102, 103 form a parallel connection of the first 107 and the second 108 bridge arm. The first bridge branch 107 is formed from a series connection of the first high-side switch A and the first low-side switch B. The second bridge branch 108 is formed from the series connection of the second high switch C and the second low-side switch D. The first high-side switch A has the control connection 104a, the first low-side switch B has the control connection 104b, the second high-side switch C has the control connection 104c and the second low-side switch D has the control connection 104d. The control connections 104a, 104b, 104c, 104d are connected to a control device, not shown in FIG. 1, which controls the switches ABC, D in a phase-shifted manner. The control is carried out by means of the control device such that essentially the first high-side switch A and the second low-side switch D are switched simultaneously. And so that the second high-side switch C and the second low-side switch B are switched simultaneously. It can also be provided that a pause or dead time is provided between the switching of the switches belonging to one another, during which no switch is activated and during which all switches are open. A switching ratio d of 50% is essentially provided during the switching process, so that essentially the switch combinations A, D and B, C are active for the same length of time.

The switch pairs A, D and and B, C, which are switched essentially simultaneously, are arranged diagonally to the coil T and / or the coil T3 _B , so that the switching of the switch pairs A, D and / or B, C for reversed current flow through the coil T3 _B ensures. The control pattern for the phase-shifted control of the switches ABC, D essentially corresponds to a control pattern or control scheme used for a phase shift switching full bridge (PSFB).

4a diagrams of various signal profiles, in particular voltage profiles and current profiles depending on a switching behavior of a bridge circuit 101 and / or a synchronous rectifier SR1, SR2 without using the

Additional transformers according to an exemplary embodiment of the invention. FIG. 4a shows a diagram 400a of an extended signal curve of a PSFB with subsequent A / B without using the additional transformer according to one

exemplary of the present invention. Circuit diagram 400a shows a selection of signal profiles for operating a phase-shifted full bridge converter _circuit without a ZVS transformer T _zvs . In the switching phase 406a or transition phase 406a, as shown at point 405a, the current I _T1A through the primary coil decreases, since in this phase the leading branch C, D is switched. During the freewheeling phase II, which follows the phase 406a, the current I _T1A continues to decrease since here, by simultaneously connecting switches B, D, a circuit with switch B, switch D and primary coil T 1 _{A is} formed. The current decreases due to the current flow circulating in this freewheeling circuit. The circuit formed in the freewheeling phase II behaves like an RL circuit, which is formed from the line resistances and the primary coil T _1A . The line resistances cause losses caused by the current which flows to break down the stored magnetic energy. Due to the losses that occur during the freewheeling phase II, the magnetic energy stored in the inductance T1 _A during the switching or transition phase 404 a of the lagging branch 107 (lagging leg transition) A, B is lower than in FIG

Transition phase 406a of leading leg transition C, D. Consequently, there is insufficient magnetic energy available to fully discharge the parasitic capacitances of switches A, B of the lagging branch, for example the parasitic capacitances of a MOSFET switch A, B

4b shows diagrams of different signal profiles, in particular voltage profiles and current profiles depending on a switching behavior of a bridge circuit 101 and / or a synchronous rectifier SR1, SR2 when using the

_{Additional transformer} T _zvs according to an exemplary embodiment of the present invention. In particular during the final phase II _{B of} the freewheeling phase II, in which the low-side switches B 104b and D 104d and synchronous rectifiers SR1 and SR2 are switched simultaneously and form the low-side freewheeling circuit 104b, 104d, T3 _B and T1 _A , The current I _T1A continues to _rise after the switchover phase 406b because the T3 _B winding of the T _zvs transformer is short-circuited, while at the same time half the input voltage is present at the T3 _A winding of T _zvs . The increase in the primary current I _{T1A continues} until the switchover instant 404b of the lagging branch (lagging leg) MB, in which the low-side switch B 104b is switched off and the high-side switch A 104a is switched on. In other words, the increase in the primary current I _{T1A allows} the available magnetic energy to be increased during the lagging leg MB transition 404b. Thus, in the

Transition phase 404b of the switching of the trailing branch 107 A / B take place as a soft transition and a soft ZVS on switching of the switch A 104a is carried out.

The circuit diagram 400b shows, as drain-source voltage Vds _A, the profile of the voltage across the high-side switch A, that is, the profile of the voltage between the connection 102 and the bridge point 105 in the first bridge _branch 107 in the event that the additional transformer T corresponds accordingly Fig. 1 is used. Circuit diagram 400a shows as drain-source voltage Vds _A the corresponding voltage curve in the event that no additional transformer is used and thus only the primary coil T1A is solely responsible for switching the high switch A.

5a shows a section from diagram 4a according to an example

Embodiment of the present invention. This shows the switching phase 404a of the switch A of the trailing branch 107, in particular that

Control voltage of switch A, for example the gate voltage, if switch A is implemented as a MOSFET. The presence of a Miller Plateau 408a in the

Voltage curve A indicates that switch A cannot be discharged before the switching process is carried out, as can also be seen from point 407a of the curve of switching signal Vds _A , so that only hard switching takes place.

5b shows a section from diagram 4b according to an example

Embodiment of the present invention. 5 shows the ZVS switching process of switch A 104a. Assuming that switch A is implemented as a MOSFET, after the switching process in switching phase 404b at point 408b of the course of the gate voltage of switch A there is no Miller Plateau available. The voltage across the switch A 104a, for example the drain-source voltage Vds _A , has already dropped to 0 V when switching, as is illustrated at point 407b. So is a

complete ZVS switching and a smooth transition possible. The voltage curve of the control voltage of the input switches A, B when controlled by PWM (pulse width modulation) is shown in Fig. 4b when using the

_{Additional transformer} T shown _zvs .

In FIGS. 4a, 4b, 5a, 5b, signal A corresponds to gate voltage Vg at switch A 104a, signal B corresponds to gate voltage Vg at switch B 104b, signal C corresponds to gate voltage Vg at Switch C 104c and the signal D of the gate voltage Vg at the switch D 1Q4d. According to FIG. 1, the switches A 104a, B 104b, C 104c, D 104d are designed as normally blocking MOSFETS. This means that the application of a voltage to the respective gate or a high pulse in diagrams 400a, 400b corresponds to a closed switch through which current can flow. The use of self-conducting MOSFETs is also possible with inverse signs.

The signal SR1 corresponds to the gate voltage at the switch SR1. The signal SR2 corresponds to the gate voltage at the switch SR2.

The signal I _1A corresponds to the time course of the primary current through the coil T1 _A , in particular through the primary coil T1 _A.

The signal Vds _A corresponds to the time course of the drain-source voltage in

Switch A.

To simplify and clarify the influence of T _zvs , in particular the current increase in IT1A caused by this, the comparisons of FIGS. 4a, 5a, 4b, 5b show the same control patterns of switches A, B, C, D, SR1, SR2, although a different control _pattern would possibly be used in a PSFB without T _zvs according to FIG. 4a, since switches SR1, SR2 would not be short-circuited during freewheeling phase II in order to achieve ZVS switching. A bridge branch 107, 108 can be referred to as a leg 107, 108. In the PSFB switching method considered below, the second bridge branch 108 or leg CD 108, which has the switches C and D, is actuated as a leading branch 108.

The first bridge branch 107 or leg AB 107, which has the switches A and B, is driven as a lagging leg 107. The reverse control is also possible. In the PSFB switching method, leg CD 108 is phase-shifted compared to leg AB in order to control and / or regulate the output voltage Vout through the phase shift.

There are essentially four main phases or four main events in a PSFB switching process or control process. In the following, the differences from a standard PSFB that are caused by the use of the ZVS transformer T _zvs are _discussed .

Diagram 400a shows the control pattern for the switches A, B, C, D in the event that no ZVS transformer T _zvs , ie no coil T3 _A and no coil T3 _{B is} used. The time profile of the diagram 400a is essentially divided into four phases I, II, III and IV.

In contrast, the diagram 400b shows the control pattern for the switches A, B, C, D in the event that the ZVS transformer T _zvs is used, that is to say that the coil T3 _A and the coil T3 _{B are} used.

The control patterns for switches A, B, C, D are essentially the same for both diagram 400a and diagram 400b. Likewise, the division of phases I, II, III and IV. The control patterns of the synchronous rectifiers SR1, SR2 in diagrams 400a and 400b also correspond.

The individual phases I, II, III, IV are described below. 1. In phase I, switches B 104b and C 104c are switched on or activated ("B & C on"). This phase I is referred to as the energizing phase. During this phase, energy and / or power from the supply source V _in , for example the HV battery, which is connected to the nodes 102, 103, is transferred to the load (not shown in FIG. 1) at the connections 1 10a, 110b, V transferred _out . The current flow therefore takes place in phase I essentially via node 102, switch C, primary coil T1 _A and, if coil T3 _{B is} present, via T3 _B , via switch B in node 103.

1.a) In the following, phase I is considered for the case that no transformer T _{zvs is} provided, for the case that neither the coil T3 _A nor the coil T3 _{B is} present. The associated signal curves are shown in Figs. 4a, 5a shown. In the event that no transformer T _{zvs is} provided, after the switch B has been switched over due to the switchover of the lagging branch 107, which _heralds phase I, the current IT _IA in the main transformer T1 _{A increases} with a slope according to the formula:

Herein, i denotes the current I _T1A through the primary coil T1 _A , Vo 'denotes the voltage at the primary coil T1 _A , which is generated by the voltage V _out at the connections 110a, 110b at the load (not shown) via the transformer T1 to the primary side of the transformer T1 _A , is reflected. Vo 'corresponds to n * Vout, where n is the number of turns of the transformer T1. L ₁ 'denotes the inductance of the coil T1 _A. The dash at Vo 'and LT indicates that these are values that have been reflected in the primary coil T1 _A.

Current I _T1A flows during phase I from node 102, via switch C, via primary coil T1 _A , via switch B into node 103. Only during the further energy supply phase III "A&D on" described further below is the reflected output inductance L ₁ 'assumed to be very much greater than the leakage inductance of the transformer T1. The leakage inductance, which changes during the transfer from T1 _A to T1 _B 1 because it is a fictional quantity that does not correspond to a physical component, but it can be set using the transformer type.

1.b) Is additionally, as in Figs. 4b, 5b, the ZVS transformer T _zvs provided in series to T1, the ZVS ransformer T T _zvs increases with its coils T3 _A and T3 _B during Phase I, the voltage across the primary coil of the main transformer T1 _A T1. Thus, the slope of the current compared to formula (1) during the energy supply _phase I, when the ZVS transformer T _{zvs is} provided, changes to:

Here again i designates the primary current I _T1A , n _zvs the number of turns of the ZVS transformer T _zvs and 2 n _zvs the double number of turns, it being assumed that the coils T3 _A and T3 _{B have} the same number of turns n. In the energy supply phase I “B&C on”, only the change in the current I _T1A through the primary coil over time is affected by the provision of the ZVS transformer T _zvs and there are therefore essentially no changes compared to the control method of a PSFB circuit in which the ZVS transformer T _{zvs is} not provided. The greater the number of turns n _{zvs of} the ZVS transformer T _zvs , the less noticeable the ZVS transformer T _zvs .

2. The essentially simultaneous switching of the switches C, D of the leading branch 108 creates the transition phase 406a, 406b between phase I and phase II. During the transition phase 406a, 406b, the switch C 104c is switched off, via the in phase I the current I _T1A has been supplied and the switch D 104d is switched on (turn-off C / turn-on D). This transition phase 406a, 406b is called the transition of the leading bridge branch 108 (leading leg transition). 2.a) In the case of Figs. 4a, 5a, that no transformer T _{zvs is} provided, in order to achieve a smooth transition (ZVS) and to avoid hard switching, the parasitic capacitance of switches C and D is substantially completely discharged or charged during transition phase 406a. The energy used for discharging and / or charging is taken up or given off by the leakage inductance (not shown in FIG. 1) of the primary winding T1 _A in the form of magnetic energy. The magnetic energy of the leakage inductance of the primary winding T1 _{A is} determined as follows:

Herein DE _L denotes the change in energy in the primary leakage inductance L _lk of transformer T1 and I the current I _T1A through the primary coil T1 _A. However, this formula generally refers to the achievement of a ZVS condition using switches A and B and is not limited to phase II only. If there is sufficient current in the main transformer T1A, C and D are switched and ZVS can also be reached for these two switches. In order to achieve a smooth transition when switching A and B (ZVS), the condition should be fulfilled:

The change in the inductive energy stored in the primary leakage inductance of T1 should be greater than the change in the capacitive energy DE _c stored in the parasitic capacitances C _{mos of} the switches A and B. In other words, the leakage inductance on the primary side should be dimensioned such that this condition is met. Formula (4) expresses that the energy stored in the leakage inductance of the coil T1 _A must be greater than the energy stored in the parasitic capacitances C _{mos of} the switches C and D and the energy stored in the parasitic capacitance C _{tr of} the transformer T1 is stored. The switching operation “turn-off C / turn-on D” of the leading branch 108 during the transition phase 406a is usually a smooth transition, since the current I _{TA1 is} at its maximum value and the energy of the leakage inductance is sufficiently large to fully charge and discharge the capacitors C _mos and C _tr .

During the transition phase 406a, which follows the end of the energy supply phase I, the two switches C, D of the leading branch 108 (leading leg) 108 are switched over essentially simultaneously. There is only a small dead time between switching off C and switching on D. The closed switch C of the leading branch 108 is opened during the transition phase 406a and the open switch D of the leading branch 108 is closed “turn-off C / turn-on D”. Since switches A, B of trailing branch 107 are not yet actuated, switch A remains open and switch B is closed.

This asymmetrical and temporally different switching of the switches C, D of the leading branch 108 and the switches A, B of the trailing branch 107 results in the state of phase II during the transition phase 406a, in which the high switches A, C open and the Low switches B, D are closed at the same time. As a result of this switching behavior, a loop or free-wheeling loop is formed in the lower region in the vicinity of the low-side node 103. The primary coil T1 _A drives the current flowing during phase I through the primary coil T1 _A and via switch B, via node 103 and via switch D. Since the switches A, C are open and / or are opened during the transition phase 406a and since the current flowing further through T1 _A is sufficiently large to discharge parasitic charges from the bridge point 106 and thus from the switches C, D, both can Switch C and switch D are switched in the transition phase 406a essentially under ZVS conditions.

2.b) If the ZVS transformer is provided, the diagrams 400b result, as shown in Figs. 4b, 5b. During the transition phase 406b, these diagrams essentially do not differ from the transition process 406a from FIG. 2a), in which no ZVS transformer is provided. The control procedure is also essentially the same. However, in the case of the coil T3 _B connected in series with the primary coil T1 _A, the primary coil T1A drives the current I _T1A that has flowed during phase I via the primary coil T1 _A and the coil because of the drop in current T3 _B , as well as via switch B, node 103 and switch D. The freewheeling loop thus has the primary coil T1 _A , the coil T3 _B , the switch B, the low-side node 103 and the switch D.

3. Phase II following transition phase 406a, 406b is referred to as free-running phase II. During this freewheeling phase 11, the low-side switches B 104b and D 104d are switched on (B & D on), i.e. closed and the high-side switches A, C opened.

3.a) In the case of Figs. 4a, 5a that there is no transformer T _zvs , the two low-side switches B and D are switched on or closed during this phase II “B & D on” and the two connections 105, 106 of the primary coil T1 _{A of} the main transformer T1 connected to the input connection 103. Thus, both input connections 105, 106 of the transformer T 1 are at the same potential and there is no voltage across the transformer T 1. However, the primary coil T1 _A continues to drive the ITIA current. However, this current I _T1A through the primary winding T1 _A decreases exponentially according to an RL circuit - a circuit with coil and resistor -:

Here, l _{p is} the peak current that flows during the transition phase 406a “transition of the leading leg branch” from 2. following the energy supply phase I. The resistance value r _{ds, on} is the on-resistance of the Switch B or D, for example the MOSFET B or D.

The end of the freewheeling phase II determines the further transition phase 404a, which is characterized in that the switches A, B of the trailing branch 107 are switched in this further transition phase 404a. The high-side switch A is switched on or closed and the low-side switch B is switched off or opened. If the leakage inductance is too small and / or insufficient current I _{T1A flows} through the primary winding T1 _{A of} the main transformer T1, the lagging switching process can occur Branch 107 in the transition phase 404a no ZVS switching can be achieved. Because if one only increased the leakage inductance of T1, this would affect the output performance. Therefore, increasing the leakage inductance is essentially avoided. The current cannot be increased easily. However, the use of the ZVS transformer and the simultaneous activation of switches SR1 and SR2 during the free-running phase managed to increase the current

3.b) If, as in Figs. 4b, 5b provided, according to FIG. 1 of the ZVS (Zero Voltage Switching) transformer T _3A , T _3B additionally provided as a series connection to the transformer T1 between the bridge points 105, 106, there are differences in the actuation to a case in which the ZVS transformer T _{zvs is} not available. Because even if the leakage inductance is too small and / or insufficient current I _T1A would flow through the primary winding T1 _{A of} the main transformer T1, the current I _T1A through the primary winding T1 _{A of} the main transformer T1 can be increased if the transformer is used by means of the ZVS that during the free-running phase II before the transition phase 404b, the synchronous rectifiers SR1, SR2 are activated or closed simultaneously. Before the switches A, B of the lagging branch 107 are switched, the synchronous rectifiers SR1, SR2 are simultaneously activated or closed, as a result of which the output 110a, 110b and in particular the secondary coil T1B are short-circuited. This simultaneous activation of SR1, SR2 generates an additional current pulse in the primary coil T1 _A , which increases the current I _T1A and thus also results in an increase in the current through the ZVS transformer.

There is therefore no need to change the switching behavior of the synchronous rectifiers SR1, SR2 compared to FIG. 4a. The actuation of the switches SR1, SR2 according to FIG. 4a could be omitted for controlling a PSFB without a ZVS transformer and is only shown in FIG. 4a for a better comparison. It also shows that the current without ZVS transformer does not increase despite switching SR1, SR2.

However, if the ZVS transformer is present, the ZVS transformer can be used to simultaneously activate the synchronous rectifiers SR1, SR2 during the freewheeling phase II, II _B during the transition phase 404b ZVS to create conditions for switching switches A, B during transition phase 404b.

In other words, in order to _take advantage of the presence of the ZVS transformer T _zvs , "B & D on" is still during phase II, during which the low-side switches B and D bridge points 105, 106 on the primary side are the same Switch potential, the secondary-side coil T1 B of the transformer T1 is short-circuited by activating or switching on the two switches SR1 and SR2 of the output rectifier circuit 1 12 or of the secondary side 112, which are realized, for example, by means of MOSFET transistors. The part of the freewheeling phase II during which the synchronous rectifiers SR1 and SR2 are switched on at the same time, is located at the end of the freewheeling phase II and is _{denoted by Ila} or II _B. If the ZVS transformer T _zvs is _present , the current i _T1A decreases during this final phase II _B , as shown in FIG is, while without ZVS transformer T _zvs in the final phase _{Ila there} is an increase in current I _T1A , as can be seen in FIG. 4a.

In phase II, by activating the low-side switches B and D, a closed circuit is formed from the series connection of the ZVS secondary coil T3 _B , the primary coil T 1 _A and the two switches B and D. While this primary-side circuit is formed by simultaneously connecting the first low-side switch B and the second low-side switch D to the negative potentials with the bridge connections 105 and 106, the two switches SR1, SR2 of the secondary-side rectifier become shortly before the activation of the switches A, B of the lagging branch 107 activated during the final phase II _A , II _B. According to FIGS. 4a and 4b, only the first synchronous rectifier SR1 is activated in the final phase II _A , II _B , since the second synchronous rectifier SR2 is already activated.

The second synchronous rectifier SR2 is switched off before the further transition phase 404a, 404b, that is to say before the switches A, B of the lagging branch 107 are actuated. By actuating the two switches SR1, SR2 in the end phase II _A , II _B , the two connections of the secondary coil T1 _{B are} simultaneously connected to the same potential, for example to the ground potential and in this way, before the transition phase 404a, 404b of the trailing branch 107 a circuit is formed on the secondary side from the secondary coil T1 _B and the two switches SR1, SR2. In the circuit formed on the primary side by the switches B and D and the coils T3 _B and T1 _A , a freewheel is generated during phase II, since the collapsing magnetic field in the coils T3 _B and T1 _{A maintains} the primary current in and the leakage inductance L _Ik of Main transformer T1 receives its current from T _zvs , in particular from T3 _B , and the current through T1 _A continues to increase. The slope of the current in is calculated according to:

This (in terms of magnitude) current increase, which is additionally caused by the leakage inductance L _Ik and the secondary-side short-circuiting, can be seen in FIG. 4b at point 409b in the region of the end of the free phase II _B , while SR1 and SR2 are on. This current rise is before the transition phase 404b, while in the same area II of FIG. 4a, without a T _zvs , a drop in the current i _T1A can be seen. As a result of the additionally injected current rise due to the discharging of coils T3 _B and T1 _A , the magnetic energy which is stored in the leakage inductance L _Ik of T 1 increases until finally all the charge stored in the switches A, B of the trailing branch 107 into one magnetic energy of the leakage inductance L _{Ik is} converted, so that the switches A, B are essentially free of charges and the transition of the lagging branch A / B (lagging leg) 107 can be carried out under ZVS conditions. The current circulates through the leakage inductance. If the leakage inductance is too small, the current must be increased to ensure sufficient energy for ZVS conditions.

4. In the transition phase 404a, 404b between phase II and III, switch B 104b is switched off and switch A 104a is switched on (turn-off B / turn-on A). This phase 404a, 404b “turn-off B / turn-on A” is referred to as the transition of lagging leg 107 (lagging leg transition) A / B.

During the energy supply phases I and III, the current increases continuously, but in the opposite direction, so that the current at points 405a, 405b of 0A differs. The amount of current increase depends on the output power of converter 100. This current provides the ZVS conditions when switching the switches C, D of the leading branch 108.

During the freewheeling phases II and IV, the current decreases in the opposite direction. As can be seen in phases II _A and / or IV _A , the conditions for ZVS switching of A and B are not reached. Only if, as can be seen in phases II _B and / or IVB, a ZVS transformer is present and the switches SR1, SR2 are switched on at the same or the same time, ZVS conditions for switching A and B can be achieved. Alternatively or additionally, the leakage inductance of T1 could also be increased, which, however, can lead to losses in the output performance and therefore, when it is carried out, is carried out only to a small extent.

Thus, the current i can _1A after a sudden surname in the transition phase 406a, 406b, the current flow i _1A during the freewheeling phase II after the other until the end portion II _A, II _B is reached. Up to the end area II _A , II _B , the courses of the current i _1A of _FIGS. 4a and 4b correspond regardless of whether the ZVS transformer T _{zvs is} present or not

Essentially, only the current i _1A, which increases linearly during phases II _B and IV _{B, is used} for the ZVS switching of A and B when the ZVS transformer is present. The current pulse of the current i _1A , immediately after the switching of A and B, is an interaction with the parasitic capacitances of the circuit and can be neglected. The different influencing of the current flow I _{T1 A} with and without ZVS transformer T _zvs in the end area II _A, II _{B of} the freewheeling _phase II is responsible for the different behavior of a circuit with a ZVS transformer and without a ZVS transformer.

4.a) If the PSFB is used without a transformer T _zvs , as shown in FIG. 4a, the phase 404a of the transition of the lagging branch 107 is a critical phase, since it is related to the freewheeling phase II connects. Because, as can be seen in FIG. 4a as well as in FIG. 5a at reference number 407a, lies in Area of transition 404a of trailing branch 107 still has voltage across switch A while switch A is actuated. The actuation of the switch is shown at reference number 408a. In the example that the switch A is realized by a MOSFET, the switch A is activated at its gate in the area 408a, the entire voltage VDSA still being present at its drain-source connection in this time area.

As a result of the low current I _T1A during phase II _A, the magnetic energy stored in the inductor T1 A is not sufficient to completely discharge the parasitic capacitances of switches A and B and the primary coil T1A when a low load is on Output 1 10a, 110b is connected. Only with a large load would there be a current in the leakage inductance or leakage inductance which would be large enough to bring about ZVS conditions in the switches A, B of the lagging branch 107. In Fig. 4a, however, a low load is assumed and consequently hard switching of switches A and B occurs. Although the time range in which soft (ZVS) can be switched can be increased by increasing the value of the inductance of primary coil T1 _A . From a certain value, however, there is then the risk that the output capability of the DC-DC converter 100 or the performance of the converter output 110a, 110b is endangered.

5a shows the detail 404a in the region of the transition of the lagging leg transition 107 (lagging leg transition) in the event that no transformer T _{zvs is} provided.

4.b) If according to the embodiment of FIG. 1 of the invention of the ZVS

Transformer T _{zvs is} provided, as shown in Figs. 4b, 5b, by activating the switches SR1, SR2 during the final phase II _B, an increased current I _{T1A flows} during the final phase II _B. Because of the increased current flow in the final phase II _B , a smooth transition when switching A and B can be generated. The current i _T1A does not _stop at the point 405b, but continues to flow, in particular it continues to rise during the final phase Ile of the freewheeling phase II up to the point 409b. This increase in

Current flow is caused by the ZVS transformer, which affects the effects of the Switching the secondary-side switches SR1, SR2 or the secondary-side

Rectifiers SR1, SR2 reinforced on the primary side.

Here, the ZVS transformer T _zvs , _i.e. the combination of the coils T3 _A or T3 _B , has the function during the freewheeling phase II or IV and in particular in an end region II _B or IV _B, ie during the time interval during which the switches SR1, SR2 are activated at the same time and short-circuit the secondary coil T1 B to increase the primary current I _1A . Since the low-side switches B and D during the freewheeling phase II

are switched on, the voltage across T3 _{B is} kept at 0V during the freewheeling phase II. The voltage reflected in the primary coil from the secondary side is therefore also zero. However, the voltage across the coil T3 _A is half that

Input voltage ½ V _in . The voltage across T3 _B is kept at 0V during freewheeling phase II. While the voltage of the T3 _A winding is not equal to zero, the current increases linearly through the ZVS transformer T _zvs . This current is proportional to the time that SR1 and SR2 are activated and inversely proportional to the leakage inductance of T _zvs .

Since the voltage is kept at 0V via the low-side switch B during the freewheeling phase II, the switch B can be switched under ZVS conditions during the transition phase 404b immediately after the freewheeling phase II, in particular the low-side switch B can be switched under ZVS Conditions are turned off. After the switch B has been switched, the current additionally impressed by the coil T3 _A into the coil T3 _B is conducted into the connection node 105 between the switches A and B and thereby helps all the charge in parasitic elements from the high-side switch A and / or to remove all voltage across the high-side switch A and to provide ZVS conditions for switching the switch A. Thus, the high-side switch A of the lagging branch 107 can be switched by a short dead time after the low-side switch B of the lagging branch 107 under ZVS conditions, as shown in FIG. 5b. 5b shows that A is actuated after V _DS = 0 and thus the voltage across A is essentially zero volts.

In one example, it may apply that the two switches A, B of the lagging branch 107 are switched essentially simultaneously during the transition phase 406b. In another example, it may apply that the high-side switch A of the lagging branch 107 is switched in time after the low-side switch B of the lagging branch 107. In yet another example, the high-side switch A of the lagging branch 107 may be switched during phase III after the low-side switch B of the lagging branch 107, which is switched during the freewheeling phase II. In yet another example, the low-side switch B of the lagging branch 107 is switched before the second synchronous rectifier SR2 and the high-side switch A of the lagging branch 107 is switched after the second synchronous rectifier SR2.

The same applies when switching the switches A, B of the lagging branch 107 following the freewheeling phase IV. However, the current ITIA flows in the reverse direction during the energy supply phases III and the freewheeling phase IV compared to those of the energy supply phase I and the freewheeling phase II Coming energy supply phase III, switch A is switched on and switch B is switched off. The free-running phase IV begins with the switching of the switches C, D of the leading branch 108. A loop or free-running loop is formed in the upper region of the circuit at the high connection 109. The freewheeling loop has switch A, coil T3 _B , coil T1 _A, high-side node 102 and switch C. The voltage is also kept at OV in this freewheeling loop. Therefore, when leaving the freewheeling phase IV in the final phase IV _B , the switch A of the trailing branch 107, which is in the freewheeling loop, is again switched first. Since the voltage in this freewheeling loop is kept at 0V, switch A of the following branch can be switched under ZVS conditions. If this switch is switched, the current additionally generated by switching the synchronous rectifiers SR1, SR2 can be used to also switch the second switch B under ZVS conditions.

The switching behavior of the switches A 104a, B 104b, C 104c, D 104d is shown in Figs. 4a, 4b, 5a, 5b the same, regardless of whether the ZVS transformer Tzvs is present, as shown in Figs. 4b, 5b is required or does not exist, as shown in Figs. 4a, 5a is required. This switching behavior corresponds to the switching behavior of a phase shifted full bridge (PSFB), so that the ZVS transformer T _zvs can be retrofitted to every PSFB without changing the switching behavior.

The ZVS transformer Tzvs provides the current increase 409a in the final phase II _{B of} the freewheeling phase II or the current increase with the opposite sign in the

Final phase IVB of free-running phase IV.

During the freewheeling phase II and also during the freewheeling phase IV the is

Input voltage V _in the sum of the drain voltages Vds _A and Vds _B if the switches A and B are related as MOSFETs.

V _in = Vds _A + Fds _B

5b shows the detail in the area of the switching interval 404b or the transition phase 404b. At the switching time 407b of the switches A and B of the lagging leg branch (lagging leg transition), in the event that a transformer T _{zvs is} provided, the voltage Vds _A across the switch A has dropped to essentially 0 V, so that ZVS Switching is possible.

On the other hand, no ZVS switching is possible without a transformer T _zvs at the switching time 407a of the switch B, as shown in FIG. 5a.

The switches A, B are connected in the first bridge connection 105 and the switches C, D are connected in series in the second bridge connection 106. The first bridge connection 105 and the second bridge connection 106 are also connected via a series connection of the coil T3 _B and the primary coil T1 _{A of} the main transformer T1. The coil T3 _B may 1 _A of the main transformer to be construed as an additional coil T3 _B TO the primary coil T, since it can be used to increase the total inductance of the series connection of T1 _A and T3 _B. A high total inductance between the nodes 105 and 106 can improve the ZVS behavior of the bridge circuit 101. The additional coil T3 _B can with a current injection device T3 _A or

Current injection device T3 _{A can be} coupled. In the example in FIG. 1, the current injection device T3 _{A is} also a coil T3 _A. The coil _B T3 may be coupled to low magnetic coupling and thus form the additional transformer T _zvs or ZVS transformer T _zvs with the auxiliary coil _B T3. The ZVS switching of switches A and B can be achieved using a small transformer T _zvs with low magnetic coupling. The Stromeinprägeeinrichtung T3 _A forms the primary coil T3 _A of the auxiliary transformer T and the auxiliary coil _zvs T3 _A, the secondary coil T3 _A forms the additional transformer T · _zvs

By connecting the additional transformer T _{zvs in series} with the main transformer T1, good output capability for the phase shifted full bridge can be achieved. If the input voltage V _in falls below a predeterminable value, the DC-DC converter cannot generate a voltage which is able to supply a load connected to the output 110a, 110b, for example the output of the DC-DC converter manages it not to charge a 12V battery if that

Input voltage V _{in is} too low. If a current injection device T3 _A or a primary winding T3 _{A is} provided, which is coupled to a secondary coil T3 _B , which is connected in series with the primary coil T1A of the main transformer, this performance of the output 110a, 110b can be increased. The good

Output performance can therefore be characterized in that even with a low input voltage V _in , a load can still be supplied with a power at the output of the DC-DC converter, but this is then also due to the low power

Input voltage V may _be low.

This good output performance enables efficient battery applications, such as auxiliary DC-DC converters for electric vehicles (EV, electro vehicles) and for plug-in hybrid electric vehicles (PHEV) in which the

Voltage range V _in depending on the state of charge of the high-voltage battery (HV battery) connected to the high-side node 102 and the low-side node 103 can be large. All switches A, B, C, connected to the primary coil T1 _{A of} the main transformer T1

D are called primary switches. These can be realized with the help of MOSFETs A, B, C, D. In order to enable a ZVS for all primary switches A, B, C, D, the series connection of the auxiliary coil T3 _B with the primary coil T1 _{A is}

Main transformer T1 provided. The additional voltage at the additional coil T3 _B enables the good output capability at the output 110 to be achieved.

The voltage drop across T3 _B affects the performance of output 110 and the addition of a primary coil T3 _A which is magnetically coupled to T3 _B increases the performance of the output by increasing the voltage applied to the primary side of main transformer T1 becomes. T _{zvs has a} two-fold effect on increasing the performance of the output. On the one hand, the voltage applied to the primary coil T1 _{A of} the main transformer T1 increases by a value given by the formula. Due to the increased input voltage of the

The output voltage Vout = Vo results in:

Here n _{tr is} the number of turns of the main transformer T1.

A ZVS transformer T _zvs can reduce switching losses. With and without ZVS

Transformer T _{zvs it} takes a predetermined time until a primary voltage on the primary coil T1 _A also appears on the secondary coil T1 _B after this

Primary voltage has been applied to the primary coil T1 _A by transformer T1.

This delay arises because the primary current through T1 _{A is} only from one

Free-running state must transition into a state in which the output current is reflected at the primary coil T 1 _A. It is desirable to increase the stored magnetic energy in order to enable a long free run in which the

magnetic energy is broken down. If this magnetic energy were increased by providing a high inductance of T _1A , this would result in high switching losses (duty loss) lead. Using the ZVS transformer, the magnetic energy can be increased by providing a current without increasing the inductance of T _1A .

In other words, by providing the ZVS transformer T _zvs, a high magnetic energy can be stored in the system, but at a low level

Stray inductance of T _1A and thus the switching losses (duty loss) can be reduced. The storage of high magnetic energy is necessary in order to establish ZVS conditions for the transition of the switches A, B of the downstream branch 107, in particular if the switches A, B are implemented as MOSFETs. The ZVS transformer T _zvs is dimensioned in such a way that the magnetic energy that enables the ZVS transition of the trailing branch 107 is stored. Storing a higher magnetic energy essentially does not improve the switching behavior.

Since the ZVS transformer also has a leakage inductance, the ZVS transformer stores the magnetic energy in its leakage inductance. This magnetic energy is proportional to the peak current which flows through the ZVS transformer T _zvs . This peak current in turn is proportional to that

Time interval of the freewheeling phase during which the switches SR1 and SR2 are switched on simultaneously. The leakage inductance of the ZVS is determined in the design of the circuit so that it can take up enough energy to bring about ZVS conditions, and is difficult to change thereafter. Thus, the current required to create ZVS conditions is determined by the length of time controlled during which the switches SR1 and SR2 are switched on simultaneously during the freewheeling phase.

The time required for the transition between the two freewheeling states in the

Freewheeling phases II and IV is required can be regarded as a duty loss, which can be quantified as follows:

Here DI is the current difference between the current through T1 _A after the

Transition phase 404a, 404b turn-off B / turn-on A ”, ie after phase 404a, 404b of the transition of the trailing branch 107 (iagging leg transition) as described in FIG. 4 and the current through the primary coil T1 _A after the phase, if the primary voltage generated by the output current at the secondary coil T1 _B appears at the time that the output current is reflected at T1 _A (reflected output current).

DD is a time value that corresponds to a region along a time axis and f is the frequency of the PWM. DD Is the amount of time it takes for the current to change. This time duration DD should be as short as possible in order to achieve good output voltage capability.

The time interval DD increases with increasing load at output 110, since the current difference DI increases. This increase in switching losses DD can only take place in a limited range, since from a certain value they are so severe that output 110 is no longer able to provide the required output voltage Vout, for example for charging a 12V battery.

In a standard PSFB without T _zvs, the ZVS region, ie the range of input voltages v _in , in which ZVS is possible can be increased by the

Inductance of the primary coil T1 _{A is} increased, but then the increase

Duty cycle losses, since it is necessary to wait longer and longer until the high stored magnetic energy eliminates the parasitic voltages of switches A, B,

C, D has broken down to enable switching under ZVS conditions. If higher voltages v _{in are present} at switches A, B, C, D, namely higher parasitic voltages are also stored in the switches. However, it is desirable to operate the bridge circuit 101 with the highest possible PWM switching frequency f and thus with the lowest possible DD.

By providing T _zvs , the ZVS region, _i.e. the area of

Input voltages V _in , at which the DC-DC converter circuit 100 can still be operated efficiently, are increased by the current I _T1A , which during the Free-running phase II flowing through T1 _A is increased, while at the same time the leakage inductance L _{ik of} the transformer T1 is kept low. Although Dl is also increased, which increases switch-on losses, more magnetic energy can also be stored at the same time. If the current is increased, more magnetic energy can be stored, but power losses and / or line losses (RMS (Root Mean Square) losses) also increase. Soft switching or ZVS switching, however, reduces the line losses

In the circuit of FIG. 1, the primary side of the DC-DC converter is configured as a phase-shifted full bridge (PSFB) with an additional small transformer T _zvs to switch the zero-voltage _switching (ZVS) of the primary-side switches A and B of the lagging branch 107 to support. By means of the provision of the transformer T _zvs , the stored magnetic energy can be increased by impressing a current, in particular the current for neutralizing parasitic charges on the switches A, B, C, D and in particular on the switches A, B of the lagging branch 107. This neutralization can take place very quickly, so that the DC-DC converter 100 can work with a high switching frequency f.

For the primary-side switches C and D of the leading branch 108 (leading leg), a soft switching or a ZVS switching can essentially always be implemented in a PSFB, that is to say regardless of whether the ZVS transformer T _{zvs is} present or not.

The DC input voltage V _in corresponds to the voltage of the HV battery. The voltage V _in can range from 240V to 470V or 400V to 800V for

Applications with stronger HV batteries, such as those used in electric buses or high-performance electric cars. The fluctuation of the

Input voltage V _in can depend on the state of charge of the HV battery. The

Duty cycle ratio of the PWM used depends on the input voltage Vin present. However, other types of

_To provide switching devices A, B, C, D and other transformers T1, T _zvs if different voltage ranges are to be supplied, for example 240V to 470V or 400V to 800V. The same voltages are present at the primary switches A, B, C, D as at points 105, 106. As may fluctuate due to the change of the state of charge of the HV battery V _in in a wide range, for example, in the range of 240V to 470V, the DC-DC converter is a control circuit is provided (not shown in Fig. 1), the duty, the duty ratio ( ratio) of the control signal of the switches A, B, C, D changed to

Keep output voltage Vout at a constant value, for example Vout =

14.5V or Vout = 12V. However, if, for example, the input voltage drops from 470V to 240V, the pulse duty factor and / or the frequency of the switches A, B, C, D must be increased in order to ensure a stable and / or constant output voltage Vout.

The pulse duty factor is determined from the quotient of the duration of the energy supply phase I and the sum of the duration of the energy supply phase I and the duration of the

Free running phase II

Frequency for the control signals for A, B, C, D, SR1, SR2 remains constant and is not varied.

Since the energy transmission and / or power transmission via the main transformer T1 depends on the primary voltage, only a small amount of power and / or energy could be transmitted via the main transformer T1 due to the reduced primary voltage and the power that can be provided with the voltage Vout would be reduced.

In other words, there is a high in the connection circuit between 105 and 106

Inductance is desirable in order _to provide high magnetic energy at a high input voltage v _in for discharging the switches A, B of the lagging branch 107 and thereby to enable ZVS switching. However, if the inductance provided by the inductance of the primary coil T1 _{A was} increased further and further by series connection of an additional inductance T3 _B , the performance of the output voltage Vout or the output power would be reduced further and further, since the discharging of the switches A, B of the lagging branch 107 especially at high voltages v _in either could not be done quickly enough or not completely. This means that the DC-DC converter could only be operated in very low voltage ranges. While the primary side of the main transformer T1 is essentially on the

High voltage V _in of 240V - 470V is present on the secondary side T1 _{B of} the

Main transformer T1 a DC voltage of 14.5V or a voltage from the range of about 12 V to 15V, which is provided as output voltage V _out, for example, a radio or other consumer of the vehicle electrical system.

The provision of the additional transformer T _zvs compensates for the loss of

Output power by increasing the voltage applied to the primary coil T1 _{A of} the main transformer T1. Since the primary coil T1 _{A of} the main transformer T 1 is connected in series with the secondary coil T3 _{B of} the ZVS transformer, the performance of the output voltage (output voltage capability) increases. In order to compensate for this influence, the primary coil T3 _{A is} provided, which is connected between the switching node 105 and the fixed potential 11. A voltage which is applied to the primary coil T3 _A generates a voltage at the secondary coil T3 _B. This voltage on the secondary coil T3 _B increases the voltage on the primary coil T1 _A and ensures good output voltage performance.

In this way, the single-stage DC-DC converter with a large

Input voltage range are operated and ZVS are still guaranteed for all primary-side MOSFETs A, B, C, D. A single-stage DC-DC converter is a DC-DC converter that converts a first voltage level into a second voltage level only once without generating further intermediate voltage levels,

One side or a connection of the primary coil T3 _{A of} the additional transformer T _zvs is connected to the first bridge connection 105 and to one side of the additional coil T3 _B or the secondary coil T3 _{B of} the additional transformer T _zvs . The other side or the other connection of the primary coil T3 _{A of} the additional transformer T _zvs is connected to a third bridge _branch 109, which is formed as a series connection of two capacitors C1 and C2. This other side of the primary coil T3 _{A of} the additional transformer T _zvs is connected between the first capacitor C1 and the second capacitor C2. The third bridge branch 109 is connected to the first input connection 102 and connected to the second input connection 103 and is connected in parallel to the first 107 and second 108 bridge branch. The third bridge arm 109 ensures that a connection of the coil T3 _{A is kept} at a fixed or constant potential. A change in the voltage of the primary coil T3 _{A of} the ZVS transformer, which impresses a current into the secondary coil T3 _B , thus depends on a change in the potential of the bridge points 105 and 106. This also affects the by switching the synchronous

Rectifiers SR1, SR2 in the primary coil T1 _A reflected pulse on the

_{Additional transformer} T _zvs off.

On the output side, the series connection of a first synchronous rectifier (Synchronous Rectifier, SR) SR1 and a second synchronous rectifier (Synchronous Rectifier, SR) SR2 is connected in parallel to the secondary coil T1 _{B of} the main transformer T1. These are connected via a first output coil L1 and a second output coil L2 and an output capacitor C0 to the output 110 of the DC-DC converter 100, via which the output voltage V _{out is} provided. The synchronous rectifier SR1, SR2 is operated in such a way that the positive or negative half-wave which is induced in the secondary coil T1 _B is passed on to the smoothing capacitor Co with the same polarity, so that an output DC voltage V _{out is} generated.

2 shows a block diagram of a DC-DC converter 200 with a bridge circuit 101 and two main transformers T1, T2 according to an example

Embodiment of the present invention. In this configuration, the current on the secondary side of the main transformer 1 is divided into four coils L1, L2, L3, L4 and four synchronous rectifiers SR1, SR2, SR 1 ', SR2', which simplifies the efficiency of the circuit and the treatment of the current leaves. In addition, the main transformer T1 from FIG. 1 is divided into the two main transformers T1, T2. The primary coil T1 _{A of} the first main transformer is coupled to the secondary coils T1 _B and T1 _{C of} the first main transformer. The primary coil T2 _{A of} the first

Main transformer is with the secondary coils T2B and T2c of the first

Main transformer coupled. The output circuits 112a, 112b essentially correspond to the output circuit 112 from FIG. 1. However, in each of them

Output circuits 112a, 112b a secondary side of the two transformers T1, T2 utilized. Here, the synchronous rectifiers SR1 and SR1 'are operated in the same way and the synchronous rectifiers SR2, SR2' are operated in the same way.

3 shows a block diagram of a DC-DC converter with a bridge circuit and a main transformer with center tap according to an exemplary embodiment

Embodiment of the present invention. In this circuit variant, only one output coil L1 is provided.

The converter circuits shown in FIGS. 1-3 can be used both as a current doubler and as a center tap configuration on the secondary side. The center tap 301 is arranged on the secondary side of the main transformer T1 ”and connected to the two sub-secondary coils T1 _B “ and T1 _C “and the coil L1. The two sub-secondary coils T1 _B "and T1 _C " are also with the

Rectifiers SR1 "and SR2" connected. There is a ground connection between the rectifiers SR1 "and SR2", which is also connected to one of the output connections. The capacitor C0 is connected in parallel to the output.

The switching behavior with and without ZVS transformer is shown in Figs. 5a, 5b shown enlarged. The voltage curves Vds _A via the high-side switch A of the

lagging leg A / B is included for a case in which there is no load at the output 110, that is to say for the no-load case or idling. In order to impress an additional current into the secondary coil T3 _B , the synchronous rectifiers SR1 and SR2 are switched on simultaneously during the freewheeling phase II _{B of} the low-side switches B, D in order to provide a short pulse across the secondary coil T1 _B through the

To generate voltage drop to 0V output voltage, which is transferred to the primary coil T1 _A. The curve Vds _A from Fig. 5b corresponds to the case in which the

_Additional transformer T _zvs in series with the main transformer T1 is present, as shown in Fig. 1.

In phase III, switch A is closed and switch B is open. It can be seen that during the switching phase 404a, 404b the initial voltage of approximately V _in = 400V across A and B for the case that T _{zvs is} used, already before the switching phase 404b 0V has dropped, as indicated at position 407b, whereas the voltage V _in = 400V, in the event that T _{zvs is} not used, still has a voltage at the end of the switching phase 404a, as indicated at position 407a. Thus, when using the ZVS transformer T _zvs , ZVS switching is also possible in the _{no-load situation} . 1, the additional transformer T _{zvs is connected} in series with the main transformer T1 and helps to impress a current into the coil T3 _{B of} the additional transformer. Regardless of the load at output 110a, 110b, ZVS switching of high switch A 104a can be achieved. Because if there is no load at the output, the output current is 0A and the output _load R _{load is} undetermined. The output voltage V _out is regulated to a constant 14.5 V independently of the load, for example by changing the frequency and / or the duty cycle of the PWM switches A, B, C, D.

The energy supply phase III is followed by another free-running phase IV, namely the free-running phase of the high-side switches A 104a and C 104c. In this a free-wheeling circuit is formed from switch A 104a, C 104c, additional coil T3 _B and primary coil T1 _A.

With the phase-shifted full-bridge topology (PSFB) with an additional inductance T3 _B , which is connected in series with the transformer T 1, ZVS switching or soft switching can be achieved if the additional inductance T3 _{B is} part of a transformer T _zvs . The additional transformer T _zvs has a low coupling factor between the primary coil T3 _A and the secondary coil T3 _B - the low coupling is achieved, for example, by a magnetic core with a slot. Energy that can be used for ZVS can be _{temporarily stored} in the additional transformer T _zvs . Due to the low coupling of the ZVS

Transformer T _{zvs maintains} a leakage inductance in T _zvs , because the part of the magnetic flux that does not _couple into the secondary coil becomes noticeable as leakage inductance. This additional leakage inductance can be regarded as a further inductance, which is in series with T3 _B , even if the leakage inductance is not a tangible component. The size of the leakage inductance can also be influenced via the coupling factor. The leakage inductance can also store magnetic energy, which can then be converted back into an electrical current flow in order to support ZVS by discharging the bridge point 105. If a configuration with only one additional coil T3 _B without primary coil T3 _A or another coil T3 _{A is} used, _i.e. no complete transformer T _zvs but only one coil is connected in series with the main transformer, the output power of the DC-DC converter is reduced ., because a voltage drops across the coil T3 _B , which then reduces the voltage across the primary coil T1 _{A of} the main transformer. As shown in Figs. 4a and 5a is shown in phase 404a, with such a configuration with only one additional coil T3 _B , no real ZVS of the high-side switch A can be achieved even with a low load, since the current through the additional coil T3 _{B is} too low is. The coil T3 _B without T3 _A can only be used to achieve ZVS switching of the low-side switch B.

However, if a complete ZVS transformer T _zvs according to FIG. 1 is used, an additional current can be impressed into the coil T3 _B by switching the synchronous rectifiers SR1, SR2 and thus the effect of the leakage inductance of the

Transformer T _zvs magnetic energy can be used. Here the energy controlled by the current determined by the length of time

while the SR1 and SR2 are activated at the same time. The ZVS transformer T _zvs thus, by activating the synchronous rectifiers SR1, SR2 over a predeterminable period of time II _B, contributes to increasing the primary current I _T1A to such an extent as it does in the freewheeling _phase II for the ZVS switching of the switches of the lagging branch 107 is required.

By using an additional complete transformer T _zvs , a very efficient and cost-effective single-stage DC-DC converter can be realized.

In addition, ZVS switching can be achieved in the primary switches A, B, C, D, regardless of the load at output 110

Provide output power at output 1 10, which can be particularly important for applications with a large input voltage range Vin.

It may be considered as an idea of the present invention to increase the magnetic energy that is stored in the transformer T _zvs by the

Primary current ITIA is increased instead of increasing the inductance of the secondary coil T3 _B ZU, which would lead to a reduction in output power. Since the magnetic energy in the secondary coil is T3 _B according to formula (3), the increase is

of the primary current I _T1A more efficiently than the increase in the inductance of the secondary coil T3 _B. Since also the duty cycle loss on the ratio of the inductance

depend, increasing the current by impressing an additional current helps to keep the ratio essentially unchanged and to increase the stored magnetic energy without increasing the loss of switching.

Each additional series connected inductor, which as a real component or as

If leakage inductance is present, the output capability of the output 110 of the DC-DC converter 100 is reduced, for example in relation to a constant output voltage v _out to be provided as a function of a wide range of input voltages V _in . This reduction in output performance can have a negative effect if the output voltage V _out of converter 100 is to be regulated to a constant output voltage,

for example to 14.5V, and the input voltage varies within a wide range, for example in the range from 240 V to 470V, depending on the state of charge of an HV battery. This is because the inductance connected in series may be necessary to enable smooth switching under ZVS conditions. A series inductance would degrade the output performance of converter 100 because, for example, it is no longer possible to generate a constant output voltage of 14.5V when the input voltage Vin is at a lower range limit, e.g. at 240V in a range of 240V to 470 V, and at the same time ZVS conditions should be met. Because it would actually be desirable to get by without the T3 _{B series} inductor. But then no ZVS would be possible and the efficiency of the converter would be low.

In addition, since no large inductances are required for T3 _B , the size of a DC-DC converter can be kept small, even though it is operated at a high switching frequency f. The high switching frequencies are possible due to the rapid discharge of node 105 and are the same for switches A, B, C, D and are determined by the durations of phases I, II, III, IV. In addition, it should be pointed out that “comprising” and “having” do not exclude other elements or steps and “one” or “on” does not exclude a plurality. Furthermore, it should be pointed out that features or steps that are described with reference to one of the above exemplary embodiments can also be used in combination with other features or steps of other exemplary embodiments described above. Reference signs in the claims are not as

View restriction.

Claims

Claims

1. Bridge circuit (101) comprising:

a first (A) and a second (C) high-side switch; a first (B) and a second (D) low-side switch; a transformer (T1) with a primary coil (T1A) and a secondary coil (T1 _B );

a coil (T3 _B );

a current injection device (T3 _A );

wherein the first high-side switch (A) and the first low-side switch (B) are connected at a first bridge connection (105) in a series connection to form a first bridge branch (107);

wherein the second high-side switch (C) and the second low-side switch (D) are connected at a second bridge connection (106) in a series connection to form a second bridge branch (108);

wherein the first (107) and second (108) bridge branches are connected to a first (102) and a second (103) input connection in a parallel connection;

wherein the secondary coil (T1 _B ) has a first (110a) and a second (110b) output connection;

wherein the primary coil (T1 _A ) and the coil (T3 _B ) are connected in a series connection to the first bridge connection with the second

To connect bridge connection;

wherein the current injection device (T3 _A ) is set up to impress a predetermined current into the coil (T3 _B ) at a predetermined time.

2. Bridge circuit (101) according to claim 1, wherein the current _{injection device} (T3 _A ) is a further coil (T3 _A ), which in combination with the coil (T3 _B ) forms a second transformer (T _zvs ).

3. bridge circuit (101) according to claim 2, wherein a coupling between the further coil (T3 _A ) and the coil (T3 _B ) is less than the coupling between the primary coil (T1 _A ) and the secondary coil (T1 _B ) of the transformer ( T1).

4. Bridge circuit (101) according to claim 2 or 3, further comprising:

a high-side capacitor (C1);

a low side capacitor (C2);

wherein the high-side capacitor (C1) and the low-side capacitor (C2) are connected in series connection at a third bridge connection (1 1 1) to form a third bridge branch (109);

wherein the third bridge branch (109) is connected to the first (102) and second (103) input connection;

wherein the further coil (T3 _A ) connects the third bridge connection (1 1 1) with at least one of the first bridge connection (105) and the second bridge connection (106).

5. Bridge circuit (101) according to one of claims 1 to 4, further comprising:

a synchronous rectifier (SR1, SR2);

wherein the synchronous rectifier is connected to the first (110a) and second output connection (110b).

6. Bridge circuit (101) according to one of claims 1 to 5, further comprising:

a controller connected to each of the first and second high switches and low switches;

the control device being set up to operate the switches (A, B, C, D) in such a way that the predetermined current is impressed into the coil (T3 _B ) by the current impressing device (T3 _A ) at the predetermined point in time.

7. bridge circuit (101) according to claim 6, wherein the control device is further configured to operate the switches (A, B, C, D, SR1, SR2) so that the predetermined current is impressed in the coil (T3 _B ), if to that

predetermined time the current through the coil is below a predetermined threshold.

8. bridge circuit (101) according to claim 6 or 7, wherein the control device is further configured to operate secondary-side switches (SR1, SR2) so that the predetermined current is impressed in the coil (T3 _B ).

9. DC / DC converter (100, 200) with the bridge circuit (101) according to one of claims 1 to 8.

10. A method for operating the bridge circuit according to one of claims 1 to 8, the method comprising:

Operate the switch of the bridge circuit so that at a predetermined time a predetermined current from the

Current impressing device is impressed in the coil.

11. Program element, comprising a program code, which is set up, when it is executed by a processor, to execute the method according to claim 10 for operating the bridge circuit (101).