EP3738200A1 - Halbbrücke mit leistungshalbleitern - Google Patents
Halbbrücke mit leistungshalbleiternInfo
- Publication number
- EP3738200A1 EP3738200A1 EP18710367.6A EP18710367A EP3738200A1 EP 3738200 A1 EP3738200 A1 EP 3738200A1 EP 18710367 A EP18710367 A EP 18710367A EP 3738200 A1 EP3738200 A1 EP 3738200A1
- Authority
- EP
- European Patent Office
- Prior art keywords
- current
- bridge
- controller
- power
- power semiconductor
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Withdrawn
Links
Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/38—Means for preventing simultaneous conduction of switches
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0003—Details of control, feedback or regulation circuits
- H02M1/0012—Control circuits using digital or numerical techniques
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0048—Circuits or arrangements for reducing losses
- H02M1/0054—Transistor switching losses
- H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0083—Converters characterised by their input or output configuration
- H02M1/0093—Converters characterised by their input or output configuration wherein the output is created by adding a regulated voltage to or subtracting it from an unregulated input
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
Definitions
- the invention relates to a half-bridge with a GE in series switched first and second power semiconductor and a controller for the power semiconductor and a method for their operation.
- Half bridges are used as one of the most important basic circuits in almost every topology.
- Half-bridges comprise first and second power semiconductors, for example MOSFETs, which are connected in series.
- the two outer terminals of the power semiconductors form a first and second terminal of the half-bridge and may be connected, for example, to a DC voltage.
- the connection node of the two power semiconductors forms a third connection of the half-bridge and is connected via a line to a load. Power flow and power in the line may be directed both towards and away from the half bridge.
- the inductive load can be a dedicated inductive component such as a choke as well as a parasitic inductance, for example, the leakage inductance of a transformer sector or the line inductance. Often the inductive load is a mixture of both elements.
- the power In order to obtain desired current and / or voltage curves, the power must semiconductor of the half-bridge with high frequency and, if necessary, alternately switched on under different switching Z who the. A simultaneous switching must be avoided who the, in order not to generate a short circuit. This results in an approximately rectangular switching voltage across a respective switch and an approximately triangular current in the output line, whose frequency corresponds to the switching frequency.
- the switching times for the power semiconductors must be turned so that a desired current in the output line results, with the average of the current is usually desired as a controlled variable.
- the dead time t db is selected only as long as necessary, in order to reliably exclude a short circuit of the half bridge.
- the controller sets the duty cycle so that the desired average value for the current in the line results.
- Switching frequencies as can be achieved with new wide-bandgap semiconductors (GaN or SiC), are advantageous for the size and weight of inductors and capacitors, which often have to be used as filters and energy storers, but lead to increased demands the control, the switching losses and the EMC.
- GaN or SiC wide-bandgap semiconductors
- the half-bridge according to the invention comprises a first and second power semiconductors connected in series, a control for the power semiconductors, a line which starts from the connection node of the power semiconductors and a device for measuring the current in the line.
- the controller is configured to compare the current with an upper and a lower threshold value and switch off the first power semiconductor when the upper threshold value is reached and turn on the second power semiconductor after a first dead time has elapsed.
- the control is configured to switch off the second power semiconductor when the lower threshold value is reached and to switch on the first power semiconductor after the expiration of a second dead time.
- the current is measured in a line extending from the connection of the power semiconductors and compared with an upper and a lower threshold value, the first level being reached when the upper threshold value is reached switched off semiconductor device and turned on after a first dead time of the second power semiconductor, and switched off upon reaching the lower threshold of the second power half conductor and turned on after a second dead time of the first power semiconductor.
- the power flow of the half-bridge can extend from the line to the outer terminals of the power semiconductors or vice versa.
- the direction of current flow in the line may be directed away from the power semiconductors, which is considered herein to be a positive current flow or to the power semiconductors be directed, which is considered as a negative current flow.
- the device for measuring the current may be provided close to the half-bridge in the line.
- the device can also be arranged in a return line from a load to one of the outer terminals of the power semiconductors, thus measuring the current in the line despite the other placement.
- the inductive load or a part of the inductive load can be arranged between the location of the current measurement and the half-bridge.
- the power semiconductors is thus not selected a fixed switching frequency, the switching eita Z defines, but the circuit of the power semiconductor is made based on the measured current values and threshold values for the current.
- the switching eita Z defines, but the circuit of the power semiconductor is made based on the measured current values and threshold values for the current.
- this allows a direct choice of the average value for the current and a direct selection of the ripple current.
- the desired average value of the current is converted by the controller within only one period.
- this can be regarded as P-behavior, which enormously simplifies the control.
- the crizfre frequency In digital regulations, it is also possible by this method, the crizfre frequency to keep well below the switching frequency. In the previous method, this would lead to difficulties, because there is usually a more complex time response.
- the method thus makes it possible for the first time to control systems with very high switching frequencies (several 100 kHz up to the megahertz range) even without great computing power, for example with simple and inexpensive microcontrollers.
- this method is very robust with changing input and output voltages and thus creates far-reaching possibilities in system design.
- Another advantage is that in this half-bridge the ripple current can be selected independently of the Ar beitstician, which was not possible with previous methods.
- the current remains even with changes in the current behavior, for example by load changes, in the range of threshold values and thus at the average current value, which is specified as the setpoint, since the switching behavior of the power semiconductors by the threshold values and the current measurement adapts to the current behavior.
- changes to the default values For example, if the setpoint for the average current - and thus the thresholds - lifted, the current reaches the upper threshold later or the lower threshold earlier than before, which shifts the switching time points of the power semiconductors and the Strommit telwert raises to the new desired value.
- the controller may include a first and second comparator to which the measured current is supplied as a first input signal, wherein the upper threshold is supplied to the first Kompa capacitor as a second input signal and the lower threshold value is supplied to the second comparator as a second input signal.
- the controller may include a digital controller that passes the upper and lower thresholds to the comparators via a DAC (Digital to Analog Converter, DAC).
- DAC Digital to Analog Converter
- the outputs of the comparators can be converted into control signals for the power semiconductors in a modulator. This results in a simple structure, since now microcontroller are available in which D / A converters, comparators and the modulator are integrated. The method can therefore be realized without additional hardware.
- the controller can calculate the threshold values from predefinable values for the mean value of the current and for the ripple current in the output line.
- the threshold values can be calculated from the sum and difference of mean value and ripple current.
- values that are relevant only to the operation outside of the control must then be specified, while the controller generates the correct control values from it.
- the controller may be configured to use a minimum value for the ripple current.
- the controller can enforce that a minimum distance between the upper and lower threshold value is maintained, with this minimum distance equals the minimum value for the ripple current. It is thereby achieved that the switching frequency resulting from the distance between the threshold values, which increases when the ripple current decreases, does not become too high.
- the controller can use upper and lower threshold values that indicate different current directions.
- the lower threshold value can each be selected such that it has a different sign than the desired average current.
- a Umla the output capacitances of the power semiconductor is particularly advantageous. This, in turn, makes it possible to turn on the power semiconductors at low voltage, ideally without voltage.
- the ripple current is chosen so large ge that the thresholds take different signs, so identify different current direction.
- Half the amplitude of the ripple current is then greater than the average current value. It may also be sufficient to use the value 0 A as one of the threshold values. Also, it is a reloading of the output capacitances of the power semiconductors he allows and thus enables a voltage-free switching.
- the controller can calculate that threshold which characterizes a different current direction than the current direction of the mean value for the current, from the summed output capacitance of the power semiconductors, the inductance in the output line and the voltage at the input and output of the half-bridge.
- the controller can set the dead times so that a voltage-free switching of the power semiconductors happens. As a result, a significant reduction in switching losses is achieved. Furthermore, a significant improvement in the EMC properties is achieved as a resonant
- Umschwingvorgang takes place.
- the edges of the switching voltage are thus significantly flatter and rounded.
- the spectrum of such a switching voltage shows considerably lower amplitudes in the harmonics.
- the controller can calculate the dead times or select from a saved value table.
- the calculation may, for example, pas from the summed output capacitance of the power semiconductors, the inductance in the output line and the voltage at the input and output of the half-bridge.
- the half-bridge may comprise means for measuring the voltage across the first and second power semiconductors. A circuit can then be done on the basis of the measured clamping voltage, which made a safe resonant switching made light.
- First and second dead time are expediently different from each other ver, since the reloading of the capacity of the power semiconductor semiconductors takes place at different absolute currents and thus takes different lengths.
- the switching losses are almost completely eliminated by adjusting the dead times.
- the effi ciency of systems with the half-bridge can be significantly increased.
- ie wise switches based on GaN or SiC can thereby be achieved significantly higher switching frequencies than before.
- the improved EMC properties at the same time significantly reduce the filter effort, thereby enabling a more compact and cost-effective design.
- the half-bridge and the method relate to the power electronics.
- the switchable power semiconductors power is at least 10 W, in particular at least 100 W or at least 1 kW.
- the switched voltage is at least 50 V, in particular at least 100 V or at least 300 V.
- FIG. 1 shows an electrical converter
- FIG. 2 shows a circuit section with a half bridge with a first drive circuit
- FIG. 3 shows a circuit diagram and current profile
- FIG. 4 shows the half-bridge with a second drive circuit
- FIG. 5 shows the half-bridge with a third drive circuit
- FIG. 6 shows a simulated switching behavior
- FIG. 7 shows a measurement result of a circuit.
- Figure 1 shows a circuit diagram for an electrical converter 10, in which an embodiment of the invention is set is.
- the converter 10 corresponds in its construction of an interconnection of a boost converter and an inverter, wherein the respective outputs are connected in series.
- the converter 10 has first and second input terminals 11A, 11B for the input voltage, the first one
- Input terminal 11A is to be used as a positive pole. Furthermore, the converter 10 has a first and second output The first output terminal 13A also typically represents the positive pole.
- the transducer 10 further includes three electrical node points 12A, 12B, 12C, with reference to which the structure will be described.
- the first node 12A is directly connected to the second input terminal 11B and further connected to ground. Between the first input terminal 11A and the two th node 12B is a first inductance LI angeord net. Between the first output terminal 13A and the two th node 12B, a first semiconductor switch S1 is arranged to. Between the second node 12 B and the first node 12 A, a second semiconductor switch S2 is arranged.
- a first capacitor CI is arranged, which represents the output of the up-converter, which is formed from the ers th semiconductor switch Sl, the second semiconductor switch S2 and the first inductance LI.
- a third semiconductor switch S3 is arranged between the first input terminal 11 A and the third node 12 C. Between the second output terminal 13B and the third node 12C, a fourth semiconductor switch S4 is arranged. Between the third node 12C and the ers th node 12A, a second inductance L2 is angeord net.
- a second capacitor C2 is arranged, which represents the output of the inverting converter, which is formed from the fourth semiconductor switch S4, the third semiconductor switch S3 and the second inductor L2.
- the semiconductor switches S1 ... 4 in the converter 10 are in this case at play GaN switch. However, other switches such as MOSFETs or IGBTs can be used.
- the boost converter During operation of the circuit, the boost converter generates a positive voltage on the first capacitor CI. This positive voltage is inherently at least as large as the A ⁇ input voltage at the input terminals 11A, 11B. In turn, the inverter generates a negative voltage at the second output terminal 13B relative to the first node 12A.
- the series connection of the two capacitors CI, C2 the output voltage between the two output terminals 13A, 13B in the sum of the amounts of the two he testified voltages.
- the gear ratio that results for a given input and output voltage for the up-converter and the inverse converter are each halved.
- the converter 10 can also be operated so that the target voltages at the capacitors CI, C2 no longer DC voltages, but other waveforms, ie in general
- a control device not shown in the figures is provided, which is configured to switch the first to fourth semiconductor switch S1 ... S4 so that the desired voltage curve at the capacitors CI, C2 results.
- Such a desired voltage curve may, for example, consist of a sequence of half-waves or of a DC voltage with an additional modulation. Since, in addition, the generated voltages at the first and second capacitors CI, C2 add to the output voltage, a high amplitude at a moderate transmission ratio for the converter can be achieved even with a clamping voltage clamping.
- the sequence of half-waves for example, both the up-converter and the inverse converter can produce a phase and amplitude equal Halbwel lenverlauf. Then the amplitudes of the add Half waves in the output voltage at the output terminals 13A, 13B.
- the converter 10 has two half-bridges, each of which is directly connected to an inductor: the half-bridge of the first and second semiconductor switches Sl, S2 is connected to the first inductance LI and the half-bridge of the third and fourth semiconductor switches S3, S4 is connected to connected to the second inductance LI.
- Figure 2 shows a much simplified section of a circuit 100 having a half-bridge 102, which corresponds for example to the pair of first and second semiconductor switches Sl, S2 and / or the pair of third and fourth semiconductor switch S3, S4 of Figure 1.
- the half-bridge 102 may be part of the converter 10 or any other power converter, such as an inverter, rectifier, power supply or other converter, or it may be implemented on its own.
- the half-bridge 102 comprises two power semiconductors 108, 110 connected in series, such as MOSFETs. Frequently, the half bridge 102 is connected to the external terminals 104, 106 to a DC voltage 114, for example to the DC link of an inverter.
- the center terminal 112 between the power semiconductors 108, 110 is connected to an inductive load 116.
- the inductive load 116 is representative of all types of loads that may be only partially inductive and for such structures in which the inductive part of the load comes about, for example by a line inductance.
- the inductive load 116 can therefore just as well be a dedicated component as a pa rasitäres element or both together.
- the control of the power semiconductors 108, 110 is performed by a control unit 120.
- the control unit 120 includes a digital controller 122, first and second comparators 124, 126 and a modulator 128. It is possible that these elements are parts of a single microcontroller and thus constructed as a single device are. Likewise, however, these elements can also be partially or completely present as separate components.
- the control unit 120 comprises a current measuring device 130, which detects the incoming or outgoing from the center terminal 112 as a signal 131 signal.
- the first comparator receives as inputs the measured current signal 131 and a first maximum current threshold 132.
- the second comparator also receives as inputs the signal 131 for the measured current and a second threshold 134 for the minimum current.
- the thresholds 132, 134 are provided by the controller 122.
- the controller 122 may, for example, calculate these from default values for the average current and the current ripple. These default values can be specified from the outside, for example by a superordinate converter control, or be determined by the controller 122 itself.
- the output signals of the comparators 124, 126 are fed to the modulator 128.
- the modulator 128 converts these, as well as stored values for applied deadlines into drive signals for the power semiconductors 108, 110, which are passed to the respective gate driver.
- Passing into the modulator 128 is achieved that when he reaches the maximum current of the active power semiconductor 108, 110 is turned off and after waiting for the dead time to prevent a short circuit in the half-bridge 102 of the other power semiconductor 108, 110 is turned on.
- the active power semiconductor 108, 110 is also switched off and, after waiting for the dead time, the other power semiconductors 108, 110 are switched on.
- a resulting circuit diagram with a switching curve 202 for the upper power semiconductor 108, a switching curve 204 for the lower power semiconductor 110, a voltage curve Course 206 via the lower power semiconductor 110 is shown together with a resulting simplified current waveform 208 in Figure 3.
- the dead times 210, 212, which elapse after switching off a respective power semiconductor 108, 110, are greatly extended for better recognition availability.
- Figure 3 shows that the resulting current waveform is approximately triangular.
- the corresponding threshold 132, 134 is reached later and the shutdown of the corresponding power semiconductor 108, 110 happens later.
- the described procedure for the control of the power semiconductors 108, 110 thus operates in contrast to the known methods no longer with a fixed switching frequency. Rather, the instantaneous effective switching frequency results from the specifications of the threshold values 132, 134 or the specifications for the average current and the ripple current, the inductance 116 and the voltages 114, 117, which co-determine the current gradient.
- the instantaneous switching frequency may also fluctuate and may change if the default values are changed.
- half bridges according to the invention can be used particularly advantageously if he testified voltage waveform is a waveform, for example, the sequence of sine half-waves.
- the half bridges then generate the se not in the usual pulse width modulation with a fixed predetermined switching frequency, but continuously adapted duty cycle. Rather, the average current value, the
- FIG. 4 again shows an excerpt from a circuit 100 with the half-bridge 102, but with a modified structure of the control unit 120.
- the dead times 210, 212 are no longer stored permanently in the modulator, but instead predetermined by the controller 122.
- the dead times 210, 212 are no longer stored permanently in the modulator, but instead predetermined by the controller 122.
- the 212 can thus be changed by the controller 122 and adapted to the operating situation Be.
- Such an adaptation can be used to reduce the switching losses, in which a resonant reloading of the output capacitances of the power semiconductors 108, 110 is allowed.
- the minimum current threshold value 134 is set to a negative value, that is, to a value with a different sign than the mean value and the threshold value 132 for the maximum current. If the average value of the current is negative, the maximum current threshold 132 is set to a positive value, that is, again to a value having a sign other than the average and the threshold value 134 for the minimum current.
- I L is the lower threshold 134 for the current
- L is the value of inductance 116 in the output line
- the voltage 117 C is the summed output capacitance of the power semiconductors 108, 110, ie
- I H denotes the upper threshold value 132 for the current
- the respective threshold value is set to 0.
- the dead times 210, 212 may be determined by the controller 122 in various ways. The appropriate determination of the dead times 210, 212 allows the voltage-free switching of the power semiconductors 108, 110. Firstly, the dead times 210, 212 can be calculated or read from a pre-agreed and stored table (look-up) who the.
- L is the value of inductance 116 in the output line
- the voltage 117 C is the summed output capacitance of the power semiconductors 108, 110
- the control unit 420 comprises one voltage measuring device 402, 404 for each of the power semiconductor lines 108, 110.
- the signals 403, 405 of the voltage measuring devices 402, 404 are supplied to a third and fourth comparator 406, 408.
- the output signals of the third and fourth comparators 406, 408 are supplied to the modulator 128 and used by this, as the turn-on Z eit Vietnamese for the respective power half conductor 108, 110 to use the time at which the clamping voltage across the power semiconductor 108, 110th is low, so for example 1 V.
- FIG. 6 shows the waveform of the voltage 206, current 207, and the turn Z nits 502a, b for the first and second power semiconductor 108, 110 as a result of a simulation.
- the switching edges of the voltage 206 are noticeably flattened.
- the output capacitances are transposed before switching on the respective power semiconductor 108, 110.
- the flatter edges of the switching voltage mean significantly lower amplitudes of the harmonics and thus also ensure better EMC properties of the structure. Since the switching frequency can become very high at very low current ripple values, it is advantageous to realize a minimum value for the current ripple.
- the controller 122 is configured to implement and maintain this minimum value. This limits the switching frequency to a desired maximum.
- FIG. 7 shows the
- the current profile 604 is not exactly linear due to the measurement.
- an input voltage of 27 V was set to about 100 V at an output power of about 90 W.
- the switching frequency is 1 Mhz and the power semiconductor 108, 110 cooled only by free convection (ie without heat sink or fan), the working temperatures remained in a non-critical range below 60 ° C. This would not be possible hard switching.
Landscapes
- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Dc-Dc Converters (AREA)
Abstract
Description
Claims
Applications Claiming Priority (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
PCT/EP2018/054788 WO2019166072A1 (de) | 2018-02-27 | 2018-02-27 | Halbbrücke mit leistungshalbleitern |
Publications (1)
Publication Number | Publication Date |
---|---|
EP3738200A1 true EP3738200A1 (de) | 2020-11-18 |
Family
ID=61622531
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
EP18710367.6A Withdrawn EP3738200A1 (de) | 2018-02-27 | 2018-02-27 | Halbbrücke mit leistungshalbleitern |
Country Status (4)
Country | Link |
---|---|
US (1) | US11316423B2 (de) |
EP (1) | EP3738200A1 (de) |
DE (1) | DE112018007167A5 (de) |
WO (1) | WO2019166072A1 (de) |
Families Citing this family (2)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
DE102019201760A1 (de) * | 2019-02-12 | 2020-08-13 | Siemens Aktiengesellschaft | Halbbrücke mit Leistungshalbleitern |
FI129607B (en) * | 2021-05-18 | 2022-05-31 | Vensum Power Oy | SYSTEM AND METHOD FOR ZERO VOLTAGE CONNECTION IN A POWER CONVERTER |
Family Cites Families (17)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US4727308A (en) | 1986-08-28 | 1988-02-23 | International Business Machines Corporation | FET power converter with reduced switching loss |
DE3901034C1 (en) * | 1989-01-14 | 1990-07-19 | Danfoss A/S, Nordborg, Dk | Inverter |
US20070109822A1 (en) | 2005-11-14 | 2007-05-17 | Kan-Sheng Kuan | Zero voltage switch method for synchronous rectifier and inverter |
US8026704B2 (en) | 2008-06-06 | 2011-09-27 | Infineon Technologies Austria Ag | System and method for controlling a converter |
US8008902B2 (en) * | 2008-06-25 | 2011-08-30 | Cirrus Logic, Inc. | Hysteretic buck converter having dynamic thresholds |
GB0912745D0 (en) * | 2009-07-22 | 2009-08-26 | Wolfson Microelectronics Plc | Improvements relating to DC-DC converters |
EP2538535B1 (de) * | 2011-06-24 | 2021-08-11 | STMicroelectronics S.r.l. | Steuervorrichtung für einen Resonanzwandler |
JP6361479B2 (ja) | 2014-02-07 | 2018-07-25 | 株式会社デンソー | 電力変換装置 |
RU2695817C2 (ru) * | 2014-08-01 | 2019-07-29 | Филипс Лайтинг Холдинг Б.В. | Схема возбуждения нагрузки |
CN104242644B (zh) * | 2014-10-11 | 2017-04-12 | 成都芯源系统有限公司 | 用于开关转换器的控制电路和控制方法 |
US9948177B2 (en) * | 2015-01-07 | 2018-04-17 | Philips Lighting Holding B.V. | Power conversion device |
US9780636B2 (en) * | 2015-01-19 | 2017-10-03 | Infineon Technologies Austria Ag | Protection from hard commutation events at power switches |
CN106033929B (zh) | 2015-03-16 | 2018-11-02 | 台达电子工业股份有限公司 | 一种功率转换器及其控制方法 |
US9759750B2 (en) | 2015-08-03 | 2017-09-12 | Alex C. H. MeVay | Low loss current sensor and power converter using the same |
US9906131B1 (en) * | 2016-08-22 | 2018-02-27 | Ferric Inc. | Zero-voltage switch-mode power converter |
EP3297162B1 (de) * | 2016-09-20 | 2020-10-28 | Mitsubishi Electric R&D Centre Europe B.V. | Verfahren und vorrichtung zur schaltsteuerung eines ersten und eines zweiten leistungshalbleiterschalters |
US10218258B1 (en) * | 2018-01-09 | 2019-02-26 | Dialog Semiconductor (Uk) Limited | Apparatus and method for driving a power stage |
-
2018
- 2018-02-27 DE DE112018007167.7T patent/DE112018007167A5/de not_active Ceased
- 2018-02-27 EP EP18710367.6A patent/EP3738200A1/de not_active Withdrawn
- 2018-02-27 US US16/975,484 patent/US11316423B2/en active Active
- 2018-02-27 WO PCT/EP2018/054788 patent/WO2019166072A1/de unknown
Also Published As
Publication number | Publication date |
---|---|
US20210028690A1 (en) | 2021-01-28 |
DE112018007167A5 (de) | 2020-12-10 |
US11316423B2 (en) | 2022-04-26 |
WO2019166072A1 (de) | 2019-09-06 |
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