EP3553887A1

EP3553887A1 - Microwave antenna array communication system and communication method

Info

Publication number: EP3553887A1
Application number: EP17878581.2A
Authority: EP
Inventors: Wei Yao
Original assignee: ZTE Corp
Current assignee: ZTE Corp
Priority date: 2016-12-06
Filing date: 2017-12-06
Publication date: 2019-10-16
Also published as: CN108155479A; WO2018103677A1; EP3553887A4; CN108155479B

Abstract

Disclosed are a microwave antenna array communication system and communication method. The method comprises: replacing a related double-sided dual-polarized antenna with a phased antenna array, directly connecting each horizontally polarized radio frequency signal transmission device in a microwave transmission device to each subarray of a corresponding horizontally polarized antenna array in the phased antenna array respectively to transmit a horizontally polarized radio frequency signal to an opposite end, connecting each vertically polarized radio frequency signal transmission device to each subarray of a corresponding vertically polarized antenna array respectively to transmit a vertically polarized radio frequency signal to the opposite end; and controlling phase shifters of each subarray of the horizontally polarized antenna array and phase shifters of each subarray of the vertically polarized antenna array directly by a controller of the phased antenna array to configure a phase relationship between the multiple radio frequency signals transmitted by the horizontally polarized antenna array and vertically polarized antenna array independently of the physical distance between the antenna arrays and the installation precision.

Description

TECHNICAL FIELD

The present invention relates to the field of microwave communication and, in particular, to a microwave antenna array communication system and communication method.

BACKGROUND

Nowadays, the demand for wireless data transmission is rapidly increased, and wireless communication technology has been rapidly developed. Several ways that are commonly used to increase the transmission capacity and the transmission rate of the wireless communication system are frequency diversity, spatial diversity and the use of polarized antennas. To describe the technical background of the present invention, a microwave Light of Sight (LoS) Multiple Input Multiple Output (MIMO) system is illustrated as an example.
The microwave transmission has the advantages of high speed, high stability, small land resource occupation and the like. The microwave transmission generally uses LoS transmission. The microwave spatial multiplexing mainly uses a multi-antenna technology, also referred to as the MIMO technology, and in order to distinguish from the general MIMO technology, the multi-antenna technology of the microwave system is referred to as a Line of Sight MIMO (LoS MIMO) technology. The LoS MIMO technology greatly increases the throughput of the system in the relevant bandwidth. Currently, the manufacturer produces most the 2×2 LoS MIMO (2×2 may be understood as a generalized monopole antenna array, and for a bipolar antenna array, may be understood as 4×4 MIMO in the narrow sense). As the technology is enhanced, 4×4 (8×8 MIMO in the narrow sense for the bipolar antenna array) to N×N LoS MIMOs have been gradually applied.
The transmission capacity C of the MIMO system is obtained according to the Shannon theorem described below. $C = \log_{2} (\det (I_{n_{R}} + \frac{ρ}{n_{T}} H' H'^{H}))$
In the above formula (1), ρ is a signal-to-noise ratio on a receive side, H' is a normalized matrix of channel transmission characteristics, I_nR is an n_R ordered unit matrix and (·) ^H denotes Hermitian transformation. The maximum equivalent of the system transmission capacity is a maximized H'H'^H determinant, that is, under the system maximum capacity, a channel matrix needs to satisfy a Vandermonde matrix, whose any unitary transformation may ensure the maximum transmission capacity. Using 2×2 microwave LoS MIMO as an example, a channel matrix Vandermonde array is denoted below. $H_{van} = [\begin{matrix} 1 & 1 \\ 1 & - 1 \end{matrix}]$
The following is obtained after two times of unitary transformation. $H_{opt} = [\begin{matrix} e^{jπ / 2} & 0 \\ 0 & 1 \end{matrix}] [\begin{matrix} 1 & 1 \\ 1 & - 1 \end{matrix}] [\begin{matrix} 1 & 0 \\ 0 & e^{- jπ / 2} \end{matrix}]$
For the above 2×2 MIMO, any path of Tx will send a corresponding Tx signal in a path to a receive end Rx of the opposite end, and send a Tx signal having a phase delay of 90° to another receive end Rx. For example, referring to FIG. 1, a transmit end Tx1 respectively sends a Tx signal and a Tx' signal to the receive end Rx1 and the receive end Rx2 simultaneously, and the TX' signal is sent with the phase delay of 90° with respect to the Tx signal. When the method is popularized to other N×N LoS MIMOs, the process of achieving the maximization of the link transmission capacity is finally represented as the requirement on the layout spacing between the send and receive antennas. Still using the 2×2 LoS MIMO in FIG. 1 as an example, in a case that a communication distance D is known, the dual-polarized antenna spacing h is accurately measured and the antenna is laid out, so as to determine a corresponding phase shift angle. The correspondence between h and D is as follows. $h = \sqrt{λD / 2}$
In the formula (2), λ is a wavelength. In the related bipolar antenna array implementation shown in FIG. 1, the 2×2 LoS MIMO corresponds to the 4×4 MIMO in a narrow sense. The architecture of the related bipolar array 4×4 MIMO design scheme is shown in FIGS. 2-1 and 2-2. In FIGS. 2-1 and 2-2, a site 1 and a site 2 are one-hop 4×4 MIMO links. Using the site 1 as an example, H0, V0, H1 and V1 respectively represent four microwave transmission devices thereof (H represents that the device is connected to a horizontally polarized antenna, and V represents that the device is connected to a vertically polarized antenna), which all operate at the same radio frequency point. H0 and V0 form a Cross-polarization Interference counteracter (XPIC, corresponding to TX 1 in FIG. 1) group. Both H0 and V0 are connected to an Ortho-Mode Transducer (OMT) and then connected to a one-side parabolic dual-polarized antenna that is mounted on the iron towers shown in FIG. 2-2. The iron towers are laid out in a high/low station manner. The dual-polarized antenna spacing h satisfies the requirement in the above formula (2). H1 and V1 form another XPIC group (corresponding to the TX 2 in FIG. 1) whose connection manner is similar to that of the XPIC group formed by H0 and V0. Finally, both XPIC operating groups are combined into a 4×4 MIMO operating group, which is similar to the situation of the Site 2. When the microwave device uses a Frequency Division Dual (FDD) operating manner, it may be known that the transceiving frequencies of the Site 1 and Site 2 are reciprocal. Therefore, in order to meet the requirement of normal operation of the microwave 4×4 MIMO transmission, the center spacing between double-side dual-polarized antennas on the site 1 side is required to satisfy the formula (2), and in the practical deployment, on the iron towers (holding poles) of the site 1 side and the site 2 side, the requirement of MIMO station deployment needs to be fully considered, and after accurate measurement and calculation, the double-side dual-polarized antennas are correctly installed at a proper spacing. It not only has requirements on the structure and height of the iron tower (holding pole) for mounting the antenna, which increases the cost of a communication system, but also is affected by the distance measurement accuracy, the antenna installation accuracy and other factors, which greatly affects the antenna performance, causes the poor antenna reliability, and makes the antenna fail to achieve the advantages claimed by the MIMO antenna.

SUMMARY

A summary of the subject matter is described hereinafter in detail. This summary is not intended to limit the scope of the claims.
Embodiments of the present invention provide a microwave antenna array communication system and a communication method, which solve the problems of high cost, difficult installation and poor reliability caused by rigid requirements of the related microwave antenna array on the installation physical distance between dual-polarized antennas and the installation precision.
The embodiments of the present invention provide a microwave antenna array communication system. The system includes a phased array antenna array and $\frac{N}{2}$
pairs of microwave transmission devices, where N is an order of a bipolar antenna array and a value of N is greater than or equal to 4.
The phased array antenna array includes a controller and $\frac{N}{2}$
pairs of polarized antenna arrays that are in one-to-one correspondence with the $\frac{N}{2}$
pairs of microwave transmission devices.
A horizontally polarized radio frequency signal transmission device in the microwave transmission device is connected to $\frac{N}{2}$
antenna sub-arrays of a horizontally polarized antenna array in a corresponding polarized antenna array so as to send $\frac{N}{2}$
horizontally polarized radio frequency signals to an opposite end, and a vertically polarized radio frequency signal transmission device in the microwave transmission device is connected to $\frac{N}{2}$
antenna sub-arrays of a vertically polarized antenna array in the polarized antenna array so as to send $\frac{N}{2}$
vertically polarized radio frequency signals to the opposite end.
The controller is configured to configure a phase of a horizontally polarized radio frequency signal transmitted by each of the antenna sub-arrays through a phase shifter of the each of the antenna sub-arrays in the horizontally polarized antenna array, and is configured to configure a phase of a vertically polarized radio frequency signal transmitted by each of the antenna sub-arrays through a phase shifter of the each of the antenna sub-arrays in the vertically polarized antenna array.
The embodiments of the present invention provide a communication method of the microwave antenna array communication system described above. The method includes:

controlling, by the controller, a phase shifter of each of the antenna sub-arrays of the horizontally polarized antenna array to configure a phase of a horizontally polarized radio frequency signal transmitted by the each of the antenna sub-arrays, and controlling a phase shifter of each of the antenna sub-arrays of the vertically polarized antenna array to configure a phases of a vertically polarized radio frequency signal transmitted by the each of the antenna sub-arrays; and
transmitting, by a horizontally polarized radio frequency signal transmission device in the microwave transmission device, $\frac{N}{2}$
horizontal polarization radio frequency signals to an opposite end through each of the antenna sub-arrays in a corresponding horizontally polarized antenna array, and transmitting, by a vertically polarized radio frequency signal transmission device, $\frac{N}{2}$
vertically polarized radio frequency signals to the opposite end through each of the antenna sub-arrays in a corresponding vertically polarized antenna array.

The embodiments of the present invention further provide a computer-readable storage medium, which is configured to store computer-executable instructions for executing the method described above.
The beneficial effects are described below.
According to the microwave antenna array communication system and the communication method, by replacing a related double-side dipolar antenna with a phased antenna array, each horizontally polarized radio frequency signal transmission device in a microwave transmission device is directly connected to each antenna sub-array of a corresponding horizontally polarized antenna array in the phased antenna array respectively to send $\frac{N}{2}$
horizontally polarized radio frequency signals to an opposite end, and each vertically polarized radio frequency signal transmission device in the microwave transmission device is connected to each antenna sub-array of a corresponding vertically polarized antenna array respectively to send $\frac{N}{2}$
vertically polarized radio frequency signals to the opposite end; and a relationship between phases of the $\frac{N}{2}$
radio frequency signals sent by the horizontally polarized antenna array and the vertically polarized antenna array is directly configured by controlling phase shifters of each antenna sub-array of the horizontally polarized antenna array and phase shifters of each antenna sub-array of the vertically polarized antenna array through the controller of the phased antenna array without relying on a physical distance between the antenna arrays and an installation precision. Therefore, the antenna performance reliability is improved, the antenna may achieve the advantages of the MIMO antenna, and the satisfaction degree of the user on communication experience is further improved while the engineering cost and the installation difficulty are reduced.
Other aspects can be understood after the drawings and detailed description are read and understood.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is an architecture diagram of a 2×2 LoS MIMO;
FIG. 2-1 is an architecture diagram of a related double-side bipolar 4×4 MIMO;
FIG. 2-2 is a schematic diagram of related double-side bipolar 4×4 MIMO iron towers;
FIG. 3 is a schematic diagram of antenna oscillator element radiation according to an embodiment two of the present invention;
FIG. 4 is a schematic diagram of a connection of phase shifters of antenna oscillator elements according to an embodiment two of the present invention;
FIG. 5-1 is a schematic diagram of an N×N MIMO antenna array according to an embodiment two of the present invention;
FIG. 5-2 is a schematic diagram of an N×N MIMO iron tower according to an embodiment two of the present invention;
FIG. 6 is a schematic diagram of a 4×4 MIMO antenna array according to an embodiment two of the present invention;
FIG. 7 is a structural diagram of an antenna bearing plate according to an embodiment two of the present invention;
FIG. 8 is a schematic diagram of a connection of a vertically polarized antenna array according to an embodiment two of the present invention;
FIG. 9 is a schematic diagram of a vertically polarized antenna array of a local end transmitting a signal according to an embodiment two of the present invention;
FIG. 10 is a flowchart of phase control according to an embodiment two of the present invention;
FIG. 11 is a schematic diagram of 4×4 MIMO signal transmission according to an embodiment three of the present invention;
FIG. 12 is a flowchart of phase closed-loop control according to an embodiment three of the present invention;
FIG. 13 is a schematic diagram of a minimum phased antenna array according to an embodiment four of the present invention; and
FIG. 14 is a schematic diagram of a connection of minimum phased antenna arrays according to an embodiment four of the present invention.

DETAILED DESCRIPTION

The present invention will be further described in detail in conjunction with the drawings and specific embodiments.

Embodiment one

The embodiment provides a microwave bipolar antenna array communication system. It should be understood that the microwave bipolar antenna array communication system in the embodiment may be deployed at a transmit end, may be deployed at a receive end, or may be directly deployed both the transmit end and the receive end. In a Frequency Division Dual (FDD) operating mode, the transmit end and the receive end are relative. That is, when transmitting a radio frequency signal to an opposite end, the transmit end also as the receive end receives a radio frequency signal sent by the opposite end. Therefore, the embodiment is described in an example where the transmit end and the receive end are respectively replaced with the local end and the opposite end (also referred to as a remote end). The microwave bipolar antenna array communication system in the embodiment may be correspondingly deployed at the local end and the opposite end.
The microwave bipolar antenna array communication system in the embodiment includes a phased array antenna array and $\frac{N}{2}$
pairs of microwave transmission devices, where N is an order of a bipolar antenna array and a value of N is greater than or equal to 4. For example, if a 4×4 MIMO bipolar antenna array is implemented, N is 4; if an 8×8 MIMO bipolar antenna array is implemented, N is 8; and so on.
The phased array antenna array in the embodiment includes a controller and $\frac{N}{2}$
pairs of polarized antenna arrays that are in one-to-one correspondence with the $\frac{N}{2}$
pairs of microwave transmission devices. A pair of microwave transmission devices includes a horizontally polarized radio frequency signal transmission device and a vertically polarized radio frequency signal transmission device. A pair of polarized antenna arrays includes a horizontally polarized antenna array formed by $\frac{N}{2}$
antenna sub-arrays and a vertically polarized antenna array formed by $\frac{N}{2}$
antenna sub-arrays. An antenna sub-array includes at least one antenna oscillator element and a phase shifter controlling the phase of the at least one antenna oscillator element. It should be understood that each antenna oscillator element (i.e., a radiation unit) in the antenna sub-arrays in the embodiment may use a phase shifter separately, or multiple antenna oscillator elements may share one phase shifter, which may be flexibly set according to specific requirements. For example, in a setting, an antenna sub-array is formed by multiple antenna oscillator elements, and each antenna oscillator element uses a phase shifter, that is, the antenna oscillator element is in one-to-one correspondence with the phase shifter.
In the embodiment, each horizontally polarized radio frequency signal transmission device in each pair of microwave transmission devices is respectively connected to each antenna sub-array in the corresponding horizontally polarized antenna array to send $\frac{N}{2}$
horizontally polarized radio frequency signals to the opposite end, and each vertically polarized radio frequency signal transmission device in each pair of microwave transmission devices is respectively connected to each antenna sub-array in the corresponding vertically polarized antenna array to send $\frac{N}{2}$
vertically polarized radio frequency signals to the opposite end.
The controller of the phased antenna array is configured to configure a phase of a horizontally polarized radio frequency signal transmitted by each antenna sub-array of each horizontally polarized antenna array through the phase shifter of each antenna sub-array of each horizontally polarized antenna array, so as to make a phase difference of $\frac{N}{2}$
horizontally polarized radio frequency signals transmitted by one horizontally polarized antenna array meet the requirement of the N×N MIMO bipolar antenna array. Similarly, for a phase difference of $\frac{N}{2}$
vertically polarized radio frequency signals transmitted by one vertically polarized antenna array, the controller controls the phase shifter of each antenna sub-array of the vertically polarized antenna array to configure a phase of a vertically polarized radio frequency signal transmitted by each antenna sub-array, so as to make the phase difference of $\frac{N}{2}$
vertically polarized radio frequency signals transmitted by one vertically polarized antenna array meet the requirement of the N×N MIMO bipolar antenna array. The phase difference of $\frac{N}{2}$
horizontally polarized radio frequency signals transmitted by one horizontally polarized antenna array and the phase difference of $\frac{N}{2}$
vertically polarized radio frequency signals transmitted by one vertically polarized antenna array are determined according to the specific order of the N×N MIMO bipolar antenna array.The following is described with an example of a generalized monopole antenna array in conjunction with a channel matrix Vandermonde array.
A Vandermonde array corresponding to the generalized monopole antenna array N×N MIMO is as follow. $H' = H_{van} = [\begin{matrix} 1 & 1 & \dots & 1 \\ 1 & e^{- j \frac{2 π}{n}} & \dots & e^{- j \frac{2 π (n - 1)}{n}} \\ ⋮ & ⋮ & ⋱ & ⋮ \\ 1 & e^{- j \frac{2 π (n - 1)}{n}} & \dots & e^{- j \frac{2 π {(n - 1)}^{2}}{n}} \end{matrix}] .$
A Vandermonde array corresponding to the generalized 4×4 monopole antenna array (corresponding to the 8×8 MIMO dipolar antenna array) is as follow. $H_{van} = [\begin{matrix} 1 & 1 & 1 & 1 \\ 1 & e^{- j \frac{π}{n}} & e^{- j \frac{2 π}{n}} & e^{- j \frac{3 π}{n}} \\ 1 & e^{- j \frac{2 π}{n}} & e^{- j \frac{4 π}{n}} & e^{- j \frac{6 π}{n}} \\ 1 & e^{- j \frac{3 π}{n}} & e^{- j \frac{6 π}{n}} & e^{- j \frac{8 π}{n}} \end{matrix}] = [\begin{matrix} 1 & 1 & 1 & 1 \\ 1 & - j & - 1 & j \\ 1 & - 1 & 1 & - 1 \\ 1 & j & - 1 & - j \end{matrix}] .$
The following is obtained after two times of unitary transformation. $H_{opt} = [\begin{matrix} e^{j \frac{3 π}{4}} & 0 & 0 & 0 \\ 0 & e^{j \frac{2 π}{4}} & 0 & 0 \\ 0 & 0 & e^{j \frac{π}{4}} & 0 \\ 0 & 0 & 0 & e^{j 0} \end{matrix}] H_{van} [\begin{matrix} e^{j} \end{matrix}] [\begin{matrix} 0 & 0 & 0 & 0 \\ 0 & e^{- j \frac{π}{4}} & 0 & 0 \\ 0 & 0 & e^{- j \frac{2 π}{4}} & 0 \\ 0 & 0 & 0 & e^{- j \frac{3 π}{4}} \end{matrix}] = [\begin{matrix} e^{j \frac{3 π}{4}} & e^{j \frac{2 π}{4}} & e^{j \frac{π}{4}} & e^{j 0} \\ e^{j \frac{2 π}{4}} & e^{- j \frac{π}{4}} & e^{- j \frac{4 π}{4}} & e^{- j \frac{7 π}{4}} \\ e^{j \frac{π}{4}} & e^{j \frac{4 π}{4}} & e^{j \frac{7 π}{4}} & e^{j \frac{10 π}{4}} \\ e^{j 0} & e^{j \frac{π}{4}} & e^{j \frac{2 π}{4}} & e^{j \frac{3 π}{4}} \end{matrix}]$
Using the first row as an example, assuming that the local end and the opposite end corresponds to the first row are a Tx 1 and an Rx 1, four paths of Txs are required to reach the receive end antenna at intervals of $\frac{π}{4}$
(i.e., the phase difference) with respect to the Rx 1. Other high-order MIMO may be constructed according to this method. For example, for the generalized monopole antenna array N×N MIMO, the phase difference is required to be $\frac{π}{N},$
and for the narrow-sense dipolar antenna array N×N MIMO converted from the generalized monopole antenna array N×N MIMO, the phase difference is required to be $\frac{2 π}{N} .$
Therefore, in the embodiment, for each horizontally polarized antenna array, the controller configures a phase difference between each of horizontally polarized radio frequency signals transmitted by the antenna sub-arrays and each of horizontally polarized radio frequency signals transmitted by adjacent antenna sub-arrays in the horizontally polarized antenna array to be $\frac{2 π}{N}$
through the phase shifter of the each of the antenna sub-arrays in the horizontally polarized antenna array, the controller configures a phase difference between each of vertically polarized radio frequency signals transmitted by the antenna sub-arrays and each of horizontally polarized radio frequency signals transmitted by adjacent antenna sub-arrays in the vertically polarized antenna array to be $\frac{2 π}{N}$
through the phase shifter of the each of the antenna sub-arrays in the vertically polarized antennas array.
In the embodiment, for $\frac{N}{2}$
pairs of polarized antenna arrays of the local end or the opposite end, the $\frac{N}{2}$
pairs of polarized antenna arrays may be disposed on an antenna bearing plate, which may simplify the installation procedure and improve the installation efficiency. Each of the $\frac{N}{2}$
pairs of polarized antenna arrays may be disposed on a respective antenna bearing plate according to the actual requirements, which may improve the flexibility of the antenna installation and application and meet more application scenarios.
In the embodiment, the controller may control the phase of the radio frequency signal transmitted by each antenna sub-array of each horizontally polarized antenna array or each vertically polarized antenna array in an open-loop control manner. That is, the controller may be configured according to the above process. The configured phase difference is finally used for facilitating the demodulation of the baseband digital signal in a modem. The system gain maximization may not be achieved merely considering the requirement of the $\frac{2 π}{N}$
phase difference between the local end and the remote end antennas, because, besides the phase difference between the antennas, the phase difference may be caused by a waveguide connector, a radio frequency cable and the like used between the microwave device radio frequency unit and the phased array antenna array. The radio frequency transceiving channels are separate from each other, which ensures the maximum gain of the MIMO demodulation. According to the above process, the phase of $\frac{2 π}{N}$
is first coarsely adjusted. That is, it is ensured that the corresponding horizontally polarized antenna array or vertically polarized antenna array between the local end and remote end antennas meets the requirement of the phase of $\frac{2 π}{N} .$
Then, it is expected that the system will operate in an MIMO mode. At this moment, after the controller configures the phase of the horizontally polarized radio frequency signal transmitted by each antenna sub-array of each horizontally polarized antenna array according to the above requirement, the controller is further configured to acquire a difference between the $\frac{2 π}{N}$
and a receive phase angle of a horizontally polarized antenna array corresponding to the opposite end for receiving the horizontally polarized radio frequency signal transmitted by each antenna sub-array of the horizontally polarized antenna array of the local end, and in response to determining that the difference between the $\frac{2 π}{N}$
and the receive phase angle is greater than a preset horizontally polarized phase angle offset threshold, adjust the phase of the horizontally polarized radio frequency signal transmitted by each antenna sub-array of the horizontally polarized antenna array according to the difference between the $\frac{2 π}{N}$
and the receive phase angle (which is a fine adjustment process on the phase difference) until the difference between the $\frac{2 π}{N}$
and the receive phase angle is less than or equal to the preset horizontally polarized phase angle offset threshold.
Similarly, after the controller configures the phase of the vertically polarized radio frequency signal transmitted by each antenna sub-array of each vertically polarized antenna array according to the above requirement, the controller is further configured to acquire a difference between the $\frac{2 π}{N}$
and a receive phase angle of a vertically polarized antenna array corresponding to the opposite end for receiving the vertically polarized radio frequency signal transmitted by each antenna sub-array of the vertically polarized antenna array of the local end, and in response to determining that the difference between the $\frac{2 π}{N}$
and the receive phase angle is greater than a preset vertically polarized phase angle offset threshold, adjust the phase of the vertically polarized radio frequency signal transmitted by each antenna sub-array of the horizontally polarized antenna array according to the difference between the $\frac{2 π}{N}$
and the receive phase angle until the difference between the $\frac{2 π}{N}$
and the receive phase angle is less than or equal to the preset vertically polarized phase angle offset threshold.
In the embodiment, the difference between the $\frac{2 π}{N}$
and the receive phase angle may be calculated at the local end, or may be calculated at the opposite end, and the specific calculation manner may be any manner that may obtain the receive phase angle according to a performance index of the received signal or directly obtain the difference (i.e., a receive phase angle error) between the $\frac{2 π}{N}$
and the receive phase angle, which is not repeated herein.
The phase shifter in the embodiment may be a discrete digital phase shifter or may be a non-discrete analog phase shifter. In the above fine adjustment process, the controller may use a stepping adjustment manner. The adjustment is performed again after the opposite end updates its receiver MIMO performance index again, and the adjustment stops when the performance index fed back by the remote end reaches a certain threshold range. At this moment, it is considered that the closed-loop phase adjustment process of the MIMO system is completed. Since the transceiving channel is reciprocal, after the local end is adjusted, the link from the opposite end to the local end is adjusted by default, and the MIMO system enters a long-term stable operating state.
In the embodiment, for each horizontally polarized antenna array or each vertically polarized antenna array, the controller may also adjust its transmit power before or after configuring its phase. The adjustment process is described below.
For each horizontally polarized antenna array, transmit power of the horizontally polarized antenna array, receive power of a horizontally polarized antenna array corresponding to the opposite end and a difference of a path insertion loss to the opposite end are acquired as a horizontally polarized power difference. In condition that the acquired horizontally polarized power difference is greater than or equal to a preset horizontally polarized power difference threshold, a main lobe radiation angle of the horizontally polarized antenna array is adjusted until the horizontally polarized power difference is less than or equal to the preset horizontally polarized power difference threshold. Of course, the transmit power of the horizontally polarized antenna array of the local end may be directly adjusted to achieve the above effect. The transmit power of the horizontally polarized antenna array may also be adjusted by combining these two manners or from other aspects as long as the above effect is achieved.
For each vertically polarized antenna array, transmit power of the vertically polarized antenna array, receive power of a vertically polarized antenna array corresponding to the opposite end and a difference of a path insertion loss to the opposite end are acquired as a vertically polarized power difference. When the acquired vertically polarized power difference is greater than or equal to a preset vertically polarized power difference threshold, a main lobe radiation angle of the vertically polarized antenna array is adjusted until the vertically polarized power difference is less than or equal to the preset vertically polarized power difference threshold. Of course, the transmit power of the vertically polarized antenna array of the local end may be directly adjusted to achieve the above effect. The transmit power of the vertically polarized antenna array may also be adjusted by combining these two manners or from other aspects as long as the above effect is achieved.
However, it should be understood that the above power adjustment process may be directly skipped in a case where the initial power is set well, or the power may be adjusted in real time in the subsequent operating process. In addition, the specific value of the various thresholds in the embodiment may be flexibly chosen according to specific communication environment requirement.
The embodiment, directly connects each horizontally polarized radio frequency signal transmission device in the microwave transmission device to each sub-array of a corresponding horizontally polarized antenna array in the phased antenna array respectively to transmit a horizontally polarized radio frequency signal to the opposite end and connects each vertically polarized radio frequency signal transmission device to each sub-array of a corresponding vertically polarized antenna array to transmit a vertically polarized radio frequency signal to the opposite end by replacing a related double-side bipolar antenna with a phased antenna array, configures a relationship between phases of radio frequency signals transmitted by the horizontally polarized antenna array and vertically polarized antenna array by directly controlling the phase shifter of each sub-array of the horizontally polarized antenna array and the phase shifter of each sub-array of the vertically polarized antenna array through the controller of the phased antenna array, and sends corresponding signals to the opposite end through each antenna sub-array of the corresponding horizontally polarized antenna array and each antenna sub-array of the corresponding vertically polarized antenna array. The reliability of the antenna performance is improved and the antenna may exert the advantages of the MIMO antenna while the engineering cost and the installation difficulty are reduced.

Embodiment two

To better understand the present disclosure, the present embodiment will be described below in conjunction with the specific implementation of the phased antenna array.
The horizontally polarized antenna array and the vertically polarized antenna array of the phased array antenna array are generally defined as being formed by a group of independent antenna oscillator elements, and relative amplitude and phase relationships may be ensured through related circuit design, so that a target of focusing and forming in a certain expected direction is achieved, and compared with other directions, the energy radiated by electric waves is greatly reduced (suppressed). Any antenna oscillator element is separately controllable and uniformly distributed on a straight line. For example, as shown in FIG. 3, a row of six antenna oscillator elements are distributed on a straight line, the antenna oscillator elements sequentially radiate from right to left, and finally, an electric wave with its wave front having a phase angle may be formed. That is, the main lobe radiation angle may be adjusted through programmed radiation delay. Therefore, for a phased array antenna, the phased antenna array has a capability of adjusting the main lobe radiation direction.
An implementation manner of an antenna sub-array of the horizontally polarized antenna array and the vertically polarized antenna array is shown in FIG. 4.
In FIG. 4, the circle plus the arrow denotes the phase shifter, and all antenna oscillator elements are non-directional and fed in the same phase with equal amplitude. If the phase difference of excitation currents of the adjacent antenna oscillator elements is ϕ, the corresponding radiation direction angle θ is calculated according to a formula described below. $θ = \sin^{- 1} \frac{φ}{d .2 π / λ}$
A sum of field strength vectors of all antenna oscillator elements at a certain point of a radiation field in a far region in the θ direction is calculated according to a formula described below. $E (θ) = E_{0} + E_{1} + \dots + E_{i} + \dots + E_{N - 1}$
Assuming that each antenna oscillator element is fed with the equal amplitude, the radiation field strength of each antenna oscillator element at this point is characterized as the following formula (using the antenna oscillator element No. 0 in FIG. 4 as a phase reference). $E (θ) = E \sum_{k = 0}^{N - 1} e^{jk (ψ - φ)} = E \sin [N] \frac{[\frac{}{2} (ψ - φ)]}{\sin} e^{j [\frac{N - 1}{2} (ψ - φ)]}$
When ψ=ϕ (ψ denotes an observation angle with respect to the antenna array), components with equal phase are added, and the field strength radiation is maximized (indicating that the main lobe is maximized in this aspect, namely, the effect of electrically adjusting the main lobe direction is achieved), which is described below. $|E {(θ)}_{\max}|$
When ϕ changes, it is known according to the antenna transceiving reciprocity theorem that the receive antenna also meet the corresponding conclusion. The above is popularized to a two-dimensional planar array, and the electrically main lobe electric control scanning in three dimensions of space may be completed by adjusting phase shift values of all feed sources arriving the planar array.
The embodiment is described in an example in which the local end is the site 1 and the opposite end is the site 2, as shown in FIG. 5-1. The microwave bipolar antenna array communication systems are correspondingly set at the local end and the opposite end in FIG. 5. In the figure, V0 and H0 form a pair of microwave transmission device, where V0 is a horizontally polarized radio frequency signal transmission device and H0 is a vertically polarized radio frequency signal transmission device. In the figure, a total of N pairs of microwave transmission devices, V0+H0, ..., and VN+HN, are at each end. Correspondingly, N pairs of polarized antenna arrays are disposed on an antenna bearing plate 1 at each end. Each pair of polarized antenna arrays is composed of a horizontally polarized antenna array 21 and a vertically polarized antenna array 20. In FIG. 5-1, a bipolar antenna array 2N×2N MIMO is implemented. Corresponding to the 2N×2N MIMO shown in FIG. 5-1, the iron tower installation schematic diagram is shown in FIG. 5-2. During installation, the physical distances between antennas are not precisely measured and the antennas are not installed, which is different from the related double-side polarized antenna. The phase difference of the antennas is mainly achieved through the control of phase shifter. Therefore, the practicability and reliability of the MIMO antenna system may be improved.
In FIG. 5-1, the horizontally polarized antenna array receives and transmits the corresponding horizontally polarized radio frequency signal, and the vertically polarized antenna array receives and transmits the corresponding vertically polarized radio frequency signal. The vertical and horizontal relationships in the figure are relative to the ground plane. The corresponding geometric combination relationship of array units needs to be designed according to the operating frequency band, the antenna gain and the like in the practical implementation of the phased array, which is not necessarily the topology shown in FIG. 5-1 that is only an example to facilitate the description. After the microwave transmission devices are respectively connected to the corresponding vertically or horizontally polarized antenna array as shown in FIG. 5-1, the antenna bearing plate 1 is mounted on the iron tower (the holding pole) through a bracket or a structural member. The phase shifter and the controller are integrated in the antenna bearing plate 1. The corresponding radiation beam phase adjustment and beam forming may be completed through the corresponding algorithm or software, so as to meet the requirements of LoS MIMO on the transmission channel matrix, thereby improving in multiples the transmission capacity and the performance.
On the basis of FIG. 5-1, the following will be described in an example where the bipolar antenna array 4×4 MIMO is implemented. Referring to FIG. 6, two pairs of polarized antenna arrays are disposed on the antenna bearing plate 1 at the two ends of the site 1 and the site 2, and two pairs of microwave transmission devices, V0+H0 and V1+H1, are disposed at the two ends. The connection of each pair of microwave transmission devices and each polarized antenna array is shown in FIG. 6. The V0 and V1 are connected to the corresponding vertically polarized antenna array 20, and the H0 and H1 are connected to the corresponding horizontally polarized antenna array 21. In FIG. 6, each horizontally polarized antenna array 21 and each vertically polarized antenna array 21 include two antenna sub-arrays.
The specific structure of the antenna bearing plate 1 is shown in FIG. 7. In FIG. 7, the black micro rectangular module represents the antenna oscillator element. The antenna oscillator element may be various types of elements, such as an antenna radiation element surface-mounted with a low-cost flame resistance rating (FR) 4 printed circuit board (PCB). The two pairs of horizontally polarized antenna arrays 21 and vertically polarized antenna arrays 20 are respectively connected to the corresponding V0, H0, V1 and H1. Both the horizontally polarized antenna array 21 and the vertically polarized antenna array 20 include two antenna sub-arrays. Each antenna oscillator element of each antenna sub-array corresponds to a phase shifter (not shown), and each phase shifter is connected to the controller. The controller completes the self-adaptive processing on the radiation main lobe and power for the four paths of signals. Specifically, the phase and the gain of each radiation element in any one antenna array in four paths of signals need to be correspondingly set. For the conventional double-side polarized antenna array, a corresponding space distance between dipolar antennas needs to be calculated according to the operating frequency of the device and the spacing between one-hop microwave links through a corresponding theoretical formula, and then the dual-polarized parabolic antennas are installed on the iron tower (the holding pole) at this space distance. Different from the conventional double-side polarized antenna array, in the embodiment, the horizontally polarized antenna array 21 and the vertically polarized antenna array 20 are both fixed on the antenna bearing plate 1, the physical form of the antenna arrays is fixed, and the radio frequency signal spacing relationship in the same polarization direction is fixed. Therefore, the problem of the remote antenna feeder does not need to be particularly considered when the phased antenna array is used. The MIMO transmission channel may be constructed through the electrically-adjusted phased array for different frequencies and communication distances. In addition, for the baseband, since specific calculation of antenna layout is avoided, for the application of integrated whole-city external device and the like, an overlong MIMO mutual transmission channel between devices is avoided, thereby greatly reducing the device complexity and reducing the product cost (EMC, lightning protection and the like). The MIMO device may get rid of the conventional installation with high difficulty and high precision, so that the MIMO device may be rapidly deployed. The related one-hop communication distance and frequency points are configured to automatically perform the related phase shift and MIMO transmission channel implementation through the controller after entering the device.
An example of the specific implementation of each horizontally polarized antenna array 21 and vertically polarized antenna array 20 is shown in FIG. 8. FIG. 8 exemplarily illustrates the implementation of the vertically polarized antenna array 20. The antenna oscillator elements in the two antenna sub-arrays 201 and 202 and the connection of the antenna oscillator elements and phase shifters (PS) to power dividers are shown in FIG. 8. The controller determines a corresponding phase shift value. A pre-stage power adjustment module completes the power control of each path of beam forming. In order to implement the phase shift of $\frac{2 π}{4} = 90 °,$
the local end RF Tx Lo provides an antenna local oscillator to a lower half array through the phase shift by 90° after the power division. The implementation of the horizontally polarized antenna array 21 is similar to the method shown in FIG. 8.
Assuming that the 4×4 MIMO shown in FIG. 6 has an RF operating frequency band of 15G and a one-hop communication distance of 5 Km, using the local site 1, as an example, referring to FIG. 9, the inside of any one vertically polarized antenna array 21 is divided into two antenna sub-arrays with the same polarization, which are respectively the antenna sub-array in a radiation lobe 021, which corresponds to the receive array V0 path of the opposite site 2, and the antenna sub-array in a radiation lobe 121, which corresponds to the receiving array site V1 path of the opposite site 2. The V0 path and V1 path at the opposite end are two groups of independent vertically polarized antenna array unit groups, which are distributed and designed according to fixed positions in the integrated antenna, so that corresponding main lobe focusing and alignment may be implemented by performing beam forming control on two antenna sub-arrays in the V0 path of the local end. The most important 90° phase required by the maximum MIMO channel transmission capacity may also be set through the automatic electric regulation in the phased antenna array. The example here requires that the radiation lobe 021 is 90° ahead of the radiation lobe 121, so as to meet the antenna spacing requirement in the conventional dual-polarized MIMO (the remaining H0, H1 and V1 at the local end also have two antenna sub-arrays, whose physical requirements on the lobe of the corresponding array radiated to the opposite site 2 are consistent with the behavior relationship of VO). That is, the LoS MIMO operating necessary condition in the conventional solution, an 90° electric wave transmission path phase difference implemented through spatial arrangement, is implemented through the phase shifter in the phased antenna array, and this phase shift relationship may be adjusted and finely adjusted in real time according to the requirements of users, so that the engineering installation of the LoS MIMO antenna of the microwave device becomes a job as simple as the conventional single-polarized 1+0 single-polarization microwave. For the control of the power and the phase of each horizontally polarized antenna array 21 and vertically polarized antenna array 20, reference is made to steps shown in FIG. 10.
In step S1001, assuming that the one-hop spacing D and the frequency point F between the local end and the opposite end have been determined, the transmit power Ptx (x may be V0, H0, V1 and HI) corresponding to each polarized antenna array (two horizontally polarized antenna arrays 21 and two vertically polarized antenna arrays 20 corresponding to V0, H0, V1 and H1) and the path insertion loss Ld from the local end to the opposite end are acquired, and the corresponding power difference threshold is set (each horizontally polarized antenna array 21 and each vertically polarized antenna array 20 may be provided with a preset horizontally polarized power difference threshold and a preset vertically polarized power difference threshold respectively, and of course may use the same power difference threshold).
In step S1002, the receive power Prx (x may be V0, H0, V1 and H1) corresponding to each polarized antenna array (two horizontally polarized antenna arrays 21 and two vertically polarized antenna arrays 20 corresponding to V0, H0, V1 and H1,) is acquired after the opposite end antenna is manually aligned. This step may be performed simultaneously with step S1001. In step S1003, it is decided whether Ptx-Ld-Prx is less than or equal to the power difference threshold; if yes, go to step S1005; if no, go to step S1004.
In step S1004, a main lobe radiation angle of the polarized antenna array corresponding to the Ptx is adjusted until Ptx-Ld-Prx is less than or equal to the power difference threshold; and go to step S1005.
In step S1005, after the current Ptx main lobe adjustment is completed, go to step S1003 to traverse the next polarized antenna array until all polarized antenna arrays are traversed.
In step S1006, the phase shifter in each polarized antenna array is adjusted to ensure the 90° phase shift between receive antenna sub-arrays of the opposite end.
At this point, the one-hop 4×4 LoS MIMO completes the phased array antenna configuration at the local end, and the corresponding phased array antenna configuration at the opposite end is also completed, so as to ensure that each path of transmitted signals of the local end meets the requirement of the 90° phase difference for constructing the maximum transmission channel after reaching the opposite end. The baseband MIMO processing function is started, and a receiver system of the opposite end completes the baseband operation and processing of capturing, synchronizing and locking and then completes the normal receiving and demodulation of each path of data, thereby doubling the transmission capacity. The processing mechanisms from the opposite end to the local end are consistent with the mechanism described above, which will not be repeated herein.

Embodiment three

Besides the open-loop control on the phase shifter in each polarized antenna array shown in the above embodiments, the embodiment provides a closed-loop precise control process. The closed-loop control is particularly applicable to for different one-hop communication distances and device operating frequency bands.
The embodiment uses the bipolar antenna array 4×4 MIMO as an example. In the MIMO demodulation process, in order to demodulate any path of signal, the baseband is equivalent to a receiver structure with a path of main path signal and three paths of slave path signals. For example, using the V0 path reception of Rx0 shown in FIG. 11 as an illustration object, after H0, V1 and H1 paths are filtered in the main received signal, the data of the V0 path may be recovered and then be correctly demodulated.
FIG. 11 is a 4×4 MIMO, and if H0 and H1 in FIG. 11 are removed first, it becomes a case of the 2×2 MIMO in the single polarization. The first path of receive signal is R0=V0+V1^∗e^j(θ0), and the second path of receive signal is R1=V1+V0^∗e^j(θ1). Using the first path of receive signal as an example, the MIMO demodulation is to estimate the θ0, then to send this angle θ0 to the transmit side by a closed-loop control channel, and to dynamically adjust by a transmitter the phase to adjust the θ0 to about 90°. The similar processing is performed on the θ1. Finally, the Rx1 will send its own reception to Rx0, the final demodulated signal in the first path is: R0-e^j(θ0)^∗(V1+V0^∗e^j(θ1))=V0-V0^∗e^j(θ0+θ1), and ideally θ0 = θ1 = 90°, so 2R0 is finally demodulated.
The θ angle herein is the phase angle difference, and ideally corresponds to 90° required in the case of 4×4 MIMO described above. Since the phase difference is finally used for facilitating the demodulation of the baseband digital signal in a modem, the system gain maximization may not be achieved only considering the requirement of the 90° phase difference between the local end and the remote end antennas, because besides the phase difference between the antennas, the phase difference may be caused by the waveguide connector, the radio frequency cable and the like used between the microwave device radio frequency unit and the phased array antenna array. The radio frequency transceiving channels are separate from each other, which ensures the maximum gain of the MIMO demodulation. The phase difference may also be adaptively adjusted through a feedback loop. According to the process shown in the embodiment two, the coarse adjustment of the 90° phase is first completed according to the configuration in the device user interface. That is, it is ensured that the corresponding array unit group between the local end and the opposite end antennas meets the 90° phase requirement. It is then contemplated that the system will operate in the MIMO mode. Because the system index is not optimal, the closed-loop phase fine adjustment process is started, and the link establishment of a closed-loop control channel is performed in a one-hop interval according to a lower modulation mode (such as the quadrature phase shift keying (QPSK), 16 quadrature amplitude modulation (QAM) and the like requiring a lower signal-noise ratio (SNR)). After the link is established, the error between the phase angle received by the opposite end and the ideal angle may be estimated (the specific algorithm may be any related error estimation algorithm, and the details are not repeated herein). The observable indicators include mean square error (MSE) and forward error correction (FEC) decoding conditions. The error is sent to the local end through the established closed-loop control channel, and the actual phase condition of the receiver at the opposite end is calculated according to the error distribution condition after the error is received by the local end. After the actual phase condition is compared with the ideal 90° phase relationship, the phase shift angle of the corresponding array unit group is adjusted by notifying the phased array antenna control module of the local end by transmitting a specific phase regulation instruction. Since the circuit structure shown in FIG. 8 is used, all phase adjustments are electrically adjustable, and the phase shift relationship may correspond to a specific circuit. In the adjustment process, a stepping mode may be used. After the opposite end updates the MIMO performance index of the receiver again, the adjustment is performed again. If the performance index fed back by the opposite end reaches a certain threshold range, the adjustment is stopped, and it is considered that the closed-loop phase adjustment process of the MIMO system is completed. Since the transceiving channel is reciprocal, after the local end is adjusted, the link from the opposite end to the local end is adjusted by default, and the LoS MIMO system enters a long-term stable operating state.
For the above closed-loop control process, reference is made to steps shown in FIG. 12.
In step S1201, assuming that the one-hop spacing D and the frequency point F between the local end and the opposite end have been determined, the transmit power Ptx (x may be V0, H0, V1 and HI) corresponding to each polarized antenna array (two horizontally polarized antenna arrays 21 and two vertically polarized antenna arrays 20 corresponding to V0, H0, V1 and HI) and the path insertion loss Ld from the local end to the opposite end are acquired, and the corresponding power difference threshold is set (each horizontally polarized antenna array 21 and each vertically polarized antenna array 20 may be provided with a preset horizontally polarized power difference threshold and a preset vertically polarized power difference threshold respectively, and of course may use the same power difference threshold).
In step S1202, the receive power Prx (x may be V0, H0, V1 and H1) corresponding to each polarized antenna array (two horizontally polarized antenna arrays 21 and two vertically polarized antenna arrays 20 corresponding to V0, H0, V1 and H1,) is acquired after the opposite end antenna is manually aligned.
In step S1203, it is decided whether Ptx-Ld-Prx is less than or equal to the power difference threshold; if yes, go to step S1205; if no, go to step S1204.
In step S1204, a main lobe radiation angle of the polarized antenna array corresponding to the Ptx is adjusted until Ptx-Ld-Prx is less than or equal to the power difference threshold; and go to step S1205.
In step S1205, after the current Ptx main lobe adjustment is completed, go to step S1203 to traverse the next polarized antenna array until all polarized antenna arrays are traversed.
In step S1206, the phase shifter in each polarized antenna array is adjusted to ensure the 90° phase shift between receive antenna sub-arrays of the opposite end.
In step S1207, the local end and the opposite end force the modulation mode to a preset modulation mode (such as the QPSK), and enable the closed-loop control channel.
In step S1208, the Prx performs 90° phase shift fine adjustment on each polarized antenna array until the phase angle estimation error of each polarized antenna array is less than or equal to a set threshold, and the MSE reaches the MIMO threshold.
In step S1209, the local end and the opposite end change the modulation mode back to the original user configuration mode, and enter the stable operation.
The phase difference self-adaptive adjustment of the feedback loop provided by the embodiment may further improve the antenna performance, and ensure the MIMO antenna array reliability. After the phase is configured based on the above process, that is, signals may be sent to the opposite end, so that the engineering cost and the installation difficulty may be reduced, the antenna performance reliability may be improved, and the antenna may exert the advantages of the MIMO antenna.
It should be understood by those skilled in the art that each of the modules or steps in the embodiments of the present invention described above may be implemented by a general-purpose computing apparatus, the modules or steps may be concentrated on a single computing apparatus or distributed on a network composed of multiple computing apparatuses, and alternatively, the modules or steps may be implemented by program codes executable by the computing apparatuses, so that the modules or steps may be stored in a computer storage medium (such as a ROM/RAM, a magnetic disk or an optical disk) and executed by the computing apparatuses. In some circumstances, the illustrated or described steps may be executed in sequences different from those described herein, or the modules or steps may be made into various integrated circuit modules separately, or multiple modules or steps therein may be made into a single integrated circuit module for implementation.

Embodiment four

As analyzed in Embodiment one, in this embodiment, the horizontally polarized antenna array and the vertically polarized antenna array of a pair of polarized antenna arrays may be disposed on an antenna bearing plate. For example, for user applications, it may not be required that all scenarios are MIMO systems, i.e. physically integrated phased array antenna arrays, and correspondingly, the problem of multiplexing flexibility will not exist in XPIC and protection scenarios. The embodiment proposes another implementation. That is, the original N×N antenna array is decomposed into physically independent minimum units according to the requirements of XPIC group which includes a pair of polarized antenna arrays (i.e. a horizontally polarized antenna array and a vertically polarized antenna array). As shown in FIG. 13, a horizontally polarized antenna array and a vertically polarized antenna array are included. In an application scenario, the operating frequency band of 15G and the one-hop communication distance of 5 km are still used as an example (see FIG. 6). The distance between the dual-polarized antennas is theoretically required to be 7.07 m. Considering the requirements of the installation on the iron tower (the holding pole), the double-side minimum phased antenna array is installed at a distance of 1 m, and since it is not installed at the ideal 7-m ideal distance, the 90° phase adjustment and device described in Embodiment two needs to be first performed on the double-side phased array antenna. As shown in FIG. 14, since the antenna distance between the double-side antennas is determined at the installation and engineering implementation stage and is not a fixed value, the phase difference finally appearing at the receive end may be a random angle within a certain range around 90°. At this moment, the closed-loop phase adjustment described in Embodiment three is performed, and the phase between the local end and the radiation array unit corresponding to the opposite end is automatically adjusted and finely adjusted. After the final adjustment is completed, the random antenna distance (determined by the user according to specific installation conditions) has no influence on the dipolar antenna MIMO system in the embodiment, and the system will be automatically adjusted and converge to the optimal operating state. That is, it is ensured that the transmission channel matrix meets the requirements of the Vandermonde matrix, and that each radiation array units are constructed in the optimal phase difference relationship, thereby maximizing the MIMO system gain and optimizing the system transmission capacity and the system gain. At the same time, the minimum phased antenna array shown in FIG. 13 may also flexibly construct 2+0, 2+2, 1+0 and other microwave applications when the MIMO application is not performed, which makes its size more compact and the weight lighter, thereby further optimizing the engineering installation.
The above content is a further detailed description of the present invention in conjunction with the specific preferred embodiments, and the specific implementation of the present invention is not limited to the description.

INDUSTRIAL APPLICABILITY

According to the microwave antenna array communication system and the communication method provided in the embodiments of the present invention, by replacing a related double-side dipolar antenna with a phased antenna array, each horizontally polarized radio frequency signal transmission device in a microwave transmission device is directly connected to each antenna sub-array of a corresponding horizontally polarized antenna array in the phased antenna array respectively to send $\frac{N}{2}$
horizontally polarized radio frequency signals to an opposite end, and each vertically polarized radio frequency signal transmission device is connected to each antenna sub-array of a corresponding vertically polarized antenna array respectively to send $\frac{N}{2}$
vertically polarized radio frequency signals to the opposite end; and a relationship between phases of the $\frac{N}{2}$
radio frequency signals sent by the horizontally polarized antenna array and the vertically polarized antenna array is directly configured by controlling the phase shifter of each antenna sub-array of the horizontally polarized antenna array and the phase shifter of each antenna sub-array of the vertically polarized antenna array through the controller of the phased antenna array without relying on a physical distance between the antenna arrays and an installation precision. Therefore, the antenna performance reliability is improved, the antenna may achieve the advantages of the MIMO antenna, and the satisfaction degree of the user on communication experience may be further improved while the engineering cost and the installation difficulty are reduced.

Claims

A microwave antenna array communication system, comprising a phased array antenna array and $\frac{N}{2}$
pairs of microwave transmission devices, wherein N is an order of a bipolar antenna array and a value of N is greater than or equal to 4; wherein
the phased array antenna array comprises a controller and $\frac{N}{2}$
pairs of polarized antenna arrays that are in one-to-one correspondence with the $\frac{N}{2}$
pairs of microwave transmission devices;
wherein a horizontally polarized radio frequency signal transmission device in the microwave transmission device is connected to $\frac{N}{2}$
antenna sub-arrays of a horizontally polarized antenna array in a corresponding polarized antenna array so as to transmit $\frac{N}{2}$
horizontally polarized radio frequency signals to an opposite end, and a vertically polarized radio frequency signal transmission device is connected to $\frac{N}{2}$
antenna sub-arrays of a vertically polarized antenna array in the polarized antenna array so as to transmit $\frac{N}{2}$
vertically polarized radio frequency signals to the opposite end; and
wherein the controller is configured to configure a phase of a horizontally polarized radio frequency signal transmitted by each of the antenna sub-arrays through a phase shifter of the each of the antenna sub-arrays in the horizontally polarized antenna array, and is configured to configure a phase of a vertically polarized radio frequency signal transmitted by each of the antenna sub-arrays through a phase shifter of the each of the antenna sub-arrays in the vertically polarized antenna array.
The microwave antenna array communication system of claim 1, wherein the controller is configured to configure a phase difference between each of horizontally polarized radio frequency signals transmitted by the antenna sub-arrays and each of horizontally polarized radio frequency signals transmitted by adjacent antenna sub-arrays in the horizontally polarized antenna array to be $\frac{2 π}{N}$
through the phase shifter of the each of the antenna sub-arrays in the horizontally polarized antenna array; and is configured to configure a phase difference between each of vertically polarized radio frequency signals transmitted by the antenna sub-arrays and each of horizontally polarized radio frequency signals transmitted by adjacent antenna sub-arrays in the vertically polarized antenna array to be $\frac{2 π}{N}$
through the phase shifter of the each of the antenna sub-arrays in the vertically polarized antennas array.
The microwave antenna array communication system of claim 1, wherein the $\frac{N}{2}$
pairs of polarized antenna arrays are located on an antenna bearing plate;
or,
each of the $\frac{N}{2}$
pairs of polarized antenna arrays is located on a respective antenna bearing plate.
The microwave antenna array communication system of claim 2, wherein the controller is further configured to, after the phase of the horizontally polarized radio frequency signal transmitted by the each of the antenna sub-arrays in the horizontally polarized antenna array is configured, acquire a difference between $\frac{2 π}{N}$
and a receive phase angle of a horizontally polarized antenna array corresponding to the opposite end for receiving the horizontally polarized radio frequency signal transmitted by the each of the antenna sub-arrays of the horizontally polarized antenna array, and in response to determining that the difference between $\frac{2 π}{N}$
and the receive phase angle of the horizontally polarized radio frequency signal is greater than a preset horizontally polarized phase angle offset threshold, adjust the phase of the horizontally polarized radio frequency signal transmitted by the each of the antenna sub-arrays in the horizontally polarized antenna array according to the difference between $\frac{2 π}{N}$
and the receive phase angle of the horizontally polarized radio frequency signal until the difference between $\frac{2 π}{N}$
and the receive phase angle of the horizontally polarized radio frequency signal is less than or equal to the preset horizontally polarized phase angle offset threshold; and
the controller is further configured to, after the phase of the vertically polarized radio frequency signal transmitted by the each of the antenna sub-arrays in the vertically polarized antenna array is configured, acquire a difference between $\frac{2 π}{N}$
and a receive phase angle of a vertically polarized antenna array corresponding to the opposite end for receiving the vertically polarized radio frequency signal transmitted by the each of the antenna sub-arrays of the vertically polarized antenna array, and in response to determining that the difference between $\frac{2 π}{N}$
and the receive phase angle of the vertically polarized radio frequency signal is greater than a preset vertically polarized phase angle offset threshold, adjust the phase of the vertically polarized radio frequency signal transmitted by the each of the antenna sub-arrays in the vertically polarized antenna array according to the difference between $\frac{2 π}{N}$
and the receive phase angle of the vertically polarized radio frequency signal until the difference between $\frac{2 π}{N}$
and the receive phase angle of the vertically polarized radio frequency signal is less than or equal to the preset vertically polarized phase angle offset threshold.
The microwave antenna array communication system of any one of claims 1 to 4, wherein the controller is further configured to acquire transmit power P_ht of the horizontally polarized antenna array, receive power P_hr of the horizontally polarized antenna array corresponding to the opposite end and a path insertion loss L_hd to the opposite end, calculate a horizontally polarized power difference value ΔP_h according to a formula ΔP_h = P_ht -P_hr-L_hd , and in response to determining that the horizontally polarized power difference value is greater than or equal to a preset horizontally polarized power difference threshold, adjust a main lobe radiation angle of the horizontally polarized antenna array until the horizontally polarized power difference value is less than the preset horizontally polarized power difference threshold; and
the controller is further configured to acquire transmit power P_vt of the vertically polarized antenna array, receive power P_vr of the vertically polarized antenna array corresponding to the opposite end and a path insertion loss L_vd to the opposite end, calculate a vertically polarized power difference value ΔP_v according to a formula ΔP_v = P_vt -P_vr -L_vd, and in response to determining that the vertically polarized power difference value is greater than or equal to a preset vertically polarized power difference threshold, adjust a main lobe radiation angle of the vertically polarized antenna array until the vertically polarized power difference value is less than the preset vertically polarized power difference threshold.
The microwave antenna array communication system of any one of claims 1 to 4, wherein each of the antenna sub-arrays comprises a plurality of antenna oscillator elements and phase shifters in one-to-one correspondence with the plurality of antenna oscillator elements.
A communication method of the microwave antenna array communication system of any one of claims 1 to 6, comprising:
controlling, by the controller, a phase shifter of each of the antenna sub-arrays of the horizontally polarized antenna array to configure a phase of a horizontally polarized radio frequency signal transmitted by the each of the antenna sub-arrays, and controlling a phase shifter of each of the antenna sub-arrays of the vertically polarized antenna array to configure a phases of a vertically polarized radio frequency signal transmitted by the each of the antenna sub-arrays; and

transmitting, by a horizontally polarized radio frequency signal transmission device in the microwave transmission device, $\frac{N}{2}$
horizontal polarization radio frequency signals to an opposite end through each of the antenna sub-arrays in a corresponding horizontally polarized antenna array, and transmitting, by a vertically polarized radio frequency signal transmission device, $\frac{N}{2}$
vertically polarized radio frequency signals to the opposite end through each of the antenna sub-arrays in a corresponding vertically polarized antenna array.
The communication method of the microwave antenna array communication system of claim 7, wherein the controller is configured to control the phase shifter of the each of the antenna sub-arrays in the horizontally polarized antenna array to configure a phase difference between each of horizontally polarized radio frequency signals transmitted by the antenna sub-arrays and each of horizontally polarized radio frequency signals transmitted by adjacent antenna sub-arrays of the horizontally polarized antenna array to be $\frac{N}{2}$
and control the phase shifter of the each of the antenna sub-arrays in the vertically polarized antennas array to configure a phase difference between each of vertically polarized radio frequency signals transmitted by the antenna sub-arrays and each of horizontally polarized radio frequency signals transmitted by adjacent antenna sub-arrays of the vertically polarized antenna array to be $\frac{2 π}{N} .$
The communication method of the microwave antenna array communication system of claim 8, further comprising:
after the controller configures the phase of the horizontally polarized radio frequency signal transmitted by the each of the antenna sub-arrays in the horizontally polarized antenna array, acquiring a difference between $\frac{2 π}{N}$
and a receive phase angle of a horizontally polarized antenna array corresponding to the opposite end for receiving the horizontally polarized radio frequency signal transmitted by the each of antenna sub-arrays of the horizontally polarized antenna array, and in response to determining that the difference between $\frac{2 π}{N}$
and the receive phase angle of the horizontally polarized radio frequency signal is greater than a preset horizontally polarized phase angle offset threshold, adjusting the phase of the horizontally polarized radio frequency signal transmitted by the each of the antenna sub-arrays in the horizontally polarized antenna array according to the difference between $\frac{2 π}{N}$
and the receive phase angle of the horizontally polarized radio frequency signal until the difference between the $\frac{2 π}{N}$
and the receive phase angle of the horizontally polarized radio frequency signal is less than or equal to the preset horizontally polarized phase angle offset threshold; and

after the controller configures the phase of the vertically polarized radio frequency signal transmitted by the each of the antenna sub-arrays in the vertically polarized antenna array, acquiring a difference between $\frac{2 π}{N}$
and a receive phase angle of a vertically polarized antenna array corresponding to the opposite end for receiving the vertically polarized radio frequency signal transmitted by the each of antenna sub-arrays of the vertically polarized antenna array, and in response to determining that the difference between $\frac{2 π}{N}$
and the receive phase angle of the vertically polarized radio frequency signal is greater than a preset vertically polarized phase angle offset threshold, adjusting the phase of the vertically polarized radio frequency signal transmitted by the each of the antenna sub-arrays in the vertically polarized antenna array according to the difference between $\frac{2 π}{N}$
and the receive phase angle of the vertically polarized radio frequency signal until the difference between $\frac{2 π}{N}$
and the receive phase angle of the vertically polarized radio frequency signal is less than or equal to the preset vertically polarized phase angle offset threshold.
The communication method of the microwave antenna array communication system of any one of claims 7 to 9, further comprising:
acquiring, by the controller, transmit power P_ht of the horizontally polarized antenna array, receive power P_hr of the horizontally polarized antenna array corresponding to the opposite end and a path insertion loss L_hd to the opposite end, calculating a horizontally polarized power difference value ΔP_h according to a formula ΔP_h = P_ht -P_hr -L_hd , and in response to determining that the horizontally polarized power difference value is greater than or equal to a preset horizontally polarized power difference threshold, adjusting a main lobe radiation angle of the horizontally polarized antenna array until the horizontally polarized power difference value is less than the preset horizontally polarized power difference threshold; and

acquiring, by the controller, transmit power P_vt of the vertically polarized antenna array, receive power P_vr of the vertically polarized antenna array corresponding to the opposite end and a path insertion loss L_vd to the opposite end, calculating a vertically polarized power difference value ΔP_v according to a formula ΔP_v = P_vt -P_vr -L_vd , and in response to determining that the vertically polarized power difference value is greater than or equal to a preset vertically polarized power difference threshold, adjusting a main lobe radiation angle of the vertically polarized antenna array until the vertically polarized power difference value is less than the preset vertically polarized power difference threshold.
A computer-readable storage medium, which is configured to store computer-executable instructions for executing the method of any one of claims 7 to 10.