EP2768246B1 - Verfahren zum Betreiben eines Hörgeräts und Hörgerät - Google Patents

Verfahren zum Betreiben eines Hörgeräts und Hörgerät Download PDF

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EP2768246B1
EP2768246B1 EP13155108.7A EP13155108A EP2768246B1 EP 2768246 B1 EP2768246 B1 EP 2768246B1 EP 13155108 A EP13155108 A EP 13155108A EP 2768246 B1 EP2768246 B1 EP 2768246B1
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Prior art keywords
sample
input
buffer level
samples
signal
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English (en)
French (fr)
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EP2768246A1 (de
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Svend Feldt
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Sennheiser Communications AS
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Sennheiser Communications AS
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Priority to EP13155108.7A priority Critical patent/EP2768246B1/de
Priority to DK13155108.7T priority patent/DK2768246T3/da
Priority to US14/179,324 priority patent/US9894445B2/en
Priority to CN201410050440.8A priority patent/CN103987009B/zh
Publication of EP2768246A1 publication Critical patent/EP2768246A1/de
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    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04RLOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
    • H04R1/00Details of transducers, loudspeakers or microphones
    • H04R1/10Earpieces; Attachments therefor ; Earphones; Monophonic headphones
    • H04R1/1091Details not provided for in groups H04R1/1008 - H04R1/1083
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04RLOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
    • H04R25/00Deaf-aid sets, i.e. electro-acoustic or electro-mechanical hearing aids; Electric tinnitus maskers providing an auditory perception
    • H04R25/50Customised settings for obtaining desired overall acoustical characteristics
    • H04R25/505Customised settings for obtaining desired overall acoustical characteristics using digital signal processing
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04RLOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
    • H04R25/00Deaf-aid sets, i.e. electro-acoustic or electro-mechanical hearing aids; Electric tinnitus maskers providing an auditory perception
    • H04R25/55Deaf-aid sets, i.e. electro-acoustic or electro-mechanical hearing aids; Electric tinnitus maskers providing an auditory perception using an external connection, either wireless or wired
    • H04R25/554Deaf-aid sets, i.e. electro-acoustic or electro-mechanical hearing aids; Electric tinnitus maskers providing an auditory perception using an external connection, either wireless or wired using a wireless connection, e.g. between microphone and amplifier or using Tcoils

Definitions

  • the present invention relates to a method for operating a hearing device and to a corresponding hearing device. More specifically, the present invention relates to sample-rate conversion in such hearing devices.
  • the invention may e.g. be useful in applications such as e.g. a hearing aid or a listening device, which receives acoustic signals from a person's surroundings, modifies the acoustic signals electronically and transmits the modified acoustic signals into the person's ear or ear canal, or such as e.g. a headset, which receives audio signals electronically and transmits corresponding acoustic signals into the person's ear or ear canal.
  • Hearing devices that perform digital signal processing of digital audio signals received from another device are known in the art. Such hearing devices may be designed so that the processing rate, i.e. the rate with which the signal processing requires input samples, equals the input rate, i.e. the rate with which audio signal samples are received.
  • the processing rate i.e. the rate with which the signal processing requires input samples
  • the input rate i.e. the rate with which audio signal samples are received.
  • various factors such as production tolerances, variations in the clock frequencies of the transmitting and receiving devices and errors in the transmission, may cause the input rate to at least temporarily deviate from the processing rate.
  • the input signal is preferably pre-processed to ensure that the signal processing receives the required samples at the processing rate.
  • a known and rather simple solution is to provide the hearing device with an input buffer into which the received samples are written on arrival and from which the respective oldest unread sample is read by the signal processing when it requires a new sample.
  • a buffer under-run occurs, i.e. when the buffer is empty when a read operation is due, then the last sample read is repeated as input to the signal processing.
  • a buffer over-run occurs, i.e. when the buffer is full when a write operation is due, then the sample to be written is discarded (skipped).
  • This simple form of sample-rate conversion in the following referred to as skip-and-repeat, has the disadvantage that the skipping and repeating of samples causes clearly audible artefacts in the processed signal. It is further only successfully applicable when the input rate and the processing rate are close to each other.
  • the conversion ratio is defined as the ratio of the processing rate to the input rate.
  • the range of conversion-ratio variations with which a device has to cope may differ between different systems and scenarios. If the range of variations is large and/or if variations occur fast, then a more sophisticated control of the skipping and repeating in the upsampled skip-and-repeat method is required to avoid that multiple skip operations or multiple repeat operations occur immediately after each other, which could otherwise significantly reduce the quality of the processed signal.
  • the upsampled skip-and-repeat method is more sensitive to such variations than the simple method because in the upsampled method a single missing input sample may cause several consecutive repetitions of an upsampled signal sample.
  • the content of entire packets may be lost due to transmission errors.
  • One such known alternative method comprises performing sample-rate conversion with a piecewise constant conversion ratio, e.g. by upsampling and downsampling as described above, however without skipping and repeating upsampled samples.
  • the received input samples are buffered, and the conversion ratio is increased when the buffer level, i.e. the number of buffered samples, sinks below a predetermined lower threshold, and conversely decreased when the buffer level grows above a predetermined upper threshold.
  • the thresholds are preferably selected so that their difference is smaller than the total buffer size and they thus provide room both below the lower threshold and above the upper threshold. This extra room is used to allow for overshoot of the control algorithm.
  • the buffer size, the buffer thresholds and the step sizes with which the conversion ratio is respectively increased and decreased are chosen such that they together allow the method to adapt to expected variations in the input rate without risking actual buffer under-runs or over-runs.
  • the method also works when one of the upsampling and the downsampling factor equals unity.
  • Signal processing generally requires resources, such as logic circuits, memory space, computation time or power, each of which is typically a limited resource in hearing devices - and even more so in battery-powered hearing devices. It is therefore desirable to implement sample-rate conversion in hearing devices using methods that are particularly efficient. In most sample-rate conversion methods, and particular in such methods comprising upsampling and subsequent downsampling of the resampled signal, filtering is required to avoid aliasing artefacts in the processed signal. This filtering adds significantly to the complexity and the resource consumption of the sample-rate conversion, and the design of the filters is therefore of high importance in hearing devices. Upsampling followed by - or integral with - filtering to remove aliased frequencies is in the art generally referred to as interpolation. In the following, the term upsampling should be interpreted to cover both upsampling without such filtering and interpolation.
  • An efficient known method for sample-rate conversion comprises the use of so-called polyphase filters in the upsampling and in the downsampling steps.
  • Each received input sample is fed in parallel to a set of digital filters, which each represents a specific phase of an interpolation filter to be applied.
  • the outputs of the phase filters in the set are cyclically sampled at the resampling rate, one cycle for each input sample, and the thus resampled samples are cyclically fed to individual digital filters in a second set.
  • Each filter in the second set represents a specific phase of a downsampling filter to be applied, and the outputs of these phase filters are added at the processing rate, thus providing a sample-rate-converted signal at the processing rate.
  • the interpolation filter is typically configured to suppress frequencies above half the input rate
  • the downsampling filter is typically configured to suppress frequencies above half the processing rate.
  • the interpolation filter and the downsampling filter thus suppress aliasing artefacts in the processed signal.
  • the sum of the lengths of the phase filters in a set is equal to or slightly larger than the length of the respective interpolation or downsampling filter.
  • the interpolation filter and the downsampling filter are combined in a single filter configured such that it suppresses frequencies above half the minimum of the input rate and the processing rate.
  • the method allows the computation of the sample-rate converted signal to be performed basically by buffering the received input samples and at the processing rate computing a scalar product of a subset of the buffered samples and a subset of filter coefficients of the filter.
  • the actual subsets to be used are selected dynamically from respectively the input buffer and a set of filter coefficients, using respective indices that are dynamically updated for each output sample to be computed.
  • the above described methods based on polyphase filters may obviously be used in combination with the above described method wherein a piecewise constant conversion ratio is changed when the buffer level exceeds predetermined buffer thresholds.
  • the conversion ratio may be changed by changing the upsampling factor and/or the downsampling factor. Since, however, the lengths and the coefficients of the phase filters depend on the up- and downsampling factors, the lengths and the coefficients of the phase filters must either be recomputed or be read from a repository of pre-stored filter configurations each time the conversion ratio changes. This obviously requires computation time and/or storage space in addition to the resources required for the conversion itself.
  • changing the conversion ratio based on predetermined buffer thresholds as described above causes the latency, i.e. the time delay from an input sample is received until the corresponding output sample is delivered to the signal processing, to vary with time and, at least to some extent, unpredictably.
  • the buffer thresholds are chosen in advance and such that adaptation is possible when the input rate variations are within a range expected at the time of implementing the method or manufacturing the device. If the variations are larger than expected, the sample-rate conversion may not function properly, and if the variations are smaller than expected, the latency will be longer than needed.
  • a consistent and short latency is often important, particularly when a hearing-device user is able to see the source of the sound he or she is listening to, e.g. when the user listens to the sound of a video or a TV broadcast.
  • the size of the input buffer and the value of the lower buffer threshold are typically selected to allow for a predetermined number of consecutive retransmissions of the same lost packet before a buffer under-run occurs. This, however, causes the latency to be larger than needed during time periods without retransmission.
  • US 6,061,410 discloses a digital control arrangement used in connection with sampling rate conversion system having an input signal of one frequency converted to an output signal of another frequency.
  • US 7,246,057 discloses a receiver system in a communication system supporting packet-based communication, including a receiver, speech decoder and a jitter buffer for handling delay variations in the reception of a speech signal consisting of packets containing frames with encoded speech.
  • Controlling the conversion ratio in sample-rate conversion of an input signal buffered in an input buffer in a manner aiming at minimising the difference between estimated average or smoothed input-buffer levels and a target buffer level allows for robust sample rate conversion with precise control of the latency.
  • Performing the sample-rate conversion by upsampling the input signal and subsequently downsampling the upsampled signal in combination with controlling the conversion ratio by operations on the upsampled signal further allows for efficient and robust sample rate conversion with precise control of the latency at the cost of only minor artefacts in the processed signal.
  • the artefacts may be barely noticeable in speech and other every-day sounds, and the method may therefore advantageously be used in hearing devices.
  • a “hearing device” refers to a device, such as e.g. a hearing aid, a listening device or an active ear-protection device, which is adapted to improve, augment and/or protect the hearing capability of a user by receiving acoustic signals from the user's surroundings, generating corresponding audio signals, optionally modifying the audio signals and providing the received or modified audio signals as audible signals to at least one of the user's ears.
  • a “hearing device” further refers to a device such as an earphone or a headset adapted to receive audio signals electronically, optionally modifying the audio signals and providing the received or modified audio signals as audible signals to at least one of the user's ears.
  • Such audible signals may e.g. be provided in the form of acoustic signals radiated into the user's outer ears, acoustic signals transferred as mechanical vibrations to the user's inner ears through the bone structure of the user's head and/or through parts of the middle ear as well as electric signals transferred directly or indirectly to the cochlear nerve and/or to the auditory cortex of the user.
  • a hearing device may be configured to be worn in any known way, e.g. as a unit arranged behind the ear with a tube leading air-borne acoustic signals into the ear canal or with a loudspeaker arranged close to or in the ear canal, as a unit entirely or partly arranged in the pinna and/or in the ear canal, as a unit attached to a fixture implanted into the skull bone, as an entirely or partly implanted unit, etc.
  • a hearing device may comprise a single unit or several units communicating electronically with each other.
  • a hearing device comprises an input transducer for receiving an acoustic signal from a user's surroundings and providing a corresponding input audio signal and/or a receiver for electronically receiving an input audio signal, a signal processing circuit for processing the input audio signal(s) and an output means for providing an audible signal to the user in dependence on the processed audio signal(s).
  • Some hearing devices may comprise multiple input transducers, e.g. for providing direction-dependent audio signal processing.
  • the receiver may be a wireless receiver.
  • the receiver may be e.g. an input amplifier for receiving a wired signal.
  • an amplifier may constitute the signal processing circuit.
  • the output means may comprise an output transducer, such as e.g. a loudspeaker for providing an air-borne acoustic signal or a vibrator for providing a structure-borne or liquid-borne acoustic signal.
  • the output means may comprise one or more output electrodes for providing electric signals.
  • the vibrator may be adapted to provide a structure-borne acoustic signal transcutaneously or percutaneously to the skull bone.
  • the vibrator may be implanted in the middle ear and/or in the inner ear.
  • the vibrator may be adapted to provide a structure-borne acoustic signal to a middle-ear bone and/or to the cochlea.
  • the vibrator may be adapted to provide a liquid-borne acoustic signal in the cochlear liquid, e.g. through the oval window.
  • the output electrodes may be implanted in the cochlea or on the inside of the skull bone and may be adapted to provide the electric signals to the hair cells of the cochlea, to one or more hearing nerves and/or to the auditory cortex.
  • a “hearing system” refers to a system comprising one or two hearing devices
  • a “binaural hearing system” refers to a system comprising one or two hearing devices and being adapted to cooperatively provide audible signals to both of the user's ears.
  • Hearing systems or binaural hearing systems may further comprise "auxiliary devices", which communicate with the hearing devices and affect and/or benefit from the function of the hearing devices.
  • Auxiliary devices may be e.g. remote controls, remote microphones, audio gateway devices, mobile phones, public-address systems, car audio systems or music players.
  • Hearing devices, hearing systems or binaural hearing systems may e.g. be used for compensating for a hearing-impaired person's loss of hearing capability, augmenting or protecting a normal-hearing person's hearing capability and/or conveying electronic audio signals to a person.
  • the first embodiment of a hearing device 100 shown in FIG. 1 comprises a receiver 101, an input buffer 102, a sample processor 103, an estimator 104, a ratio controller 105, an amplifier 106 and an output transducer 107.
  • the hearing device 100 may e.g. be an earphone or a headset and may be intended to be worn in an operating position by a user, such as e.g. at, in, on or close to an ear of the user.
  • a transmitter 108 transmits samples of a digital audio signal to the hearing device 100, e.g. by means of radio signals or other wired or wireless electronic signals.
  • the transmitter may be any kind of device that is capable of transmitting a digital audio signal, such as e.g. a laptop computer, a mobile phone, a wireless microphone, a further hearing device etc.
  • the transmission may be made using any suitable protocol, including continuous data protocols or packet data protocols such as e.g. Bluetooth Advanced Audio Distribution Profile (A2DP), which is a part of the Bluetooth 1.0 standard, or Bluetooth Low Energy (BLE), which is a part of the Bluetooth 4.0 standard.
  • A2DP Bluetooth Advanced Audio Distribution Profile
  • BLE Bluetooth Low Energy
  • the transmitter 108 is external to the hearing device 100, but in some embodiments the transmitter 108 may instead be comprised by the hearing device 100.
  • the receiver 101 receives the samples of the digital audio signal and feeds the received samples as a digital input signal x(n) to the input buffer 102.
  • the receiver 101 optionally assisted by the transmitter 108, may apply any known method and technology, such as e.g. message checksums, forward error correction coding and decoding, retransmission of lost data packets etc., to ensure that errors in the received samples are corrected and/or to avoid that transmission errors cause gaps or other discontinuities in the audio signal provided in the digital input signal x(n).
  • the sample processor 103 processes the buffered samples to provide samples of a digital output signal y(m) such that the digital output signal y(m) is a sample-rate converted representation of the digital input signal x(n) with a predetermined target sample rate F y , which is preferably equal to the processing rate of the amplifier 106.
  • the sample processor 103 may use any of the sample-rate conversion methods known in the prior art, in particular any of the methods described further above, provided that the implemented method allows for a variation of the conversion ratio.
  • the sample processor 103 preferably uses one or more of the methods explained below in relation to FIGs. 3 , 4 and 5 .
  • the estimator 104 repeatedly determines an average or smoothed input-buffer level l(t) indicating an actual buffer level, i.e. the number N (see FIG. 6 ) of unprocessed samples in the input buffer 102.
  • the average or smoothed input-buffer level l(t) may be determined using any known averaging or smoothing method, such as e.g. repeatedly counting the number N of unprocessed samples in the input buffer 102 and computing an average over a sliding time window of the count results or low-pass filtering the count results.
  • the counting may be achieved by any known counting method, such as e.g.
  • the estimator 104 preferably uses one or more of the methods explained below in relation to FIG. 6 .
  • the ratio controller 105 is adapted to control the sample processor 103 in a manner aiming at minimising the difference between the estimated average input-buffer level l(t) and a target buffer level L.
  • the ratio controller 105 may achieve this by means of any known method of minimising said difference, such as e.g. adaptive least-mean-squares (LMS) methods or proportional control methods.
  • the ratio controller 105 preferably comprises or constitutes a proportional-derivative (PD) controller or a proportional-integral-derivative (PID) controller and thus preferably achieves its goal by means of respectively proportional-derivative control or proportional-integral-derivative control.
  • the ratio controller 105 may preferably control the sample processor 103 by providing one or more control parameters to the sample processor 103.
  • the control parameters may comprise any parameter that the sample processor 103 may use to determine that, and/or when, a skip or a repeat operation should be performed and/or to determine that the conversion ratio should be changed by other means, such as by changing an upsampling or a downsampling factor.
  • Suitable control parameters may indicate e.g. a target conversion ratio, an amount of change to be effected of the current conversion ratio or a time shift to be effected.
  • the sample processor 103 may adjust the conversion ratio to match as good as possible and within predetermined constraints the indicated target conversion ratio or a conversion ratio resulting from adding the indicated amount of change to the current conversion ratio, or to effect as good as possible and within predetermined constraints the indicated time shift in the processing of buffered input samples x(n).
  • the amplifier 106 receives the samples of the digital output signal y(m) from the sample processor 103 and provides a corresponding amplified signal to the output transducer 107.
  • the amplifier 106 may be or comprise any known type of amplifier suitable for providing an amplified signal corresponding to the digital output signal y(m), such as e.g. a digital pulse-width-modulator, a digital-to-analog converter followed by an analog amplifier or by an analog pulse-width-modulator, etc.
  • the amplifier 106 may further comprise a signal processor (not shown) adapted to modify the digital output signal y(m) in any known way, e.g.
  • the output transducer 107 converts the amplified signal into an acoustic signal.
  • the acoustic signal may enter the ear or the ear canal of the user, which may thus hear the acoustic signal and thus ideally perceive the original audio signal comprised in digital form by the samples received from the transmitter 108 or an enhanced or improved version thereof.
  • the output transducer 107 and optionally the amplifier 106 may be replaced by other suitable output means, such as the ones mentioned in the general description of hearing devices in the section "Disclosure of the invention".
  • the hearing device 100 may further comprise a latency controller 109 that controls the target buffer level L in dependence on an estimated quality of reception of the digital audio signal.
  • the latency controller 109 may preferably increase the target buffer level L in dependence on the estimated quality of reception decreasing and vice versa.
  • the quality of reception may preferably be estimated on the basis of repeated determinations of the actual buffer level N.
  • the latency controller 109 preferably uses one or more of the methods explained below in relation to FIG. 7 .
  • the hearing device 100 may further comprise a battery or an accumulator 110 that supplies electric power to the electronic circuits of the hearing device 100.
  • the hearing device 100 may receive electric power from an external device, such as the transmitter 108, e.g. via the transmission signal or by other wired or wireless power transmission means.
  • the second embodiment of a hearing device 200 shown in FIG. 2 comprises a receiver 101, an input buffer 102, a sample processor 103, an estimator 104, a ratio controller 105, an amplifier 106, an output transducer 107, an optional latency controller 109 and an optional battery or accumulator 110 having construction, connections and function as the corresponding elements of the first embodiment of FIG. 1 , except for deviations explained below.
  • the hearing device 200 may e.g. be a hearing aid, a listening device or an active ear-protection device and may be intended to be worn in an operating position at, in, on or near an ear of a user.
  • the hearing device 200 may receive samples of a digital audio signal from a transmitter 108 as described above in relation to FIG. 1 .
  • the hearing device 200 further comprises microphone 201, a preamplifier 202, a digitiser 203, a signal processor 204 connected with the amplifier 106 and the output transducer 107 to form an audio signal path.
  • the microphone 201 is arranged such that it may receive an acoustic input signal from the user's surroundings when the hearing device 200 is worn in the operating position by the user and thus provide a corresponding microphone signal to the preamplifier 202.
  • the preamplifier 202 amplifies the microphone signal and provides the amplified microphone signal to the digitiser 203.
  • the digitiser 203 digitises the amplified microphone signal and provides a digitised main audio signal to the signal processor 204, which may modify the main audio signal in accordance with the purpose of the hearing device 200, e.g. to improve, augment and/or protect the hearing capability of the individual.
  • the signal processor 204 provides the modified audio signal to the amplifier 106 instead of the digital output signal y(m).
  • the signal processor 204 is further connected to receive the digital output signal y(m) from the sample processor 103 and provides the modified audio signal to the amplifier 106 in dependence on a combination of the digital output signal y(m) and the digitised main audio signal. If, for instance, the transmitter 108 is a mobile phone, the signal processor 204 may switch between providing a modified or unmodified version of the digital output signal y(m) and a modified version of the digitised main audio signal in the modified audio signal to the amplifier 106, depending on whether the mobile phone is respectively engaged in a call or not. The signal processor 204 may, alternatively or additionally, combine the audio signals in the two input signals in any other known way, such as e.g. by adding them.
  • FIG. 3 illustrates upsampling of a digital input signal x(n) using a polyphase filter as it may be comprised by embodiments of the invention, such as the ones shown in FIGs. 1 and 2 .
  • the sample processor 103 and/or the input buffer 102 may thus comprise the elements shown in FIG. 3 .
  • the use of polyphase filters for upsampling is well known in the art, and so are its advantages with respect to efficiency and ease of implementation (see e.g. [1], chapter 10.5.2).
  • the receiver 101 may perform error correction, request retransmission of lost samples or packets of samples and/or discard duplicate samples or packets of samples in order to provide a digital input signal x(n) with as few errors and discontinuities as possible.
  • the digital input signal x(n) has a sample rate F x that may vary over time.
  • the sample rate F x may be directly reflected in the arrival times for the individual samples of the digital audio signal at the receiver 101 if the samples are transmitted continuously. If, however, the transmission from the transmitter 108 to the hearing device 100, 200 is packet-based, the exact instantaneous sample rate F x is undefined. In the latter case, or in the case that the samples arrive irregularly due to transmission errors or other causes, references in the following to the sample rate F x should be interpreted to include average values over an appropriate number of samples, e.g. over one or more packets.
  • the number U of upsampling phase filters P u equals the upsampling factor.
  • the interpolation filter Hu is a conventional finite-impulse-response (FIR) filter of length J, and its coefficients hu(j) are computed such that the interpolation filter Hu would suppress frequencies above half the sample rate F x and pass frequencies below unaltered if applied to a digital signal.
  • FIR finite-impulse-response
  • the interpolation filter Hu is padded with zero coefficients to achieve this.
  • FIG. 4 illustrates downsampling of a digital signal, such as the upsampled signal z(r) provided in FIG. 3 , using a polyphase filter as it may be comprised by embodiments of the invention, such as the ones shown in FIGs. 1 and 2 .
  • the sample processor 103 may thus comprise the elements shown in FIG. 4 .
  • the use of polyphase filters for downsampling is well known in the art, and so are its advantages with respect to efficiency and ease of implementation (see e.g. [1], chapter 10.5.2).
  • the number D of downsampling phase filters Q d equals the downsampling factor, and each downsampling phase filter Q d represents a specific phase of a downsampling filter H D .
  • the downsampling filter H D is a conventional FIR filter of length K. If the filter length K is initially not an integer multiple of D, then the downsampling filter H D is padded with zero coefficients to achieve this.
  • the coefficients h D (k) of the downsampling filter H D are computed such that the downsampling filter Hu would suppress frequencies above half the sample rate F y of the digital output signal y(m) and pass frequencies below unaltered if applied to a digital signal.
  • sample-rate conversion may be achieved by combining the elements shown in FIGs. 3 and 4 .
  • the sample processor 103 may control the conversion ratio U/D by performing skip and/or repeat operations on the upsampled samples z(r) and/or by changing the upsampling and/or downsampling factor U, D.
  • the sample processor 103 works in one mode, wherein it performs skip and/or repeat operations, as long as this suffices to minimise the difference between the estimated average or smoothed input-buffer level l(t) and the target buffer level L, and works in another mode, wherein it only changes the upsampling and/or the downsampling factor U, D, when said difference is too large or changing too fast to allow minimising by skip and/or repeat operations without either risking multiple skip or multiple repeat operations on consecutive samples z(r) or risking a too large deviation from the target buffer level L, e.g. a buffer over-run or a buffer under-run.
  • the ratio controller 105 controls the working mode of the sample processor 103 in dependence on said difference and/or on the speed of variation of said difference, e.g. by comparing one or both of these to predetermined thresholds.
  • the upsampling and downsampling described above and possible implementations are discussed in detail in [1], chapter 10.5.2.
  • sample-rate conversion based on a time-variant filter structure as disclosed in [1], chapter 10.5.3 may be used and may thus be comprised by embodiments of the invention, preferably by the sample processor 103 and/or the input buffer 102 described above.
  • EQ1 corresponds to equation (10.5.8) from [1], however with slightly deviating letter symbols. The meanings of the remaining letter symbols are explained below.
  • up- and downsampling as well as the accompanying filtering to prevent aliasing may be accomplished by buffering samples of the digital input signal x(n) and for each sample to provide of the digital output signal y(m), preferably at the target sampling rate F y , computing the sample y(m) as a scalar product of a dynamically selected subset of the buffered samples x(n) and a dynamically selected subset of filter coefficients h(j) of an FIR filter H of length J.
  • the filter coefficients h(j) of the filter H are preferably computed such that the filter H would suppress frequencies above the lower one of the cut-off frequencies of respectively the interpolation filter Hu and the downsampling filter H D described further above.
  • the filter H would thus suppress frequencies above half the minimum of the input sample rate F x and the target sampling rate F y and pass frequencies below unaltered if applied to a digital signal. If J is initially not an integer multiple of the upsampling factor U, then the filter H is padded with zero coefficients to achieve this.
  • the filter coefficients h(j) are preferably stored in a coefficient table 501, which may be comprised by the sample processor 103.
  • the high efficiency of this sample-rate conversion is partly due to the fact that only J/U multiplications need be performed for each output sample y(m), namely the J/U products between respective elements of the two subsets. Since each of the indices i and m is preferably traversed in steps of unity, the product iU may be computed by successively adding U, and the term ( mD ) U may be computed by successively adding D and subtracting U whenever the sum exceeds U. Similarly, the integer portion of the fraction mD/U may be computed by successively adding D, subtracting U when the sum exceeds U, and increasing a counter for each made subtraction; the value of the counter yields said integer portion.
  • the actual subsets to be used in the scalar product may be selected dynamically from respectively an input buffer 102 and the predetermined set of filter coefficients h(j) in the coefficient table 501, using respective indices that are dynamically updated using only addition, subtraction and comparison operations.
  • the computations may be made in an arithmetic unit 502, which may be comprised by the sample processor 103. Further details and possible implementations are discussed in detail in [1], chapter 10.5.3. In particular, preferred embodiments include those disclosed in [1], chapter 10.5.3, such as e.g. the block processing algorithm visualised in [1], Figure 10.16, the filter structure disclosed in [1], Figure 10.17 and the embodiment with U polyphase filters disclosed in the last part of the chapter.
  • error correction etc. may be performed, e.g. by the receiver 101, and the observations as to the meaning of the sample rate F x of the digital input signal x(n) are also the same as for FIG. 3 .
  • the sample processor 103 may control the conversion ratio by changing the upsampling and/or downsampling factor U, D. Since, however, the upsampled signal z(r) is not directly available in EQ1 or EQ3, nor in the implementations disclosed in [1], performing skip or repeat operations on the upsampled signal z(r) is not possible without further. However, EQ1 and EQ3 as well as the implementations disclosed in [1] may be modified to allow an operation similar to skipping and repeating samples of the upsampled signal z(r).
  • the primary sample index modifier S(m) is preferably updated for each output sample y(m) to compute.
  • Increasing or decreasing the primary sample index modifier S(m) before computation of an output sample y(m) causes the time-variant filter to operate on a subset of the buffered samples x(n) that is shifted one sample respectively forwards or backwards in time.
  • the effect hereof is equal to the effect of respectively a skip or a repeat operation on the digital input signal x(n) as described further above in the respect that it temporarily modifies the conversion ratio.
  • the digital output signal y(m) has less pronounced artefacts than a corresponding digital output signal y(m) obtained by sample-rate conversion using simple skip or repeat operations on the digital input signal x(n); the level of artefacts produced is in fact comparable to the level produced when using simple skip or repeat operations on an upsampled signal z(r).
  • the ratio controller 105 may thus control the conversion ratio in the sample processor 103 by causing a change of the primary sample index modifier S(m).
  • the ratio controller 105 may provide the primary sample index modifier S(m) as a control parameter to the sample processor 103.
  • the ratio controller 105 may provide a secondary sample index modifier s(m) as a control parameter to the sample processor 103, and the sample processor 103 may explicitly or implicitly compute the primary sample index modifier S(m) in dependence on the secondary sample index modifier s(m), e.g.
  • ⁇ s ( m ) provides the accumulated sum of the secondary sample index modifier s(m) over time.
  • EQ5 may be transformed into EQ4 by expressing the primary sample index modifier S(m) as a function of the secondary sample index modifier s(m), m, D and U.
  • the secondary sample index modifier s(m) may be an integer function, but is preferably a fractional function in order to allow for a more even distribution in time of shifts in the subset of the buffered samples x(n).
  • the value of the secondary sample index modifier s(m) for a specific index m thus indicates a desired time shift in the sample-rate conversion in the form of a fraction s(m)/U of an input sample interval 1/F x
  • the primary sample index modifier S(m) more or less corresponds to an integral over time of the secondary sample index modifier s(m) and thus indicates a sum of all preceding desired time shifts.
  • the index to x(n) may be computed merely by adding, subtracting and comparing values as described further above.
  • the ratio controller 105 may compute or update the primary or the secondary sample index modifier S(m), s(m) once for each output sample y(m) or at larger intervals depending on the required speed of controlling the conversion ratio. In some embodiments, the ratio controller 105 may compute or update the primary or the secondary sample index modifier S(m), s(m) during computation of an output sample y(m), such that the primary or the secondary sample index modifier S(m), s(m) becomes in effect a function of both indices m and i. In this case, less than the entire subset is shifted in time for the particular output sample y(m), and the artefacts produced are typically more pronounced. Nevertheless, such embodiments may be preferable in some situations.
  • the modifications may comprise increasing the length of the shift register and providing a multiplexer before each hold-and-sample device that allows the hold-and-sample device to select its input from any one of a number of consecutive memory cells in the shift register. In the case that the sample-rate conversion is implemented in software, the required modifications are rather straight forward.
  • the sample processor 103 works in one mode, wherein it performs the above described time shift operations, as long as this suffices to minimise the difference between the estimated average or smoothed input-buffer level l(t) and the target buffer level L, and works in another mode, wherein it only changes the upsampling and/or the downsampling factor U, D, when said difference is too large or changing too fast to allow minimising by the above described time shift operations without risking a too large deviation from the target buffer level L, e.g. a buffer over-run or a buffer under-run.
  • the ratio controller 105 controls the working mode of the sample processor 103 in dependence on said difference and/or on the speed of variation of said difference, e.g. by comparing one or both of these to predetermined thresholds.
  • the ratio controller 105 may provide one or more control parameters, such as e.g. the primary or the secondary sample index modifier S(m), s(m) or any other suitable control parameter, e.g. a target conversion ratio or a desired shift in the current conversion ratio.
  • the ratio controller 105 preferably computes or updates the one or more control parameters repeatedly, e.g. once for every output sample y(m) or any other intervals, such as e.g. larger or smaller intervals, depending on the required speed of controlling the conversion ratio.
  • the ratio controller 105 may preferably compute or update the one or more control parameters by means of the well known proportional-derivative control algorithm or the well known proportional-integral-derivative control algorithm.
  • the ratio controller 105 may preferably compute the difference e(k) between the average or smoothed input-buffer level l(t) received from the estimator 104 and the target buffer level L, which may e.g. be a fixed value stored in the ratio controller 105 or a varying value received from the latency controller 109, and use this difference e(k) as an error input to an error-minimising control algorithm that computes consecutive values of a control parameter w(k).
  • K i is set to zero.
  • the ratio controller 105 may use other known types of control algorithms, such as e.g. lead, lag or lead-lag compensators.
  • the estimator 104 may preferably provide the average or smoothed input-buffer level l(t) as illustrated in FIG. 6 .
  • the curve 601 illustrates an example of the variation over time t of the actual buffer level N.
  • the input buffer 102 may have a lower threshold 602 and an upper threshold 603 which may equal respectively zero and the buffer size, or alternatively lie away from these extremes in order to allow overshoot of the control algorithm applied by the ratio controller 105.
  • the receiver 101 feeds packets of samples x(n) into the input buffer 102, which causes the curve 601 to increase by the packet size, e.g. during the write operation 604. The packets arrive irregularly due to transmission errors.
  • the sample processor 103 reads a number N of buffered samples x(n) at regular intervals, which causes the curve 601 to decrease by the number of samples read, e.g. during the read operation 605.
  • the time line 606 is divided into three time periods, wherein vertical dotted lines illustrate different methods for determining an actual input-buffer level N.
  • the actual buffer level N is sampled every time a read operation 605 by the sample processor 103 occurs.
  • the actual buffer level N is sampled every time a write operation 604 by the receiver 101 occurs.
  • the actual buffer level N is sampled at regular time intervals which are preferably substantially shorter than the intervals between successive write operations 604 and substantially shorter than the intervals between successive read operations 605.
  • the estimator may use any of the illustrated sample methods, or any combination thereof, to provide input data to a smoothing or averaging algorithm for providing the average or smoothed input-buffer level l(t).
  • the estimator 104 may preferably compute an average over a sliding time window of the sample results or, alternatively or additionally, low-pass filter the sample results to obtain the average or smoothed input-buffer level l(t). Due to the different sampling times relative to upwards and downwards shifts in the actual buffer level N, the illustrated sample methods typically cause the smoothing or averaging algorithm to provide different results, and the estimator 104 may therefore preferably subtract from the output of the average or low-pass filter a constant value that reflects a typical estimation error for the implemented sampling method before providing the average or smoothed input-buffer level l(t) to the ratio controller 105.
  • the width of the time window, the characteristics of the low-pass filter and/or typical estimation errors may be determined in advance by experimentation or analytically from knowledge of the packet sizes, of the number of samples read in batch by the sample processor 105, of expected variations in the arrival times of the digital input signal x(n), and of the characteristics of the hardware and/or software involved in the sample-rate conversion.
  • the latency controller 109 may preferably provide the target buffer level L adaptively as illustrated in FIG. 7 .
  • the curve 601 illustrates an example of the variation over time t of the actual buffer level N
  • the input buffer 102 may have a lower threshold 602 and an upper threshold 603 as described above.
  • the sample processor 103 reads a number of buffered samples x(n) at regular intervals, e.g. during the read operation 605, and the receiver 101 feeds packets of samples x(n) into the input buffer 102, e.g. during the write operation 604. In the shown example, the packets arrive irregularly due to a decrease over time of the quality of reception of the digital audio signal.
  • the difference between the local minimum 701 and the local maximum 702 of the actual buffer level N is relatively small, whereas in the rightmost part, the difference between the local minimum 703 and the local maximum 704 of the actual buffer level N is relatively large.
  • the average or smoothed input-buffer level l(t) provided by the estimator 104 is also shown.
  • the latency controller 109 may preferably control the target buffer level L in dependence on the estimated quality of reception of the digital audio signal in order to achieve a short latency when the quality of reception is high and a long latency when the quality of reception is low.
  • the latency is correlated with the target buffer level L because the ratio controller 105 works to minimise the difference e(k) between the estimated average or smoothed input-buffer level l(t) and the target buffer level L, and the latency controller 109 may thus preferably decrease the target buffer level L when the estimated quality of reception is high and increase the target buffer level L when the estimated quality of reception is low.
  • the latency controller 109 may achieve control of the target buffer level L in dependence on the estimated quality of reception of the digital audio signal in an indirect manner, e.g. by determining local minima 701, 703 in the actual buffer level N and controlling the target buffer level L in dependence on the determined local minima 701, 703.
  • the latency controller 109 may e.g. increase the target buffer level L whenever a determined local minimum 701, 703 is below a predetermined threshold 705 and iteratively decrease the target buffer level L as long as the determined local minima 701, 703 are above the predetermined threshold 705.
  • the predetermined threshold 705 may equal the lower threshold 602 or may be higher in order to allow overshoot of the latency control algorithm without risking buffer under-run.
  • the latency controller 109 may e.g. immediately increase the target buffer level L by an amount equal to the amount which a determined local minimum 701, 703 is below the predetermined threshold 705 and subsequently decrease the target buffer level L, e.g. exponentially, towards a minimum target buffer level L min , e.g. with a time constant between 1 and 3 s, between 2 and 6 s or between 3 and 10 s.
  • the latency controller 109 may track the local minima 701, 703 and employ the above described increases and decreases of the target buffer level L in dependence on a low-pass filtered or otherwise smoothed minimum track value.
  • the latency controller 109 may further determine local maxima 702, 704 in the actual buffer level N and control the target buffer level L in dependence on both the determined local minima 701, 703 and the determined local maxima 702, 704. For instance, the latency controller 109 may increase the time constant for decreasing the target buffer level L in dependence on the difference between the determined local maxima 702, 704 and the determined local minima 701, 703 increasing. Alternatively, or additionally, the latency controller 109 may increase the target buffer level L in dependence on the difference between the determined local maxima 702, 704 and the determined local minima 701, 703 increasing. The latter could e.g.
  • the latency controller 109 may determine the difference between local maxima 702, 704 and local minima 701, 703 by tracking the local minima 701, 703 and the local maxima 702, 704 and determine said difference in dependence on low-pass filtered or otherwise smoothed minimum and maximum track values.
  • the latency controller 109 may use the difference between the average or smoothed input-buffer level l(t) provided by the estimator 104 and the determined local minima 701, 703, since this difference is highly correlated with the former difference when the ratio controller 105 works as described further above.
  • the latency controller 109 may use any known methods and/or means to track or otherwise determine the development over time of the local minima 701, 703 and/or the local maxima 702, 704 and may control the target buffer level L in dependence on the determined development.
  • the latency controller 109 may receive a quality signal from the receiver 101 indicating e.g. the frequency of transmission errors, lost packets and/or packet retransmissions etc., and the latency controller 109 may estimate the quality of reception of the digital audio signal based on the quality signal alone or on the quality signal in combination with the methods described above using the actual buffer level N.
  • the latency controller 109 may further low-pass filter the target buffer level L with an appropriate cut-off frequency before providing it to the ratio controller 105 in order to avoid abrupt changes of the conversion ratio.
  • the hearing device 100, 200 is preferably implemented using mainly digital circuits operating in the discrete time domain, but some parts hereof may alternatively be implemented as analog circuits operating in the continuous time domain.
  • Digital functional blocks of the hearing device 100, 200 such e.g. the functional blocks shown in the drawings, may be implemented in any suitable combination of hardware, firmware and software and/or in any suitable combination of hardware units.
  • any single hardware unit may execute the operations of several functional blocks in parallel or in interleaved sequence and/or in any suitable combination thereof.
  • the input buffer 102 may preferably be implemented as a ring buffer or as a first-in-first-out (FIFO) buffer.
  • FIFO first-in-first-out
  • the hearing device 100, 200 may be part of a binaural hearing system, and the hearing device 100, 200 may e.g. receive the digital audio signal from the other hearing device in the binaural hearing system.
  • the binaural hearing system may comprise one or more auxiliary devices.
  • Embodiments of the invention may be used in any type of device, and most advantageously in battery-driven and/or portable devices.

Claims (15)

  1. Verfahren zum Betrieb eines Hörgeräts (100, 200), wobei das Verfahren umfasst
    ein Empfangen von Abtastungen eines Digitalaudiosignals,
    ein Einspeisen von empfangenen Abtastungen als ein Digitaleingangssignal (x(n)) in einen Eingangspuffer (102),
    ein Verarbeiten der gepufferten Abtastungen (x(n)), um Abtastungen eines Digitalausgangssignals (y(m)) derart bereitzustellen, dass das Digitalausgangssignal (y(m)) eine abtastratenkonvertierte Repräsentation des Digitaleingangssignals (x(n)) mit einer vorbestimmten Sollabtastrate (Fy) ist,
    ein Schätzen eines durchschnittlichen oder geglätteten Eingangspufferpegels (l(t)), der die Anzahl (N) von unverarbeiteten Abtastungen (x(n)) in dem Eingangspuffer (102) indiziert, und
    ein Steuern des Verarbeitens der gepufferten Abtastungen (x(n)) auf eine Weise, die auf ein Minimieren der Differenz (e(k)) zwischen dem geschätzten durchschnittlichen oder geglätteten Eingangspufferpegel (l(t)) und einem Sollpufferpegel (L) abzielt,
    ein Vergleichen eines Parameters, der eine Differenz zwischen dem geschätzten durchschnittlichen oder geglätteten Eingangspufferpegel und dem Sollpufferpegel (L) oder eine Variationsgeschwindigkeit der Differenz zwischen dem durchschnittlichen Eingangspufferpegel und dem Sollpegel (L) repräsentiert, mit einer vorbestimmten Schwelle,
    ein Steuern eines Umwandlungsverhältnisses zwischen der Eingangsrate und der Abtastrate gemäß einer aus einer ersten Betriebsart und einer zweiten Betriebsart, die gemäß einem Ergebnis des Vergleichs zwischen dem Parameter und der vorbestimmten Schwelle ausgewählt ist, wobei
    bei der ersten Betriebsart das Umwandlungsverhältnis durch Durchführen eines Weglass- und/oder Wiederholbetriebs zur Minimierung der Differenz zwischen dem durchschnittlichen Eingangspufferpegel und dem Sollpufferpegel gesteuert wird,
    bei der zweiten Betriebsart das Umwandlungsverhältnis durch Ändern eines Hochabtastfaktors und/oder eines Runterabtastfaktors gesteuert wird,
    die erste Betriebsart für ein Steuern des Umwandlungsverhältnisses ausgewählt wird, wenn der Parameter unterhalb der vorbestimmten Schwelle ist, und
    die zweite Betriebsart zur Steuerung des Umwandlungsverhältnisses ausgewählt wird, wenn der Parameter die vorbestimmte Schwelle überschreitet und dadurch indiziert, dass die Differenz zwischen dem durchschnittlichen Eingangspufferpegel und dem Sollpufferpegel zu groß oder eine Änderung zu schnell ist für eine Minimierung der Differenz durch Weglass- und/oder Wiederholbetriebe.
  2. Verfahren nach Anspruch 1, wobei das Verarbeiten ein Hochabtasten der gepufferten Abtastungen (x(n)) zur Bereitstellung eines hochabgetasteten Signals (z(r)) und ein Runterabtasten des hochabgetasteten Signals (z(r)) umfasst, und wobei das Steuern des Verarbeitens ein Durchführen von Betrieben bezüglich des hochabgetasteten Signals (z(r)) umfasst.
  3. Verfahren nach Anspruch 2, wobei das Hochabtasten und/oder das Runterabtasten die Verwendung eines Polyphasenfilters (Pu, Qd) umfasst.
  4. Verfahren nach einem der vorstehenden Ansprüche, wobei das Verarbeiten ein Berechnen jeder Abtastung des Ausgangssignals (y(m)) als ein Skalarprodukt eines Untersatzes der gepufferten Abtastungen (x(n)) und eines Untersatzes von Filterkoeffizienten (h(j)) umfasst, wobei die Untersätze neu für jede berechnete Abtastung (y(m)) aus jeweils dem Eingangspuffer (102) und einem Satz von vorbestimmten Koeffizienten (h(j)) ausgewählt werden, und wobei die Auswahl des Untersatzes der gepufferten Abtastungen (x(n)) abhängig von dem geschätzten durchschnittlichen oder geglätteten Eingangspufferpegel (l(t)) ist.
  5. Verfahren nach einem der vorstehenden Ansprüche, wobei das Steuern des Verarbeitens der gepufferten Abtastungen (x(n)) eine Proportionalableitungssteuerung oder eine Proportionalintegralableitungssteuerung umfasst.
  6. Verfahren nach einem der vorstehenden Ansprüche, ferner mit einem Erhöhen des Sollpufferpegels (L) in Abhängigkeit von einer Verringerung einer geschätzten Qualität eines Empfangs des Digitalaudiosignals und andersherum.
  7. Verfahren nach Anspruch 6, wobei die Qualität des Empfangs basierend auf wiederholten Bestimmungen der Anzahl (N) von unverarbeiteten Abtastungen (x(n)) in dem Eingangspuffer (102) geschätzt wird.
  8. Hörgerät (100, 200) mit einem Empfänger (101), einem Eingangspuffer (102), und einem Abtastprozessor (103), wobei der Empfänger (101) dazu eingerichtet ist, um Abtastungen eines Digitalaudiosignals zu empfangen und empfangene Abtastungen als ein Digitaleingangssignal (x(n)) zu dem Eingangspuffer (102) einzuspeisen, wobei der Abtastprozessor (103) dazu eingerichtet ist, um die gepufferten Abtastungen (x(n)) zu verarbeiten, um Abtastungen eines Digitalausgangssignals (y(m)) derart bereitzustellen, dass das Digitalausgangssignal (y(m)) eine abtastratenkonvertierte Repräsentation des Digitaleingangssignals (x(n)) mit einer vorbestimmten Sollabtastrate (Fy) ist,
    dadurch gekennzeichnet, dass das Hörgerät (100, 200) ferner aufweist
    ein Schätzelement (104), das dazu eingerichtet ist, um einen durchschnittlichen oder geglätteten Eingangspufferpegel (l(t)) zu schätzen, der die Anzahl (N) von unverarbeiteten Abtastungen (x(n)) in dem Eingangspuffer (102) indiziert,
    ein Verhältnissteuerelement (105), das dazu eingerichtet ist, um den Abtastprozessor (103) auf eine Weise zu steuern, die auf eine Minimierung der Differenz (e(k)) zwischen dem geschätzten durchschnittlichen oder geglätteten Eingangspufferpegel (l(t)) und einem Sollpufferpegel (L) abzielt,
    ein Vergleichselement, das dazu eingerichtet ist, um einen Parameter, der eine Differenz zwischen dem geschätzten durchschnittlichen oder geglätteten Eingangspufferpegel und dem Sollpufferpegel (L) oder eine Variationsgeschwindigkeit der Differenz zwischen dem durchschnittlichen Eingangspufferpegel und dem Sollpegel (L) repräsentiert, mit einer vorbestimmten Schwelle zu vergleichen, und
    ein Verhältnissteuerelement, das dazu eingerichtet ist, um ein Umwandlungsverhältnis zwischen der Eingangsrate und der Abtastrate gemäß einer aus einer ersten Betriebsart und einer zweiten Betriebsart, die gemäß einem Ergebnis des Vergleichs zwischen dem Parameter und der vorbestimmten Schwelle ausgewählt ist, zu steuern,
    wobei bei der ersten Betriebsart das Umwandlungsverhältnis durch Durchführung eines Weglass- und/oder Wiederholbetriebs zur Minimierung der Differenz zwischen dem durchschnittlichen Eingangspufferpegel und dem Sollpufferpegel gesteuert ist, und bei der zweiten Betriebsart das Umwandlungsverhältnis durch Änderung eines Hochabtastfaktors und/oder eines Runterabtastfaktors gesteuert ist,
    wobei die erste Betriebsart zur Steuerung des Umwandlungsverhältnisses ausgewählt ist, wenn der Parameter unterhalb der vorbestimmten Schwelle ist, und die zweite Betriebsart zur Steuerung des Umwandlungsverhältnisses ausgewählt ist, wenn der Parameter die vorbestimmte Schwelle überschreitet und dadurch indiziert, dass die Differenz zwischen dem durchschnittlichen Eingangspufferpegel und dem Sollpufferpegel zu groß oder eine Änderung zu schnell ist für eine Minimierung der Differenz durch Weglass- und/oder Wiederholbetriebe.
  9. Hörgerät nach Anspruch 8, wobei der Abtastprozessor (103) ferner dazu eingerichtet ist, um die gepufferten Abtastungen (x(n)) hochabzutasten, um ein hochabgetastetes Signal (z(r)) bereitzustellen, und das hochabgetastetes Signal (z(r)) runterabzutasten, und wobei das Verhältnissteuerelement (105) ferner dazu eingerichtet ist, um den Abtastprozessor (103) dazu zu bringen, Betriebe bezüglich des hochabgetasteten Signals (z(r)) durchzuführen.
  10. Hörgerät nach Anspruch 9, wobei der Abtastprozessor (103) einen Polyphasenfilter (Pu, Qd) für das Hochabtasten und/oder das Runterabtasten umfasst.
  11. Hörgerät nach einem der Ansprüche 8 bis 10, wobei der Abtastprozessor (103) ferner dazu eingerichtet ist, um jede Abtastung des Ausgangssignals (y(m)) als ein Skalarprodukt eines Untersatzes der gepufferten Abtastungen (x(n)) und eines Untersatzes von Filterkoeffizienten (h(j)) zu berechnen, die Untersätze neu für jede berechnete Abtastung (y(m)) aus jeweils dem Eingangspuffer (102) und einem Satz von vorbestimmten Koeffizienten (h(j)) auszuwählen, und den Untersatz der gepufferten Abtastungen (x(n)) in Abhängigkeit von dem geschätzten durchschnittlichen oder geglätteten Eingangspufferpegel (l(t)) auszuwählen.
  12. Hörgerät nach Anspruch 11, wobei der Abtastprozessor (103) ferner dazu eingerichtet ist, um den Untersatz der gepufferten Abtastungen (x(n)) in Abhängigkeit von dem geschätzten durchschnittlichen oder geglätteten Eingangspufferpegel (l(t)) in der Zeit zu verschieben.
  13. Hörgerät nach einem der Ansprüche 8 bis 12, wobei das Verhältnissteuerelement (105) ein Proportionalableitungssteuerelement oder ein Proportionalintegralableitungssteuerelement umfasst.
  14. Hörgerät nach einem der Ansprüche 8 bis 13, ferner mit einem Latenzsteuerelement (109), das dazu eingerichtet ist, um den Sollpufferpegel (L) in Abhängigkeit von einer Verringerung einer geschätzten Qualität eines Empfangs des Digitalaudiosignals zu erhöhen und andersherum.
  15. Hörgerät nach einem der Ansprüche 8 bis 14, wobei das Latenzsteuerelement (109) ferner dazu eingerichtet ist, um die Qualität des Empfangs basierend auf wiederholten Bestimmungen der Anzahl (N) von unverarbeiteten Abtastungen (x(n)) in dem Eingangspuffer (102) zu schätzen.
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