EP2662929A1 - Phased array antenna and method for processing received signals in a phased array antenna - Google Patents

Abstract

The received signals processing method involves assigning, once or permanently, one individual phase value, which is normally distributed from a phase range within a decorrelation unit (8), to each receiving element (E-1 to E-N) of a phased-array antenna. The normally distributed phase values are added up to oscillator signals (LO-1 to LO-N). The phase displacement corresponding to a receiving direction of the antenna is deployed on complex baseband-signals (IQ-1 to IQ-N) within another decorrelation unit (9) in each receiving path. An independent claim is included for a phased-array antenna.

Description

The invention relates to a method for processing received signals in a phased array antenna according to the features of patent claim 1 and a phased array antenna according to the features of patent claim 3.

STATE OF THE ART

Out DE 600 25 064 T2 and DE 10 2007 046 566 B4 radars are known using digital beamforming.

In Fig. 1 The structure of an N-channel phased array antenna with digital beam shaping is shown schematically. The use of a plurality of receiving channels shown in FIG. 5 achieves a directivity in the surroundings of the radar device. Under ideal conditions, the dynamic range over a single receiver system increases by a factor of 10log ₁₀ (N) dB, where N expresses the number of channels used and n expresses the run index for the nth channel. Using the example of a linear phased array antenna 1 with antenna spacing d = λ / 2 (λ: wavelength associated with the carrier frequency f _c used via the speed of light c, corresponding to λ = c / f _c ), further processing is described. The high-frequency signals X ₁ ,..., X _N provided by the antenna elements E ₁ ,..., E _N are fed to the analogue receiver stages ARX ₁ ,..., ARX _N. In the complete analog receiver unit 2 the high frequency signals X ₁ , ..., X _{N are converted} to a lower intermediate frequency U ₁ , ..., U _N. For this purpose, a mixing signal BO from the block central base oscillator 6 of the analog receiver unit 2 is supplied. The internal distribution of the central BO takes place at each analogue receiver stage ARX ₁ ,..., ARX _N. The intermediate frequency signals U ₁ ,..., U _n generated by the mixing process of X ₁ ,..., X _N with the oscillator signals derived from BO are subsequently supplied to the digital receiver unit 3. There, the analog signals with the analog-to-digital converters ADC ₁ , ..., ADC _N are first converted into digital signals. In the downstream digital preprocessing unit PP ₁ ,..., PP _N , the complex baseband signals IQ ₁ ,..., IQ _{N are} generated. A complex baseband signal IQ _n is vectorially composed of a real part Re {IQ _n } and an imaginary part Im {IQ _n }. The digital beamforming processing unit 4 is part of a signal processor and uses all signals IQ ₁ , ..., IQ _N provided by the N individual channels to form a number J beams B ₁ , ..., B _J. As a rule, the number of beams is: J <N. These beams provide directional access to distance and speed information. The concept is explained independently of the selected modulation type.

As is known theoretically for linear phased arrays, an incident signal wave produces a direction-dependent phase shift in each case in the nth channel (φ _n = n * 2 * π * f _c * d / c * sin (Θ), where f _{c is} the carrier frequency, d represents the antenna distance and Θ the angle of incidence Fig. 2 , with the receiving channel 7 based on the index n = 1 used.

The signal LO ₁ provided by the local LO ₁ is extracted from the central base oscillator BO (reference 6) in FIG Fig. 1 , derived. The central base oscillator 6 provides a local LO signal of identical frequency to each receiving element. By multiplying LO ₁ and the phase-shifted X ₁ , the intermediate frequency signal U _{1 is} generated. The digitization of the value and time-continuous signal U ₁ in the signal D ₁ takes place in the analog-to-digital converter (ADC ₁ ). The generation of the complex baseband signal Re {IQ ₁ } + j * Im {IQ ₁ } takes place in the preprocessing unit PP1. By definition, ADC ₁ and PP ₁ form a digital individual receiver 10 in the network. The processing work Digital Beamforming 4 is described below Fig .1 explained in more detail. The direction-dependent phase shifts φ ₁ -φ _N remain in the conversion of the signals X ₁ , ..., X _N to IQ ₁ , ..., IQ _N obtained and can in the mixer units P ₁ , ..., P _N in 4 each be applied inversely. The multiplication of the complex signals IQ ₁ ~IQ _N with inverse rotation hands R _n = exp (-j * n * 2 * π * f _c * d / c * sin (Θ)) produces an inverse phase shift of φ _{inv, n} = φ _n in each n-th channel and allows a coherent addition (constructive superposition) of all received signals in the subsequent summing unit 12. The generated output signals B ₁ -B _J are fed to a subsequent processor unit (radar processor).

^1xN) ergibt sich ein Richtdiagramm nach G(Θ)=10log₁₀(sin²(N*π*d/λ*sin(Θ-Θ₀))/ (N²*sin²(π*d/λ*sin(Θ-Θ₀)))), wobei Θ₀ den tatsächlichen Einfallswinkel der elektromagnetischen Welle und Θ die eingestellte Vorzugsrichtung der digitalen Strahlformung.The totality of all applied inverse phase shifts can thus be in a vector Φ _inv = [φ _inv , 2 φ _inv , 3 φ _inv , 4 φ _inv , ₅ ... φ _inv , N] and thus a pointer vector R = A _taper * exp (jΦ _inv ), which is applied by multiplication to the mixer units P ₁ ~P _n . If no amplitude weighting is provided (A _taper = [11 ... 1] with A _taper ∈

^1xN ) results in a directional diagram according to G (Θ) = 10log ₁₀ (sin ² (N * π * d / λ * sin (Θ-Θ ₀ )) / (N ² * sin ² (π * d / λ * sin ( Θ-Θ ₀ )))), where Θ ₀ the actual angle of incidence of the electromagnetic wave and Θ the set preferred direction of digital beam forming.

Conventional methods have a drawback with respect to the useful noiseless dynamic range. When using beamforming, distortion products (HD ₂ , HD ₃ ... HD _i = Harmonic Distortion, interfering signals generated in nonlinear systems at multiples i of the signal frequency) experience an inverse phase shift corresponding to the useful signal. The phases associated with the distortion products have a factor Ψ deviating from the vector φ _inv , e.g. Eg Ψ = 2 for HD2. Depending on the viewing angle Θ, this leads to a partially destructive or constructive superimposition of these distortion products.

Disadvantage of a signal processing according to the prior art is therefore that there is a constructive interference of the input signal in the addition, but in the distortion products, however, in addition to a partially destructive Interference also leads to a constructive interference at the viewing angle Θ = 0 _° and larger angles according to the zeros in the function P (Θ) = 10log ₁₀ (sin ² (Ψ * N * π * d / λ * sin (Θ-Θ ₀ ) ) / (N ² * Sin ² (Ψ * π * d / λ * sin (Θ-Θ ₀ )))). This inevitably worsens the spur-free dynamic range (SFDR), which in this case represents the ratio of the power magnitude of the largest harmonic to the power level of the received signal (fundamental). The dynamic range increase 10log ₁₀ (N) dB expected by the digital beamforming can thus not be guaranteed, since distortion products are not decorrelated from channel to channel at all viewing angles Θ and thus are added up in the same way as the useful signal.

DESCRIPTION OF THE INVENTION

The object of the invention is to provide a method in which occurring in the receiver harmonics are suppressed by the selected signal processing concept and the trouble-free dynamic range is improved over all angles Θ. Another object is to provide a corresponding antenna.

The objects are achieved by the method according to the features of the valid patent claim 1 and the device according to claim 3. Advantageous embodiments of the invention are the subject of dependent claims.

Fig. 1

schematisch der Aufbau einer N-kanaligen Phased-Array Antenne mit digitaler Strahlformung,

Fig. 2

schematisch der Aufbau einer N-kanaligen Phased-Array Antenne mit digitaler Strahlformung und richtungsabhängigen Phasenverschiebungen,

Fig. 3

schematisch der erfindungsgemäße Aufbau einer N-kanaligen Phased-Array Antenne mit digitaler Strahlformung inklusive der Dekorrelationseinheiten,

Fig. 4

beispielhafte erfindungsgemäße Phasenbelegung im Empfangsteil

Fig. 5

eine schematische Darstellung eines beispielhaften Frontends, des Mischers und des Oszillators einer Phased-Array Antenne gemäß Fig.1.

The invention and further advantageous embodiments of the invention will be explained in more detail with reference to the drawing. Show it:

Fig. 1

the construction of an N-channel phased array antenna with digital beam shaping,

Fig. 2

the structure of an N-channel phased array antenna with digital beam shaping and direction-dependent phase shifts,

Fig. 3

FIG. 2 schematically the structure according to the invention of an N-channel phased array antenna with digital beam shaping including the decorrelation units, FIG.

Fig. 4

exemplary phase assignment according to the invention in the receiving part

Fig. 5

a schematic representation of an exemplary front end, the mixer and the oscillator of a phased array antenna according to Fig.1 ,

Fig. 3 shows schematically the construction according to the invention of an N-channel phased array antenna with digital beam shaping including the decorrelation units 8,9. According to the invention, receive signals X ₁ ,..., X _{N are processed} in a phased array antenna having a plurality of receive elements E ₁ ,..., E _N , each having an associated receive path, an analog intermediate frequency signal U ₁ in each receive path , ..., U _N produced by mixing the reception signal X _1, ..., X _N with an oscillator signal LO _1, ... LO _N and then by digitization and possibly digital blend into a complex base band signal IQ ₁ , ..., IQ _N , wherein in each receive path to the complex baseband signal IQ ₁ , ..., IQ _N one of the receive direction of the antenna corresponding phase shift is applied. The invention is characterized in that, when the method is carried out for the first time, each receiving element of the phased array antenna is exactly one individual phase variable from -π to: + π normal-distributed phase value φ _{r, 1} , .., φ _{r, N} within the first Is assigned to the oscillator signal these normally distributed phase values <φ _{r, 1} , .., φ _{r, N} and that within the second decorrelation unit 9 in each receive path to the complex baseband signal IQ ₁ , .. , IQ _{N is} one of the receiving direction of the antenna corresponding phase shift φ _{rx, 1} , .., φ _{rx, N} in which the normally distributed phase values φ _{r, 1} , .., φ _{r, N are} considered inversely applied.

The first decorrelation unit 8 and the second decorrelation unit 9 differ in that the first decorrelation unit 8 has a channel-dependent phase shift added in the analog part of the receiver whereas the second decorrelation unit 9 inversely applies the added channel-dependent phase shift in the digital part of the receiver and thus reverses it. The introduced blocks RX ₁ , ..., RX ₂ each denote the union of ADC and PP in each channel (see also FIG Fig. 1 .).

In applying the phase shift φ _{rx, 1} , .., φ _{rx, N} to the complex baseband signal IQ ₁ , ..., IQ _N , an amplitude weighting may additionally be performed, the application of a weighting function does not affect the subject matter of this invention.

The object underlying the invention digital each applied phase vector in the second decorrelation unit 9 can thus be expressed as Φ _dekorr = Φ _inv + Φ _rx, wherein Φ _{r x} = [φ _{rx, 1} φ _{rx, 2} φ _{rx, 3} ... φ _{rx, N} ] represents the additive, a normally distributed random process taken, and already applied in the first decorrelation unit 8 analog applied data set of phase values, which lie between +/- π. The analog-applied phase vector in the first decorrelation unit 8 corresponds to the phase vector in the second decorrelation unit 9 with the opposite sign according to Φ _r = -φ _rx . Furthermore, Φ _{inv is to be assigned to} the actual beam shaping and into those already in Fig. 2 , described complex rotary hands R ₁ ~R _N included. Φ _rx contains the additionally introduced phase values. The phase shift applied in vector form on the digital side can thus be expressed as a multiplication by the signal _vector R _dekorr = exp (j (φ _inv + φ _rx )) = exp (jφ _dekorr ) with the individual _pointers [R _{dekorr, 1} R _dekorr , ₂ ... R _{dekorr, N} ]. On the analog side, phase shifts are likewise applied to the mixer signals LO ₁ -LO _N via the vector Φ _r = [φ _r , ₁ φ _{r, 2} φ _{r, 3} ... Φ _{r, N} ]. The mixer signals with introduced phase shifts φ _{r, 1} ,..., Φ _{r, N} are combined in the signal _vector LO _dekorr = [LO _decorr, 1 LO _{decorr, 2} ... LO _decorr, N].

The probability density function of the random process underlying Φ _rx can be expressed as f (x) = 1 / (σ * sqrt (2 * π) * exp (-0.5 * (((x-μ) / σ) ² ), where μ is the expected value and σ represents the standard deviation The multiplication of the complex baseband signals IQ ₁ , ..., IQ _N with the complex R _decorr results Thus, one for the correct beam shaping due to Φ _inv = -Φ, where Φ = [φ ₁ φ ₂ φ ₃ ... φ _N ], the direction-dependent phase shifts φ _n according to Fig. 2 , and on the other hand to decorrelate the harmonics and thus to obtain an SFDR gain over all points of view Θ due to the inverse relationship of the specifically introduced phase shifts Φ _rx = -φ _r .

Φ

richtungsabhängig auftretenden Phasenverschiebungen, analogseitig, wobei Φ=[ϕ₁ ϕ₂ ϕ₃ ...ϕ_N] mit ϕ_n=n*2*n*π*f_c*sin(Θ) im n-ten Kanal.

Φ _inv

inverse Phasenverschiebungen für korrekte Strahlformung, digitalseitig, wobei Φ _inv=[ϕ_inv,2 ϕ_inv,3 ... ϕ_inv,N]=-Φ.

Φ _r

normalverteilte Phasenverschiebungen, gezielt analogseitig appliziert in der ersten Dekorrelationseinheit 8 wobei Φ _r =[ϕ_r,1 ϕ_r,2 ϕ_r,3 ... ϕ_r,N] mit entnommenen Einzelwerten aus f(x)=1/σ*sqrt(2*π)*exp(-0.5*(((x-µ)/σ)²).

Φ _rx

normalverteilte Phasenverschiebungen, gezielt digitalseitig appliziert in der zweiten Dekorrelationseinheit 9, wobei Φ _rx =[ϕ_rx,1 ϕ_rx,2 ϕ_rx,3 ...ϕ_rx,N]=-Φ _r.

Φ _dekorr

effektiv angewandte Phasenverschiebungen wobei Φ _dekorr=Φ _inv+Φ _rx.

R

Zeigervektor exp(jΦ _inv), digitalseitig appliziert, sorgt für korrekte Strahlformung.

R _dekorr

Zeigervektor exp(jΦ _dekorr), digitalseitig appliziert, sorgt für korrekte Strahlformung und Dekorrelation von Harmonischen.

LO

Signalvektor für analogseitige Mischersignale.

LO _dekorr

Signalvektor für analogseitige Mischersignale mit Phasenterm exp(jΦ_r).

An exemplary vector assignment for Φ _inv and Φ _rx for a viewing angle in radians of Θ = 30 °, d = λ / 2 and N = 100 is in Fig. 4 shown. The hitherto used vector and signal designations are summarized and explained again for clarity.

Φ

direction-dependent occurring phase shifts, analog side, where Φ = [φ ₁ φ ₂ φ ₃ ... φ _N ] with φ _n = n * 2 * n * π * f _c * sin (Θ) in the n-th channel.

Φ _inv

inverse phase shifts for correct beam shaping, digitally, where φ _inv = [φ _{inv, 2} φ _{inv, 3} ... φ _{inv, N} ] = - φ .

Φ _r

normally distributed phase shifts, specifically applied on the analog side in the first decorrelation unit 8 where Φ _r = [φ _{r, 1} φ _{r, 2} φ _{r, 3} ... φ _{r, N} ] with extracted individual values from f (x) = 1 / σ * sqrt (2 * π) * exp (-0.5 * (((x-μ) / σ) ² ).

Φ _rx

normally distributed phase shifts, specifically applied digitally in the second decorrelation unit 9, where Φ _rx = [φ _{rx, 1} φ _{rx, 2} φ _{rx, 3} ... φ _{rx, N} ] = - φ _r .

Φ _decorr

effectively applied phase shifts where Φ _decorr = Φ _inv + Φ _rx .

R

Pointer vector exp (j Φ _inv ), applied on the digital side, ensures correct beam shaping.

R _decorr

Pointer vector exp (j Φ _decorr ), applied on the digital side, ensures correct beam shaping and decorrelation of harmonics.

LO

Signal vector for analog-side mixer signals.

LO _decorr

Signal vector for analog mixer signals with phase term exp (jΦ _r ).

The phased array antenna according to the invention comprises a plurality of receiving elements E ₁ , ..., E _N , N local oscillators, which may be connected to a base oscillator, for example, for generating the oscillator signals, mixers for mixing the oscillator signals LO ₁ ,. LO _N with correspondingly received by the receiving elements E ₁ , ..., E _N received signals, analog-to-digital converter circuits and a signal processor, each receiving element E ₁ , ..., E _N a mixer LO ₁ , ..., LO _{N is} assigned. The inventive phased array antenna is characterized in an embodiment in that the oscillator LO ₁ , ..., LO _N with each mixer LO ₁ , ..., LO _{N is} connected via signal lines, each signal line a targeted additive length deviation whose length is itself normally distributed.

A phased array antenna according to the invention can thus be constructed such that either each signal line is assigned a specific additive length deviation whose length is itself normally distributed, or each oscillator receives a targeted additive phase shift whose value is also normally distributed.

The lengths of the individual signal lines can be derived from a normal distribution of a phase range from -π to + π for a given carrier frequency of the received signal.

The relationship between the lengths of the signal lines and the phase shifts generated is given by I = Φ _decorr / (2 * π) * λ. In this case, λ = c ₀ / (f _c * r), where f _{c represents} the carrier frequency and n the refractive index of the medium. Using the example of f _c = 5 GHz, n = 1 and λ = 6cm, a line length deviation of +/- 3 cm corresponds to the required phase interval of +/- π.

Fig. 5 shows a schematic representation of an exemplary front end of a phased array antenna with 4 receiving elements E1, E2, E3, E4. Each receiving element E1, E2, E3, E4 is in each case assigned a channel K1, K2, K3, K4. In each channel K1, K2, K3, K4 there is a respective mixer M1, M2, M3, M4, which is connected to a common oscillator OSZ. This oscillator OSZ is connected to the individual mixers M1, M2, M3, M4 via individual signal lines L1, L2, L3, L4. The additive length deviations of the individual signal lines L1, L2, L3, L4 in this case correspond to a normal distribution.

Claims (4)

Method for processing received signals in a phased array antenna with a plurality of receiving elements (E ₁ , ..., E _N ) each having an associated receiving path,
wherein in each receive path an analog intermediate frequency signal (U ₁ , ..., U _N ) by mixing the received signal (X ₁ , ..., X _N ) generated in each receive path with an oscillator signal (LO ₁ , ..., LON) and converted into a complex baseband signal (IQ ₁ , ..., IQ _N ) by subsequent digitization,
wherein in each reception path to the complex baseband signal (IQ ₁ , ..., IQ _N ) a phase shift corresponding to the reception direction of the antenna is applied,
characterized in that
when the method is carried out for the first time, each receive element (E ₁ ,..., E _N ) of the phased array antenna has exactly one individual phase value from a phase range from -π to + π normal distribution (φ _r , ₁ , .., φ _{r, N} ) is assigned once and permanently within a first decorrelation unit (8),
that the oscillator signal (LO ₁ , ..., LO _N ) these normal distributed phase values (φ _{r, 1} .., (φ _{r, N} ) are added up and
in a second decorrelation unit (9) in each receive path to the complex baseband signal (IQ ₁ , ..., IQ _N ) a phase shift corresponding to the receive direction of the antenna (φ _{rx, 1} , .., φ _{rx, N} ), in which the normally distributed phase values (φ _{r, 1} , .., φ _{r, N} ) are taken into account. Method according to claim 1,
characterized in that
in the application of the phase shift (φ _{rx, 1} , .., φ _{rx, N} ) to the complex baseband signal (IQ ₁ , ..., IQ _N ) an amplitude weighting is performed. Phased array antenna comprising a plurality of receiving elements, an oscillator (OSZ) for generating an oscillator signal, mixers (M1, ..., M4) for mixing the oscillator signal with received signals correspondingly received by the receiving elements (E1, ..., E4) , Analog-to-digital converter circuits and a signal processor,
wherein each receiving element (E1, ..., E4) is assigned a mixer (M1, ..., M4),
characterized in that
the oscillator (OSZ) is connected to each mixer (M1, ..., M4) via signal lines (L1, ..., L4), each signal line (L1, ..., L4) being assigned a specific additive length deviation, whose length itself is normally distributed. Phased array antenna according to claim 3,
characterized in that
the lengths of the individual signal lines (L1, ..., L4) are derived from a normal distribution of a phase range from -π to + π at a predetermined carrier frequency of the received signal.

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