DE102012009402B3 - Phased array antenna and method for processing received signals in a phased array antenna - Google Patents

Abstract

The invention relates to a method for processing received signals in a phased array antenna having a plurality of receiving elements each having an associated receiving path, wherein upon first implementation of the method each receiving element of the phased array antenna exactly one individual from a phase range of -π to + π normal distribution phase value is assigned once and permanently. The invention further relates to a phased array antenna comprising a plurality of receiving elements, a mixer for mixing the oscillator signal in accordance with the receiving signals received by the receiving elements and an oscillator for generating an oscillator signal, wherein the oscillator is connected to each mixer via signal lines, either each signal line is assigned a specific additive length deviation whose length is itself normally distributed, or each LO receives a targeted additive phase shift whose value is also normally distributed.

Description

The invention relates to a method for processing received signals in a phased array antenna according to the features of patent claim 1 and a phased array antenna according to the features of patent claim 3.

STATE OF THE ART

Out DE 600 25 064 T2 and DE 10 2007 046 566 B4 radars are known using digital beamforming.

In 1 The structure of an N-channel phased array antenna with digital beam shaping is shown schematically. The use of a plurality of receiving channels shown in FIG. 5 achieves a directivity in the surroundings of the radar device. Under ideal conditions, the dynamic range over a single receiver system increases by a factor of 10log ₁₀ (N) dB, where N expresses the number of channels used and n expresses the run index for the nth channel. Using the example of a linear phased array antenna 1 with antenna distance d = λ / 2 (λ: Wavelength associated with the carrier frequency f _c used via the speed of light c, corresponding to λ = c / f _c ), the further processing is described. The high-frequency signals X ₁ ,..., X _N provided by the antenna elements E ₁ ,..., E _N are fed to the analogue receiver stages ARX ₁ ,..., ARX _N. In the complete analogue receiver unit 2 the high frequency signals X ₁ , ..., X _{N are converted} to a lower intermediate frequency U ₁ , ..., U _N. For this purpose, a mixed signal BO from the block is a central base oscillator 6 the analogue receiver unit 2 fed. The internal distribution of the central BO takes place at each analogue receiver stage ARX ₁ ,..., ARX _N. The intermediate-frequency signals U ₁ ,..., U _n generated by the mixing process of X ₁ ,..., X _N with the oscillator signals derived from BO are subsequently sent to the digital receiver unit 3 fed. There, the analog signals with the analog-to-digital converters ADC ₁ , ..., ADC _N are first converted into digital signals. In the downstream digital preprocessing unit PP ₁ ,..., PP _N , the complex baseband signals IQ ₁ ,..., IQ _{N are} generated. A complex baseband signal IQ _n is vectorially composed of a real part Re {IQ _n } and an imaginary part Im {IQ _n }. The processing unit Digital Beamforming 4 is part of a signal processor and uses all signals IQ ₁ , ..., IQ _N provided by the N individual channels to form a number J beams B ₁ , ..., B _J. The number of beams is typically: J <N. These beams provide directional access to distance and speed information. The concept is explained independently of the selected modulation type.

As is known theoretically for linear phased arrays, an incident signal wave produces in the n-th channel a direction-dependent phase shift φ _n = n * 2 * π * f _c * d / c * sin (Θ), where f _{c is} the carrier frequency, d represents the antenna distance and Θ the angle of incidence. For further illustration becomes 2 , with the receiving channel 7 based on the running index n = 1.

The provided by the local LO ₁ signal LO ₁ is from the central base oscillator BO (reference numeral 6 ) in 1 , derived. The central base oscillator 6 provides each receiving element with a local LO signal of identical frequency. By multiplying LO ₁ and the phase-shifted X ₁ , the intermediate frequency signal U _{1 is} generated. The digitization of the value and time-continuous signal U ₁ in the signal D ₁ takes place in the analog-to-digital converter (ADC ₁ ). The generation of the complex baseband signal Re {IQ ₁ } + j * Im {IQ ₁ } takes place in the preprocessing unit PP1. By definition, ADC ₁ and PP ₁ form a single digital receiver 10 , The following is the processing work Digital Beamforming 4 out 1 explained in more detail. The direction-dependent phase shifts φ ₁ -φ _N remain in the conversion of the signals X ₁ ,..., X _N to IQ ₁ ,..., IQ _N and can be used in the mixer units P ₁ ,..., P _N in FIG each be applied inversely. The multiplication of the complex signals IQ ₁ ~ IQ _N with inverse rotational pointers R _n = exp (-j * n * 2 * π * f _c * d / c * sin (Θ)) produces an inverse phase shift of φ _{inv, n} = - φ _n in each n-th channel and allows a coherent addition (constructive superposition) of all received signals in the subsequent summing unit 12 , The generated output signals B ₁ -B _j are fed to a subsequent processor unit (radar processor).

ergibt sich ein Richtdiagramm nach G(Θ) = 10log₁₀(sin²(N·π·d/λ·sin(Θ – Θ₀))/(N²·sin²(π·d/λ·sin(Θ – Θ₀)))), wobei Θ₀ den tatsächlichen Einfallswinkel der elektromagnetischen Welle und Θ die eingestellte Vorzugsrichtung der digitalen Strahlformung.The totality of all applied inverse phase shifts can thus be in a vector Φ _inv = [φ _{inv, 2} φ _{inv, 3} φ _{inv, 4} φ _{inv, 5} ... φ _{inv, N} ] and thus a pointer vector R = A _taper · exp (jΦ _inv ), which is applied by multiplication to the mixer units P ₁ ~ P _n . If no amplitude weighting is provided

The result is a directional diagram according to G (Θ) = ₁₀ log ₁₀ (sin ² (N × π × d / λ × sin (Θ-Θ ₀ )) / (N ² × sin ² (π × d / λ × sin (Θ - Θ ₀ )))), where Θ ₀ the actual angle of incidence of the electromagnetic wave and Θ the set preferred direction of digital beam forming.

Conventional methods have a drawback with respect to the useful noiseless dynamic range. When using beamforming, distortion products (HD ₂ , HD ₃ ... HD _i = Harmonic Distortion, interfering signals generated in nonlinear systems at multiples i of the signal frequency) experience an inverse phase shift corresponding to the useful signal. The the Distorted products associated phases have a different vector Φ _inv factor Ψ, z. Eg Ψ = 2 for HD2. Depending on the viewing angle Θ, this leads to a partially destructive or constructive superimposition of these distortion products.

The disadvantage of signal processing in accordance with the prior art is therefore that the addition results in a constructive interference of the input signal, but in the case of the distortion products, apart from a partially destructive interference, this also leads to a constructive interference at the viewing angle Θ = 0 ° and larger angles the zeros in the function P (Θ) = ₁₀ log ₁₀ (sin ² (Ψ × N × π × d / λ × sin (Θ-Θ ₀ )) / (N ² × sin ² (Ψ × π × d / λ × sin (Θ - Θ ₀ )))). This inevitably worsens the spur-free dynamic range (SFDR), which in this case represents the ratio of the power magnitude of the largest harmonic to the power level of the received signal (fundamental). The dynamic range increase 10log ₁₀ (N) dB expected by the digital beamforming can thus not be guaranteed, since distortion products are not decorrelated from channel to channel at all viewing angles Θ and thus are added up in the same way as the useful signal.

DESCRIPTION OF THE INVENTION

The object of the invention is to provide a method in which occurring in the receiver harmonics are suppressed by the selected signal processing concept and the trouble-free dynamic range is improved over all angles Θ. Another object is to provide a corresponding antenna.

The objects are achieved by the method according to the features of the valid patent claim 1 and the device according to claim 3. Advantageous embodiments of the invention are the subject of dependent claims.

The invention and further advantageous embodiments of the invention will be explained in more detail with reference to the drawing. Show it:

1 the construction of an N-channel phased array antenna with digital beam shaping,

2 the structure of an N-channel phased array antenna with digital beam shaping and direction-dependent phase shifts,

3 FIG. 2 schematically the structure according to the invention of an N-channel phased array antenna with digital beam shaping including the decorrelation units, FIG.

4 exemplary phase assignment according to the invention in the receiving part

5 a schematic representation of an exemplary front end, the mixer and the oscillator of a phased array antenna according to 1 ,

3 schematically shows the construction according to the invention of an N-channel phased array antenna with digital beam shaping including the decorrelation units 8th . 9 , According to the invention, receive signals X ₁ ,..., X _{N are processed} in a phased array antenna having a plurality of receive elements E ₁ ,..., E _N , each having an associated receive path, an analog intermediate frequency signal U ₁ in each receive path , ..., U _N produced by mixing the reception signal X _1, ..., X _N with an oscillator signal LO _1, ... LO _N and then by digitization and possibly digital blend into a complex base band signal IQ ₁ , ..., IQ _N , wherein in each receive path to the complex baseband signal IQ ₁ , ..., IQ _N one of the receive direction of the antenna corresponding phase shift is applied. The invention is characterized in that, when the method is carried out for the first time, each receiving element of the phased array antenna has exactly one individual phase value φ _{r, 1} ,..., Φ _{r, N} within the first normal phase distribution from -π to + π decorrelation unit 8th is assigned once and permanently, that the oscillator signal these normal distributed phase values φ _{r, 1} , ..., φ _{r, N are} added up and that within the second decorrelation unit 9 in each receive path to the complex baseband signal I _Q1 , ..., IQ _N one of the receive direction of the antenna corresponding phase shift φ _{rx, 1} , ..., φ _{rx, N} in which the normally distributed phase values φ _{r, 1} , .. ., φ _{r, N are} considered inverse.

The first decorrelation unit 8th and the second decorrelation unit 9 differ in that the first decorrelation unit 8th a selectively added channel-dependent phase shift applied in the analog part of the receiver, whereas the second decorrelation unit 9 the added channel-dependent phase shift in the digital part of the receiver is inversely applied and thus reversed. The introduced blocks RX ₁ , ..., RX ₂ each denote the union of ADC and PP in each channel (see also FIG 1 ).

When applying the phase shift φ _{rx, 1} , ..., φ _{rx, N} to the complex baseband signal IQ ₁ , ..., IQ _N , an amplitude weighting can additionally be performed, the application of a Weighting function does not affect the subject of this invention.

The invention underlying digitally applied phase vector in the second decorrelation unit 9 can thus be expressed as Φ _decorr = Φ _inv + Φ _rx , where Φ _rx = [φ _{rx, 1} φ _{rx, 2} φ _{rx, 3} ... φ _{rx, N} ] of the additive normal-distributed random process the first decorrelation unit 8th Analogously applied set of phase values representing which lie between +/- π. The analog-applied phase vector in the first decorrelation unit 8th corresponds to the phase vector in the second decorrelation unit 9 with opposite sign according to Φ _r = Φ _rx . Furthermore, Φ _{inv is to be assigned to} the actual beam shaping and into those already in 2 , described complex rotary hands R ₁ ~ R _N included. Φ _rx contains the additionally introduced phase values. The phase shift applied in vector form on the digital side can thus be expressed as a multiplication by the signal _vector R _dekorr = exp (j (φ _inv + φ _rx )) = exp (jφ _dekorr ) with the individual _pointers [R _{dekorr, 1} R _{dekorr, 2} ... R _{dekorr, N} ]. On the analog side, phase shifts are likewise applied to the mixer signals LO ₁ -LO _N via the vector Φ _r = [φ _{r, 1} φ _{r, 2} φ _{r, 3} ... Φ _{r, N} ]. The mixer signals with introduced phase shifts φ _{r, 1} ,..., Φ _{r, N} are combined in the signal _vector LO _dekorr = [LO _{decorr, 1} LO _{decorr, 2} ... LO _{decorr, N} ].

The probability density function of the random process underlying Φ _rx can be expressed as f (x) = 1 / (σ · sqrt (2 · π) · exp (-0.5 · (((x - μ) / σ) ² ), where μ is the expected value and σ the standard deviation represented. the multiplication of the complex baseband signals IQ _1, ..., IQ _N with the complex R _dekorr thus leads to a the correct beamforming due φ _inv = -Φ wherein φ = [φ ₁ φ ₂ φ ₃ ... φ _N ] the direction-dependent phase shifts φ _n according to 2 , and, secondly, decorrelation of the harmonics and thus an SFDR gain over all angles of view Θ due to the inverse relationship of the deliberately introduced phase shifts Φ _rx = Φ _r .

Φ

richtungsabhängig auftretenden Phasenverschiebungen, analogseitig, wobei Φ =[φ₁ φ₂ φ₃ ... φ_N] mit φ_n = n·2·π·f_c·d/c·sin(Θ) im n-ten Kanal.

Φ_inv

inverse Phasenverschiebungen für korrekte Strahlformung, digitalseitig, wobei Φinv = [φ_inv,2 φ_inv,3 ... φ_inv,N] = –Φ.

Φ_r

normalverteilte Phasenverschiebungen, gezielt analogseitig appliziert in der ersten Dekorrelationseinheit 8 wobei Φ_r = [φ_r,1 φ_r,2 φ_r,3 ... φ_r,N] mit entnommenen Einzelwerten aus f(x) = 1/(σ·sqrt(2·π)·exp(–0.5·(((x – μ)/σ)²).

Φ_rx

normalverteilte Phasenverschiebungen, gezielt digitalseitig appliziert in der zweiten Dekorrelationseinheit 9, wobei Φ_rx = [φ_rx,1 φ_rx,2 φ_rx,3 ... φ_rx,N] = –Φ_r.

Φ_dekorr

effektiv angewandte Phasenverschiebungen wobei Φ_dekorr = Φ_inv + Φ_rx.

R

Zeigervektor exp(jΦ_inv), digitalseitig appliziert, sorgt für korrekte Strahlformung.

R_dekorr

Zeigervektor exp(jΦ_dekorr), digitalseitig appliziert, sorgt für korrekte Strahlformung und Dekorrelation von Harmonischen.

LO

Signalvektor für analogseitige Mischersignale.

LO_dekorr

Signalvektor für analogseitige Mischersignale mit Phasenterm exp(jΦ_r).

An exemplary vector assignment for Φ _inv and Φ _rx for a viewing angle in radians of Θ = 30 °, d = λ / 2 and N = 100 is in 4 shown. The hitherto used vector and signal designations are summarized and explained again for clarity.

Φ

direction-dependent occurring phase shifts, analog side, where Φ = [φ ₁ φ ₂ φ ₃ ... φ _N ] with φ _n = n · 2 · π · f _c · d / c · sin (Θ) in the n-th channel.

Φ _inv

inverse phase shifts for correct beam shaping, digitally, where Φinv = [φ _{inv, 2} φ _{inv, 3} ... φ _{inv, N} ] = -Φ.

Φ _r

normally distributed phase shifts, specifically applied on the analog side in the first decorrelation unit 8th where Φ _r = [φ _{r, 1} φ _{r, 2} φ _{r, 3} ... φ _{r, N} ] with extracted individual values from f (x) = 1 / (σ · sqrt (2 · π) · exp (-0.5 · (((X - μ) / σ) ² ).

Φ _rx

normally distributed phase shifts, specifically applied digitally in the second decorrelation unit 9 where φ _rx = [φ _{rx, 1} φ _{rx, 2} φ _{rx, 3} ... φ _{rx, N} ] = -φ _r .

Φ _decorr

effectively applied phase shifts where Φ _decorr = Φ _inv + Φ _rx .

R

Pointer vector exp (jΦ _inv ), applied on the digital side, ensures correct beam shaping.

R _decorr

Pointer vector exp (jΦ _decorr ), applied on the digital side, ensures correct beam shaping and decorrelation of harmonics.

LO

Signal vector for analog-side mixer signals.

LO _decorr

Signal vector for analog mixer signals with phase term exp (jΦ _r ).

The inventive phased array antenna comprises a plurality of receiving elements E1, ..., EN, N local oscillators, which z. B. can be connected to a basic oscillator, for generating the oscillator signals, mixers for mixing the oscillator signals LO ₁ , ..., LO _N with correspondingly received by the receiving elements E ₁ , ..., E _N received signals, analog-to-digital converter circuits and a signal processor, wherein each receiving element E ₁ , ..., E _N, a mixer LO ₁ , ..., LO _{N is} assigned. The inventive phased array antenna is characterized in an embodiment in that the oscillator LO ₁ , ..., LO _N with each mixer LO ₁ , ..., LO _{N is} connected via signal lines, each signal line a targeted additive length deviation whose length is itself normally distributed.

A phased array antenna according to the invention can thus be constructed such that either each signal line is assigned a specific additive length deviation whose length is itself normally distributed, or each oscillator receives a targeted additive phase shift whose value is also normally distributed.

The lengths of the individual signal lines can be derived from a normal distribution of a phase range from -π to + π for a given carrier frequency of the received signal.

The relationship between the lengths of the signal lines and the phase shifts generated is given by I = φ / (2 · π) · λ. In this case, λ = c ₀ / (f _c * r), where f _{c represents} the carrier frequency and n the refractive index of the medium. Using the example of f _c = 5 GHz, n = 1 and λ = 6 cm, a line length deviation of +/- 3 cm corresponds to the required phase interval of +/- π.

5 shows a schematic representation of an exemplary front end of a phased array antenna with 4 receiving elements E1, E2, E3, E4. Each receiving element E1, E2, E3, E4 is in each case assigned a channel K1, K2, K3, K4. In each channel K1, K2, K3, K4 there is a respective mixer M1, M2, M3, M4, which is connected to a common oscillator OSZ. This oscillator OSZ is connected to the individual mixers M1, M2, M3, M4 via individual signal lines L1, L2, L3, L4. The additive length deviations of the individual signal lines L1, L2, L3, L4 in this case correspond to a normal distribution.

Claims (4)

A method for processing received signals in a phased array antenna having a plurality of receiving elements (E ₁ , ..., E _N ) each having an associated receive path, wherein in each receive path an analog intermediate frequency signal (U ₁ , ..., U _N ) by mixing the received signal (X ₁ , ..., X _N ) in each receive path with an oscillator signal (LO ₁ , ..., LO _N ) and by subsequent digitization into a complex baseband signal (IQ ₁ , .. ., IQ _N ), wherein in each receiving path to the complex baseband signal (IQ ₁ , ..., IQ _N ) one of the receiving direction of the antenna corresponding phase shift is applied, characterized in that when first performing the method each receiving element (E ₁ , ..., E _N ) of the phased array antenna exactly one individual phase value from a phase range from -π to + π normalverteiler (φ _{r, 1} , ..., φ _{r, N} ) within a first decorrelation unit ( 8th ) is assigned to the oscillator signal (LO ₁ , ..., LO _N ) these normally distributed phase values (φ _{r, 1} , ..., φ _{r, N} ) and that within a second decorrelation unit ( 9 ) in each receive path to the complex baseband signal (IQ ₁ , ..., IQ _N ) one of the receive direction of the antenna corresponding phase shift (φ _{rx, 1} , ..., φ _{rx, N} ), in which the normally distributed phase values ( φ _{r, 1} , ..., φ _{r, N} ) is applied. A method according to claim 1, characterized in that in the application of the phase shift (φ _{rx, 1} , ..., φ _{rx, N} ) to the complex baseband signal (IQ ₁ , ..., IQ _N ) an amplitude weighting is performed , Phased array antenna comprising a plurality of receiving elements, an oscillator (OSZ) for generating an oscillator signal, mixers (M1, ..., M4) for mixing the oscillator signal with received signals correspondingly received by the receiving elements (E1, ..., E4) , Analog-to-digital converter circuits and a signal processor, wherein each receiving element (E1, ..., E4) is assigned a mixer (M1, ..., M4), characterized in that the oscillator (OSZ) is connected to each mixer (M1, ..., M4) via signal lines (L1, ..., L4), each signal line (L1, ..., L4) being assigned a specific additive length deviation, whose length itself is normally distributed. Phased array antenna according to claim 3, characterized in that the lengths of the individual signal lines (L1, ..., L4) are derived from a normal distribution of a phase range from -π to + π at a predetermined carrier frequency of the received signal.

Images