EP1836766A1

EP1836766A1 - Method and device for amplifying an amplitude- and phase-modulated electric signal

Info

Publication number: EP1836766A1
Application number: EP06707664A
Authority: EP
Inventors: Andreas Langer
Original assignee: Siemens AG
Current assignee: Siemens AG
Priority date: 2005-01-12
Filing date: 2006-01-04
Publication date: 2007-09-26
Also published as: WO2006074974A1; DE102005001496B4; DE102005001496A1; US20080261541A1; CN1969454A

Abstract

The aim of the invention is to suppress interference caused by a mismatch of the power amplifier (5, 6) of a polar-loop transmitter. To achieve this, the crest factor of the output signal from the power amplifier (5, 6) is measured during operation to detect the state of the mismatch and to identify the modification of the transfer characteristic curve of the power amplifier. The crest factors that have been determined are compared with a target value and if the crest factor deviates from the target value, the bandwidth or the amplification of the amplitude closed loop (2, 4) is adapted accordingly. The transmitter can thus regulate the non-linear distortions that occur as a result of the mismatch. Said measures improve the linear behaviour or degree of efficiency of the power amplifier (5, 6).

Description

description

Method and device for amplifying an amplitude and phase modulated electrical signal

In order to increase the data rate to be transmitted via a mobile radio connection, modulation methods are now being used which influence both the amplitude and the phase position of the signal to be transmitted. Examples include variants of the GSM standard (global standard for mobile communication), in particular EDGE (Enhanced data rate for GSM evo ^¬ lution). In order to avoid errors in the transmission of a signal which is both amplitude- and phase-modulated, the signal-generating transmitting unit and in particular its power amplifier should have as linear a behavior as possible. Although highly linear power supplies that meet the requirements are available in principle, they are more complex and therefore more expensive to manufacture.

The linearity of a transmitter used in modern mobile radio systems can be improved by applying the so-called polar-loop concept. In one based on this concept polar loop transmitter, the amplitude is separated from the phase by means of two control loops (Amplitudenregel- circular / AM loop and phase-locked loop / PM loop) the Trä ^¬ gersignal modulated (see for example WO 03/005564 o- WO 02/47249). The splitting of the amplitude- and phase-modulated signal into an exclusively amplitude-modulated component and an exclusively phase-modulated component increases the bandwidth of the two individual signals. The bandwidth of the control circuits should therefore be as large as possible in order to minimize the linear distortion of the two individual signals due to the low-pass effect of the respective control loop. Although a wide range of the AM loop reduces linear distortion, but increased gleichzei ^¬ tig the noise in the receive band (RX band) of an EDGE transmitter and receiver unit. On the other hand, the bandwidth of the AM loop changes with a mismatch of the power amplifier, while the bandwidth of the PM loop usually remains constant despite mismatching. In addition, the saturation power of the power amplifier decreases. This can lead to the amplifier being ^controlled in the nonlinear region of the characteristic and intermodulation products occurring. The da ^¬ with concomitant distortion of the output signal can cause the system requirements in terms spekt- tral purity are no longer satisfies the signal.

The invention is therefore based on the object, in particular to suppress the parasitic effects generated by a mismatch of the power amplifier of a polar loop transmitter largely. This object is according to the invention ren or by a procedural ^¬ having the features of claim 1. solved by a device having the features of claim 8. The dependent claims relate to advantageous developments and refinements of the method or. the device.

With the aid of the invention, the requirements with regard to the necessary back-off of the power amplifier of a polar loop transmitter can be considerably reduced. This is accompanied by the improvement of the efficiency of the power amplifier, which correspondingly extends the maximum operating time of the mobile telephone equipped with ^{such a} transmitter. Due to the lower power loss of the power amplifier, the sen temperature during operation also increases significantly less. In addition, the adaptive adjustment of the bandwidth of the AM control loop ensures op ^¬ timales noise behavior (as a consequence of the low bandwidth of the AM control loop) and with a possible mismatch li ^¬ neares behavior of the entire transmitter.

The invention will be explained below with reference to drawings. Show it : Fig. l: the crest factor, the linear output power and the output power of the amplifier unit of a polar loop transmitter as a function of the input power;

FIG. 2 shows the schematic structure of a polar-loop transmitter according to the invention; FIG.

As already mentioned, the saturation power of the amplifier unit of a polar loop transmitter decreases as a result of Fehlanpas solution. This reduces the back-off, which leads to non-linear distortions when the envelope is driven into the non-linear region of the amplifier characteristic. Since the state of the mismatch is not detected in known transmitters and, moreover, the change of the characteristic as a result of a mismatch is unknown, it is proposed to ^carry out a mismatch or a mismatch. to detect the resulting compression of the amplifier characteristic curve by evaluating the crest factor called the "peak-to-average" ratio of the demodulated RF output signal and to compensate for the parasitic effects associated therewith.

Mathematically, the output signal y (t) of the amplifier unit is a non-linear mapping of the Verstär ^¬ kereinheit supplied input signal x (t). Due to the non-linear transformation, the statistical properties of the input signal x (t) and thus also the crest factor CF are changed. The transfer function of the amplifier unit is described by the following, simple polynomial representation, wherein the quantities a and. c constants and s (t) denote the complex envelope:

y (t) = ax (t) -cx ³ (t)

with x (t) = Re {s (t) e ^J ""}, (Re {}: = operator for real part); The complex envelope s (t) depends in this case according to the rela ^¬ hung

s ² (t) = I ² (t) + Q ² (t)

from the two baseband signals I (t) and Q (t). The crest factor CF _{x of} the input signal x (t) is defined as

Max [x (t)] tr _v =

with σ _x ² (t) = E (x ² (t)) (E {}: = operator of the expected value).

Calculating the crest factor CF _{y of} the output signal y (t) as a result of the non-linear transformation, we obtain:

With

_{CF =} Max [y (t)]

Under the above conditions (approximate representation of the transfer function of the amplifier unit by a simple polynomial and not, as required, by a Volterra series!), The crest factor depends only on the ratio c / a, which in turn is a measure of the ldB Compression point of the amplifier characteristic is. In Fig. 1 shows the crest factor CF _y and the output power of the amplifier unit as a function of the input power for the parameter values a = 3, c = 0, 3 and CF _x = 3.5 dB.

As can be seen with reference to FIG. 1, the smaller the deviation in the output power from a linear reference output power measured in dBm dB measured crest factor CF _y . I st a limit SW, z. B. SW = I .8dB, given, the value of the crest factor CF _{y should} then not be less than the value of the predetermined threshold SW. The value of the crest factor CF _y can thus serve as a measure of the compression of the amplifier characteristic, wherein the deviation "not compressed characteristic" of a state representing the reference value of the absolu ^¬ th output power is independent. The crest factor CF _y is rather determined by the "curvature" of the characteristic curve at the instantaneous average output power of the amplifier unit.

Fig. 2 shows the structure of a polar loop transmitter according to the invention. In this transmitter, the amplified, both amplitude and phase modulated input signal U _{mod is} split into an amplitude-modulated component and a phase-modulated component and further processed these signal components in ge ^¬ separate control loops. Mathematically, the amplitude-modulated component in this case corresponds to the magnitude of the complex envelope (see above), the phase modulated component, however, the phase of the complex Einhül ^¬ loins. Separate control ^¬ exist circles for both signal components, the amplitude control circuit (AM) loop from egg ^¬ nem amplitude comparator 2, an intermediate amplifier 4 and a battery voltage modulator is (LDO 6) and ^¬ NEN the phase locked loop (PM loop) comprising a phase comparator 1 and ei the input signal x (t) of the power amplifier 5 erzeu ^¬ ing, voltage controlled Os cillator 3 includes. On the input side, the phase comparator 1 and the amplitude comparator 2 are respectively connected to the input signal U _mOd

Reference / setpoint as well as with the means of a coupler from ^¬ seized, optionally mixed down to an intermediate or base ^¬ band frequency down (mixer 8) and then amplified (repeater variable Gain Amplifier 7) output signal U _{out of} the power amplifier 5 as Ver ^¬ equivalent value acted upon , The output signal of the phase comparator 1 regulates the phase-modulated component of the output signal. Signals U _ou t by means of the voltage controlled oscillator 3 to the set by the input signal U _mOd setpoint.

In the illustrated polar-loop transmitter, the amplitude modulation is generated by varying the supply voltage U _{D of} the power amplifier 5. The amplitude _comparator 2 influenced via the controllable battery voltage _modulator 6 (control voltage U _LD0 ) the supply voltage U ₀ = f (U _LD0 , U _ßatt ) ^of power amplifier 5 and thus the envelope of the antenna 11 fed output signal U _out such that s the amplitude of the envelope of the output signal U _out a FEH ^¬ lerfreies image of the amplitude of the voltage applied to one of the two a ^¬ gears of the magnitude 2 input signal U _m o _d is. In this case, U _{D is} generated with the aid of the voltage modulator 6 from a battery voltage U _Batt .

The linear range of the power amplifier 5 is characterized by a linear relationship between the output signal U _out and the control voltage U _LDO . This linear combination ^¬ slope is given as long as the control voltage U _D sufficiently far tt of the battery voltage U _Ba is removed. Is the control voltage U ₀ approaching the battery voltage U _Ba tt _? Thus, the transfer characteristic (U _ou t as a function of U _LD O) of the power amplifier 5 due to saturation effects in the LDO 6 is compressed. As a result, the steepness of the Trans ferkennli ^¬ never drops. This results in the AM loop to a reduction of Re ^¬ gelbandbreite. The reduction of the bandwidth leads in addition to the deteriorated leadership behavior of the scheme to further effects that are typical of a polar loop transmitter and the spectrum of the modulated transmission signal significantly in ^¬ flow sen.

The splitting of the phase- and amplitude-modulated input signal U _mod into an amplitude-modulated and a phase-modulated component increases the bandwidth of the individual signal components. The bandwidth of the control circuits (AM loop and PM loop) must therefore be chosen so that the linear distortion of the two sub-signals due to the low-pass effect of the respective control loop are minimal. If, for example, the amplitude spectrum is too heavily filtered due to a too low bandwidth of the AM control loop, this results in a broadened spectrum of the output signal Uout. If the phase or amplitude spectrum is linearly distorted (ie suppression of the higher-frequency components of the amplitude spectrum), In the overall spectrum, the extinction of the higher-frequency signal components is worsened, which broadens the overall spectrum. From this the demand can be derived to choose the bandwidth of the control loops as large as possible. This results, as already explained above, al ^¬ larger companies at an increased noise level in the receive band (RX band).

The range of control circuits should, regardless of the ge ^¬ desired output power and other environmental conditions remain constant as possible. With the repeater 4 (Variable Gain Amplifier) in the forward branch, the open-

Loop gain of the transmitter. This repeater 4 has in sufficiently large overall gain of the AM loop output power does not affect the out ^¬. The second repeater 7 (variable gain amplifier) in the feedback path is also part of the cumulative gain ^¬ with, but directly affects the output power ^¬ U out. The greater the gain of the feedback (Mi ^¬ shear 8, amplifier I) ₁ the smaller the average output ^¬ performance.

As already explained above, the trans ferkennlinie kompri ^¬ mized when the output signal U _{LDO of} the repeater 4, the battery voltage U _Ba tt approaches. If this occurs, the steepness of the transfer characteristic and thus also the open loop gain decreases, which in turn means a reduction in the bandwidth of the AM control loop. Since the GSM system requirements the saturation performance of an EDGE transmitter and thus also the Power amplifier 5, the latter is operated with sufficient ^¬ the "back-off" This back-off operation guarantees a sufficiently linear behavior of the amplifier to the maximum power possible.

Despite the aforementioned back-off operation but unwanted ef ^¬ fect may occur in a mismatch of the transmitter amplifier. A Fehlanpas solution can be caused by impedance changes, for example, by changing the distance between see the antenna of the mobile phone and the head of Benut ^¬ zers, caused. Such mismatch causes the slope of the transfer characteristic in line ^¬ aren area and therefore the bandwidth of the AM control loops ses that change. In addition, the saturation power of the transmitter amplifier drops. By the lowering of the saturation power, the back-off and thus reduce the distance ferkennlinie for non ^¬ linear region of Trans. If the trans ferkennlinie is controlled as a result of the AM modulation in the non-linear range, intermodulation products occur. DIE se intermodulation products are regulated by the AM loop out ^¬, the bandwidth of the AM control provided sufficiently large. Since the latter can not always be assumed, it is proposed that the crest factor CF _{y of} the amplifier output signal Uout or. y (t) as a measure of the compression of the amplifier input signal x (t) and to adjust the band width ^{¬ of} the polar loop transmitter, according to the deviation of the gemes senen crest factor CF _y from a comparison value to adjust.

The crest factor CF _y can, for example, by incoherent demodulation of the amplifier output signal U _ou t or. y (t) can be measured by means of the device described in WO 03/096548 A2. As shown in FIG. 2, this device consists of a device with the output signal of the coupler, the instantaneous power resp. the average power (RMS power) measuring envelope demodulator (HDK) 14, a level shifter (LS) 13, an analog-to-digital converter (ADC) 12 and one of the calculation of the crest factor CF _y digital signal processing device 9th

If the measured crest factor CF _y falls below a threshold value which defines too great a compression, then the

Gain in the forward branch of the AM control loop increased. The necessary comparison of the crest factor CF _y with the smoldering ^¬ lenwert in which the signal processing device 9 downstream means 10 runs, whose first output is connected to the control input of the amplifier 4 and a second output connected to the control input of the repeater. 7 By increasing the forward gain, only the bandwidth of the AM control loop changes, but not the output power of the power amplifier 5. If the increase in gain (and thus the band width ^¬ ) selected according to the deviation of the measured crest crest factor CF _y from the threshold , the AM closed loop ^¬ can compensate nonlinear distortion of the amplifier unit. 5

Zoom in case of excessive compression of the characteristic mechanism s is the bandwidth of the AM loop very strong to take the higher order intermodulation products to it ^¬. However, this can affect the stability of the system. In order to avoid such stability problems, as already proposed, the bandwidth of the AM control loop is initially increased by triggering the repeater 4. Is the max. Increase AM bandwidth resp. the pre ^¬ Windwärts gain is not sufficient to maintain the then gemes Senen crest factors tor CF _y below the threshold, it is recommended that the gain in the reverse branch of the transmitter by dently ^¬ tion of the repeater 7 to increase so that the baking ^¬ off, with unchanged output power, increases and nonlinear distortions are reduced. At the same time must naturally lent the gain in the forward path will be adjusted accordingly, so that the bandwidth of the AM closed loop does not know ^¬ ter increases. Since the increase of the bandwidth is not uncritical for reasons of stability, it is further alternatively / additionally proposed to influence the characteristic of the AM control loop by changes in a loop filter. Such a loop filter can be integrated or installed in the phase comparator 1. Thus, for example, Cha ^¬ rakteristik be changed by switching on or off of individual filter elements. The selection of the loop filter depends on the crest factor. In this way, for example, the phase margin can be increased with high bandwidths (and high forward amplification).

Claims

claims

1. A method for amplifying an amplitude and phase modulated electrical signal in a polar loop transmitter, wherein the phase and the amplitude of the electrical signal (U _m o _d ) separated by means of a phase locked loop (1, 3) and an amplitude control loop ( 2, 4, 7, 8) are each set or readjusted by comparison with a reference value, wherein the input side of the electrical signal (U _mOd ) acted upon phase-locked loop (1, 3) an input signal of an amplifier unit (5, 6) and the just ^¬ if the input side beauf with the electrical signal (U _mod) ^¬ as estimated amplitude control circuit (2, 4, 7, 8) a control voltage (U _LD0) of the amplifier unit (5, 6) generated dadur ch labeled in zei seframe that at least a Crest factor is determined and compared with a setpoint ^¬ value, and that at a deviation of the crest factor from the desired value, the gain of the amplitude control loop in the forward or reverse zw eig (2, 4, 7, 8) is changed.

2. Method according to claim 1, characterized in that it is changed to change the bandwidth of the amplitude control loop (2, 4, 7, 8) of the forward feedforward.

3. Method according to claim 1 or 2, characterized in that the amplification of the amplitude control loop in the backward branch (7, 8) is changed.

4. The method according to any one of claims 1 to 3, dadur ch gekenn zei chnet, since ss the control voltage (U _LDO ) and the amplitude of the envelope are sampled time synchronous.

5. The method according to any one of claims 1 to 4, dadur ch labeled in zei seframe as ss the bandwidth of the amplitude control circuit (2, 4, 7, 8) by Su ^¬ the gain alteration of a first amplifier (4) into a control signal (U _LD0 ) generating forward _branch (2, 4) of the amplitude _{control loop is} changed.

6. Method according to one of claims 2 to 5, characterized in that the gain is amplified by driving a second amplifier (7) in a reverse phase branch (7, 8) of the amplitude control loop (7, 8) generating the reference values of the phase and the amplitude. 4, 7, 8) is changed.

7. An apparatus for amplifying an amplitude and phase modulated electrical signal, with an output side connected to an amplifier unit (5, 6) Phasenre ^¬ gelkreis (1, 3) and an output side with a Steuerein ^¬ gear of the amplifier unit (5, 6) connected amplitude rule - circle (2, 4, 7, 8), wherein first inputs of the phase locked loop (1, 3) and the amplitude control loop (2, 4, 7, 8) each with the electrical signal (U _mOd ) and second inputs of the phase ^¬ loop (1, 3) and the amplitude control circuit (2, 4, 7, 8) each with a decoupled output signal (U _out ) of the amplifier unit (5, 6) are charged dadur ch gekenn zei chnet, as ss the device first means (9 ) for calculating the crest factor and / or second means (10) for comparing the crest factor with a desired value and for changing the bandwidth or the gain of the amplitude control loop (2, 4, 7, 8) in dependence from the deviation of the crest factor from the nominal ^¬ value has.

8. Apparatus according to claim 9, characterized zei chnet du rch that a second amplifier (7), the input side indirectly or directly with one of the amplifier unit (5, 6) nach- switched coupler unit and the output side is in each case connected to the second inputs of the phase locked loop (1, 3) and the amplitude control loop (2, 4, 7, 8), wherein at ei ^¬ nem control input of the second amplifier (7) a second output signal of the second means ( 10) for changing the gain of the amplitude control circuit (2, 4, 7, 8) is applied.

9. Device according to claim 8 or 9, characterized in that a means is provided for controlling the amplitude control circuit (2, 4, 7, 8).