EP1573891B1 - Resonance converter and method for driving variable loads - Google Patents

Description

The present invention relates to a switching power supply for driving variable ohmic-capacitive or ohmic-inductive loads, comprising a resonant circuit, an electromechanical energy converter, a switch and a control device.

Switching power supplies with or without resonant circuit usually do not come without inductive electromagnetic components. To achieve a low-loss switching operation such circuits can be operated only up to a certain maximum frequency and only with resonant inductive elements or broadband transformers or inductors. Such components are volume-intensive and cause a significant share of the costs of the entire device.

For example, a self-or externally excited half-bridge circuit is to be cited, which works with bipolar transistors, reverse-sequence diodes, a series resonant circuit and inductive base feedback. An exemplary embodiment of such a half-bridge circuit is disclosed in the following document (1): Lowbridge, M. Maytum, K. Rutgers, "Electronic Ballasts for Fluorescent Lamps Using BUL 770/791 Transistors" (Texas Instruments, 1992 ). The load circuit is predominantly inductive, whereby a low-loss switching in different load cases is possible. This circuit can also be classified as a class D amplifier. It would have the disadvantage of capacitive clearances even using minority charge-free MOS (MOS) transistors because the switches must be turned on under voltage unless an output side resonant choke transitions the voltage to approximately zero at power up can rise to a respective switch. Thus, ZVS (ZVS = Zero Voltage Switching), which is characterized in that when switching a voltage across a power semiconductor before and during a switching operation is made to zero, achieved by a sufficiently large (resonant) inductance on the load circuit.

Furthermore, there are Class E HF (RF) high frequency (RF) amplifiers with only one switch and high efficiency. An embodiment of such a circuit is published in the following document (2): NO Sokal, AD Sokal, "Class E - A New Class of High Efficiency Tuned Single-Ended Switching Power Amplifiers", (IEEE, Journal of Solid-State Circuits, Vol. SC-10, No. 3, June 1975 ). Such amplifiers are mainly used as a transmission amplifier and operated with an externally generated clock at an optimal on-time. The switch-on time is usually about half a period (D = 0.5 corresponds to optimum). D denotes the relative (ie related to a period duration) on-time. This circuit also requires a resonant inductance in the load circuit, but achieves zero-voltage behavior (ZVS) in parallel with a sufficiently large capacitance. Whereas in the case of a half-bridge circuit the parallel capacitance to the switch is chosen to be as small as possible in order to achieve the zero-voltage behavior (ZVS) without problems by means of a resonance inductance, in the case of said class E circuit this parallel capacitance is made as large as possible to be the maximum voltage across the switch to be as small as possible during turn-off. However, if the capacitance is chosen too large, the voltage can not return to zero, and impermissible switch-on losses occur.

When using high-frequency piezoelectric transformers (piezotransformers) or other energy converters with an electromechanical energy conversion, any desired transformation ratios can be realized. However, these components usually do not provide predominantly inductive input behavior. Such electromechanical converters are usually also very narrow-band and can transmit only sinusoidal oscillations with regard to their frequency behavior. A hard-switching converter topology is therefore less suitable for their operation. Thus, resonant operation, favorably also in a resonant converter topology, must be chosen. Since a capacitive input and output behavior is essentially predetermined by a piezoceramic material, such a converter can only replace conventional inductances or transformers if care is taken in the event of a desired inductive load circuit behavior for additional inductive shaping of the load circuit. In a half-bridge circuit such an inductive load circuit behavior is required to keep the switching losses small. The simplest measure is an additional, albeit small conventional inductance in the load circuit insert. If the turn-on losses are small enough due to correspondingly low input voltage levels (eg extra-low voltages up to 24 V), a capacitive behavior of the electromagnetic converter in the half-bridge may also be acceptable.

Finally, the switching in a resonant case using a piezoelectric transformer can be designed so that the switching losses are minimized when a Umladezeit the relatively large input capacitances of the piezoelectric transformer by accurately observing required driving times by temporarily switching off both switches (dead times) bridged becomes. For this purpose, however, a precisely adjustable high-side and low-side driver circuit is required, which further usually has an integrated circuit. An embodiment of such a circuit is published in the following document (3): RL Lin, Lee FC, EM Baker, DY Chen, "Inductor-less Piezoelectric Transformer Electronic Ballast for Linear Fluorescent Lamps ", APEC2001, Anaheim, CA, USA, Proceedings, Vol. 2, pp. 664-669 ,

In a class E resonant circuit, the predominantly capacitive input behavior of a piezo transformer is beneficial in that the magnitude of the input capacitance can be adjusted to an electrically required value and thus not disturbing, as is the case with a half-bridge or other inductively-acting load circuit. Such class E circuits with a piezoelectric transformer are already known from the following document (4): T.Abe, Sh. Jomura, T. Tamakai, "Discharge tube driving device and piezoelectric transformer," EP 0 665 600 B1 , European Patent of 21.07.1999.

However, such circuitry is not used in document (4) for the case of a large input voltage and a small output voltage, which is technically used in mains voltage applications, but is used to up-transform from a smaller voltage to a larger one. This limitation to small input voltages was previously determined mainly by the lack of availability dynamically faster, high-blocking circuit breaker, which can now be produced inexpensively, for. B. Fieldstop IGBT (IGBT = Integrated Gate Bipolar Transistor) up to 1700 V or Cool MOS transistors up to 800 V.

In low-voltage applications, a class E circuit according to document (4) and according to document (2) is usually used in optimum operation with the relative on-time of D = 0.5. In the case of the high transformation, such a circuit usually requires an additional input parallel capacitance if the input capacitance of the piezoelectric transformer is not sufficiently large. This is not the case in a transformer-off case where the input capacitance of some embodiments of piezoelectric transformers may be too large.

In addition, there are single-transistor circuits with a piezotransformer which require a resonant inductance which does not, or not exclusively, be smoothing and thus must be suitable for a high frequency of about 50 to 200 kHz by a suitable choice of a magnetic material and a stranded wire. An embodiment of such an arrangement is shown in the document (5), US 6,052,300 , disclosed. In addition, an input-side smoothing choke prevents a direct action of high-frequency current oscillations on an input or on a smoothing capacitor with respect to a non-input side smoothing or Resonanzinduktivität, so that an input-side smoothing reactor (hereinafter called inductor inductance) is preferable to other arrangements of inductance.

With regard to the control of circuits with a piezoelectric transducer, phase lock loop (PLL) is a typical way of frequency tuning. In Scripture (7), US 5,866,968 , a possibility is described to adjust the phase shift between the output voltage and the drive signal of a circuit according to (4) so that a PLL circuit with a simple oscillator / driver IC can be realized. This class E control circuit is well suited for piezotransformers with high transformation properties, since the maximum voltage at the output of the transformer is a significant point at the same time for the rated power. Because of the low current load in the high transformation, the frequency characteristic of the output voltage will almost correspond to an idle case, so that the transformation ratio between idle and nominal load changes little. In (8) is thus essentially a phase control over the voltage waveforms between input and output, so that always sets a maximum output voltage when the correct phase position (in this case about 90 ° or slightly less) is set. This also applies to other topologies with heavy up-transformation the voltage, for example for the half-bridge circuit. In the case of the transformation, a flattening of the transmission characteristic of the output voltage can be observed, since the secondary-side current load significantly influences the voltage transmission ratio. In this case, with an inaccurate fixation of the nominal point in applications, for example for low-pressure gas discharge lamps or power supplies, very different output powers occur if an adjustment to the phase between the voltages would take place. Gas discharge lamps have a negative differential resistance characteristic, and are sufficiently fixed by the lamp current with respect to their rated power. If one uses the phase shift between the maximum of the output current and an input variable as the basis for a control, so the desired nominal power is hardly adjustable by the Exemplarstreungen of lamp (nominal voltage) and piezotransformer, regardless of the topology. Thus, the regulation must be made to a certain nominal value of the output current, which is not necessarily the maximum transmittable current. A fundamental solution for setting a PLL control according to this principle with this same disadvantage has become known from (3). Consequently, for adjusting the lamp current in (3), a very accurate control circuit must be used which either requires a special nominal value adjustment for each device in order to reach the nominal point. Or the value of the output current is sampled with sufficient processing effort. A phase control by sampling the zero crossings of output voltage and output current in a half-bridge circuit is again inaccurate because of the dispersion of Umladezeiten at the input of the piezoelectric transformer, so that there is an evaluation of the amplitude of the output current is required to adjust the rated power.

Out " A Very Simple DC / DC Converter Using Piezoelectric Transformer ", MJ Prieto et al., IEEE, 2001, pages 1755 to 1760 are known power supplies with piezo transformer in a half-bridge topology.

From the EP 0782374 A1 For example, an inversion circuit for igniting a cold cathode arc tube by using a piezoelectric transformer is known. The inversion circuit includes a piezoelectric transformer for supplying a voltage signal to the arc tube, a driver for driving the piezoelectric transformer, and a voltage controlled oscillator for generating an oscillation pulse voltage signal having an oscillation frequency controlled by a control voltage. For generating the control voltage, a control circuit is used, which generates the control voltage based on a detected phase difference between a voltage at the output of the driver and a voltage at a connected to the output of the piezoelectric transformer voltage divider.

Furthermore, from the US6188163B1 an inversion circuit for igniting a fluorescent tube by using a piezoelectric transformer, the piezoelectric transformer having a transformation ratio in normal operation, and having a transformation ratio in the ignition mode.

The object of the present invention is to provide a resonant converter and a method for efficiently driving variable loads.

This object is achieved by a resonant converter for driving variable loads according to claim 1 and a method for driving variable loads according to claim 16.

The present invention is based on the finding that a piezotransformer can be used for driving variable loads in a rated load operation for the Abtransformationsfall by a switch for switching a voltage applied to the piezotransformer voltage signal is used whose switching frequency based on a phase shift between a switch current and a load current is controlled.

By the presented invention, a switching power supply or oscillator is defined, which is basically constructed as a class E amplifier with a piezoelectric transformer, but in its operation from an optimum, which is given by D = 0.5, deviates downward, so that the switch current during the on-time is only increasing, where D is typically in an interval of 0.25 to 0.45, and a maximum of a switch voltage is limited to an approximately three times the value of the input voltage. D is considered as the relative on-time of only the positive history of the switch current. In addition, a negative switch current profile can and should occur by eg an antiparallel diode to the switch in all operating cases, whereby the zero voltage switching (ZVS) can always be guaranteed.

This measure for sensibly limiting the switch voltage in mains voltage applications is from document (6), EP 0 681 759 B1 , and further known in the following document (8): LR Nerone, Novel Self-Oscillating Class E Ballast for Compact Fluorescent Lamps, IEEE Trans. On Power Electronics, Vol. 16, No. 2, March 2001, pages 175-183 , described. Thus, a rectified mains voltage of about 80 to 150 volts or 170 to 260 volts can be applied to an input of a class E amplifier without exceeding a maximum allowed switch maximum (eg, 600 V for 120 V AC and DC) 1200 V for 240 V AC voltage). In addition, a piezoelectric transformer on the input side can be connected directly in parallel with a switch, which takes the Abtransformation to the load and guaranteed by its capacitive input behavior a desired return of the switch voltage to zero over a defined load or input voltage range.

In order to avoid the need for additional reactive load circuit components in this circuit, a voltage transformation ratio of the piezoelectric transformer is just chosen to match the load impedance, and an input capacitance of the piezotransformer is selected so that it can resonantly store the required reactive power component, so that neither the Switch voltage is exceeded, nor does the voltage return to zero. Compared with the circuit according to document (4), the external capacitance shown there parallel to the switch is unnecessary, since the input capacitance of the piezoelectric transformer for a mains voltage application can be chosen sufficiently large, while their value is achieved in low-voltage applications of a piezoelectric transformer less well and below Circumstances is too small.

In addition, the circuit according to the invention requires comparatively to half-bridge circuits for mains voltage applications only a low-side driver and thus has a reasonable tax expense. This simplifies the control effort for the entire circuit and is comparable to a driving effort of a hard-switching DC-DC converter (flyback or boost arrangement).

In addition, the switch comes only for a short time, and comparable to the effect of a power source, in a reverse operation and therefore operates very low loss especially at a use of MOS transistors, but also when using IGBT with reverse diode even at high frequencies up to 100 kHz ,

The present invention makes it possible, within certain limits, to drive variable loads with low loss and with a simple driving effort at high frequencies, with only a minimal amount of circuitry including, for example, a switch (MOSFET or IGBT with a reverse diode), an input direct current choke (Drosselihduktivität) and an electromechanical Energy converter (piezoelectric transformer) is obtained. In this case, a rectified mains voltage with certain fluctuations in the input voltage is just as useful as a constant input DC voltage. The converter (resonant converter) generates a nearly sinusoidal output voltage due to a high quality of the electromechanical transformer, whereby the crest factor at downstream ohmic loads, such as gas discharge lamps, can be kept sufficiently small. This is possible in a conventional circuit, as disclosed for example in document (2), only with a large load circuit quality, which in turn would result in an increased current density load of a decoupling capacity, as well as a switch-parallel capacity. The circuit used in this way works in total with low losses and the maximum usable frequency is essentially limited only by the dynamic losses of the switch. A fieldstop IGBT is through its short tail time and a relieved switch off are very well suited for this application.

Thus, in the invention, a few low-cost components are combined to meet the technical requirements of off-line rectification offsetting for typical burn voltages for low pressure gas discharge lamps. At the same time, the electromechanical transducer (piezoelectric transformer) satisfies the requirement of up-transformation in an unloaded state, so that a low-pressure gas discharge lamp can be easily ignited. Prior to ignition, such a lamp represents a very large resistance, which in an ignited state (combustion mode) changes into a defined load with a negative differential resistance, and can be approximately approximated with an ohmic resistance at an operating point. The ignition circuit realized by further components in a conventional ballast can be realized in a ballast with a piezoelectric transformer exclusively by this transformer, whereby a further cost reduction is brought about.

Furthermore, by utilizing a load-dependent phase shift between a load current and a switch current, such a rated load point is set that it can be regulated via a phase locked loop (PLL). In this case, with a sufficient bandwidth of a piezoelectric transformer, a simple integrated drive circuit can be used. A detection of an input or a lamp voltage is not required for an adjustment of an operating point, since a parameter dependence of a phase shift is small enough to adjust the lamp power alone via a setpoint adjustment of the phase shift. Also, the amplitude of the output current need not be sampled for approximate power adjustment because of the change in the transformation ratio If the load changes, the nominal power can be reproduced exactly enough on the phase shift of the current zero crossings of the switch and load current.

Fig. 1

ein grobes Blockschaltbild, das einen prinzipiellen Aufbau eines erfindungsgemäßen Resonanzkonverters zeigt;

Fig. 2

ein Schaltungsdiagramm eines Resonanzkonverters, wobei eine Steuereinrichtung zum Steuern der Schaltfrequenz des Schalters nicht dargestellt ist;

Fig. 3

ein detailliertes Schaltungsdiagramm des Resonanzkonverters aus Fig. 2;

Fig. 3a

eine frequenzabhängige Spannungsübertragungsfunktion eines piezoelektrischen Transformators im lastfreien Zündbetrieb und im Lastbetrieb.

Fig. 4

qualitative Kurvenverläufe von einem Schalterstrom I_S und einem Laststrom I_L;

Fig. 5

und 5a Verläufe eines Phasenwinkels Φ_LT in Abhängigkeit von der Frequenz, sowie frequenzabhängige Spannungsübertragungsfunktionen in Abhängigkeit von der Ausgangslast und der Eingangsspannung;

Fig. 6

den Phasenwinkel Φ_LT bei einer konstanten Frequenz in Abhängigkeit von einer Eingangsspannung U_in;

Fig. 7

eine Schaltung zum Treiben einer veränderlichen Last gemäß einem weiteren Ausführungsbeispiel der vorliegenden Erfindung; und

Fig. 8

eine Ansteuerschaltung gemäß einem weiteren Ausführungsbeispiel der vorliegenden Erfindung;

Preferred embodiments of the present invention will be explained below with reference to the accompanying drawings. Show it:

Fig. 1

a rough block diagram showing a basic structure of a resonance converter according to the invention;

Fig. 2

a circuit diagram of a resonant converter, wherein a control device for controlling the switching frequency of the switch is not shown;

Fig. 3

a detailed circuit diagram of the resonant converter of Fig. 2;

Fig. 3a

a frequency-dependent voltage transfer function of a piezoelectric transformer in no-load ignition and load operation.

Fig. 4

qualitative curves of a switch current I _S and a load current I _L ;

Fig. 5

and FIG. 5a shows characteristics of a phase angle Φ _LT as a function of the frequency, as well as frequency-dependent voltage transfer functions as a function of the output load and the input voltage;

Fig. 6

the phase angle Φ _LT at a constant frequency as a function of an input voltage U _in ;

Fig. 7

a variable load driving circuit according to another embodiment of the present invention; and

Fig. 8

a drive circuit according to another embodiment of the present invention;

In Fig. 1 is a rough representation of a resonance converter according to the invention is shown, which comprises a source 101, a switch 103, a piezo transformer 105, a variable load 107 and a control device 109. A voltage supplied by the source 101 or a current supplied thereby is switched by means of the switch 103 at a switching frequency, whereby an input signal is applied to the piezo transformer 105, which is converted into an output signal having a frequency which is the switching frequency of the switch 103 depends. This output signal serves to drive a load 107, for example a low-pressure gas discharge lamp, whose load characteristic is variable. The switching frequency at which the switch 103 is switched is controlled by the controller 109 based on a phase shift between the current through the switch 103 and the load current through the load 107. This phase shift can be determined from a plurality of signals which can be tapped, for example, before and after the piezo transformer 105 and before or after the switch 103.

Fig. 2 shows an embodiment of a resonance converter, wherein a control device for controlling the switching frequency is not shown. In this case, the source 101 is coupled to a first terminal 2011 of an input choke 201. A second terminal 2013 of the input choke 201 is coupled to a first input 1031 of the switch 103. The first input 1031 of the switch 103 is coupled to a first terminal 1051 of an input port 1052 of the piezotransformer. The source 101 is further coupled to a second input 1033 of the switch 103, which is further coupled to a second terminal 1053 of the input port 1052 of the piezo transformer 105. The variable load 107 is connected between a first terminal 1055 of an output gate 1056 of the piezotransformer and a second terminal 1057 of the output gate 1056. The switch 103 further has a control input 1035, to which a control signal can be applied, which controls the switching frequency of the switch 103. In the following the operation of the resonant converter shown in Fig. 2 will be described in more detail.

From the source 101, which may be a DC voltage source, an approximately constant or sawtooth DC current is fed via the input choke 201. The switch 103 is operated with a relative on time D and an operating frequency f, so that a resonance of the converter 105 is reached, and an output signal, such as a voltage, the variable load 107, such as a gas discharge lamp or other ohmic-capacitive load, drives.

Fig. 3 is a detailed circuit diagram of a resonant converter including a class E amplifier. The source 101 is first coupled to the first terminal 2011 of the input choke 201. The second terminal of the inductor inductor is coupled to the first input 1031 of the switch 103, the first input 1031 being further coupled to the first terminal 1051 of the converter 105. The source 101 is also coupled to the second input 1033 of the switch 103, the second input 1033 being further coupled to the second terminal 1053 of the converter 105. Between the first terminal 1055 and the second terminal 1057 of the output port of the converter 105, the load 107 is arranged. The switch 103 in this embodiment comprises a voltage-controlled power switch 1037 whose source or emitter is coupled to the first input 1031 of the switch and whose drain or collector is coupled to the second input 1033 of the switch 103 are. The control input 1035 of the switch 103 is simultaneously implemented as a gate of the voltage-controlled circuit breaker 1037 in this embodiment. Between the second input 1033 and the first input 1031, a diode 1039 is connected in the flow direction.

Furthermore, a simplified equivalent circuit diagram of a piezotransformer 105 is shown in FIG. The equivalent circuit comprises an input capacitance 10501, which is connected between the first terminal 1051 and the second terminal 1053 of the input port of the piezo transformer 105 and is thus arranged parallel to the switch 103. Further, the equivalent circuit of the converter 105 comprises a resonant circuit consisting of a series circuit of a capacitance 10502, an inductor 10503 and a resistor 10504. In addition, the equivalent circuit of converter 105 includes a transformer arrangement 10505 whose inverse transfer ratio is 1 / μ (1 / μ = effective input voltage to effective output voltage) frequency dependent and, in a nominal load operation where the reactive power component is less than the active power component, is between 1, 5: 1 and 5: 1. The resonant circuit, which is further characterized by a high quality, consists of the capacitance 10502, the inductor 10503 and the resistor 10504, is connected between the first terminal 1051 of the converter 105 and another terminal 10506 a primary side of the transformer assembly 10505. Parallel to a secondary side of the transmission arrangement 10505, an output capacitance 10508 is arranged.

The piezo transformer 105 is characterized in that the transmission ratio ü is subject to change depending on the load 107. The gas discharge lamp 301 connected between the terminals 1055 and 1057, which is embodied by the load resistor 107, is connected to the converter as a load. The voltage controlled power switch 1037 may be, for example, a fast IGBT (eg, a field stop IGBT) or a MOS transistor (For example, a cool MOS transistor), which is used together with an anti-parallel reverse diode. In the following the operation of the circuit shown in Fig. 3 will be explained.

When the voltage-controlled power switch 1037 is made conductive by application of a control signal to the control input 1035, a current flowing through the voltage-controlled power switch can not rise abruptly due to the input reactor 201. In addition, the input capacitance 10501 of the converter 105 is discharged. If the voltage-controlled power switch 1035 is switched off again by applying a corresponding control signal, that is to say in a blocking state, then a voltage across the voltage-controlled power switch only grows slowly, since the input capacitance 10501 charges up. Due to the effect of the resonant circuit of the converter 105 in the steady state, in spite of positively flowing further input current through the inductor 201, a current reversal in the switch is achieved, as a result of which the capacitance 10501 is again discharged. The voltage across the switch 103 thus becomes zero again, and a negative current begins to flow through the switch. The freewheeling diode 1039 has the task of conducting a reverse current even before switching on the voltage-controlled circuit breaker 1037. In this case, a relative or an absolute switch-on time of the switch remain close to constant, since the diode does not have to be switched, but is subject to a current-controlled mode of operation. As long as the diode carries the reverse current, the voltage-controlled power switch at the gate 1035 can therefore be de-energized with respect to the collector / emitter or drain / source, so that no turn-on losses occur. Such a current-controlled antiparallel diode is not necessarily designed as a fast diode, so that in this case also a low-cost slow diode can be used.

If the switch 103 is now operated at a predetermined frequency, the resonant circuit consisting of the capacitance 10502, the inductor 10503 and the resistor 10504 is excited. If a resonant frequency of the resonant circuit is reached, the converter 105 reaches a maximum voltage transfer ratio u. When using a piezoelectric transformer can be, for example, a voltage transfer function (at a defined input voltage 101 and a defined load 107) with respect to the frequency as described by a Gaussian function (bell curve), as illustrated for example in Fig. 3a. At a resonant frequency f _R , the voltage transfer function reaches a maximum value in the load state. If the resonance frequency f _{R is} exceeded, which corresponds to an over-resonant case, then the voltage transfer function decreases such that it follows a course of the Gaussian curve. For example, at a frequency f ₁ above the resonant frequency, the voltage transfer function has assumed a value that is significantly less than the value of the voltage transfer function in the resonant case. If the frequency decreases again in the over-resonant mode, the voltage transfer ratio ü increases again.

This frequency dependent voltage transfer ratio of a piezoelectric transformer shown in Fig. 3a is now exploited according to the present invention to drive variable loads. The gas discharge lamp exemplarily connected to the secondary side of the transformation arrangement 10505 is characterized by a variable load characteristic. In order to ignite a gas discharge lamp, it is necessary that at the output port of the converter 105, a high voltage is applied, which enables the ignition of the gas discharge lamp 301. If the gas discharge lamp 301 is ignited in the no-load state, then the voltage applied to the gas discharge lamp 301 drops while the load current flowing through the gas discharge lamp increases. Is a nominal load operation (load condition) achieved, that is, the gas discharge lamp is converted into a combustion operation, so its load behavior, as already stated, be approximated by a variable resistance in each operating point, wherein a negative differential characteristic of the ohmic resistance occurs.

Now, if the voltage transfer function chosen by a suitable design of the electro-mechanical transducer 105 so broad that a deviation from the resonant frequency, a suitable reduction of the voltage transfer ratio occurs, it can be counteracted an increase in a voltage at the gas discharge lamp during load operation. If the output voltage between the first terminal 1055 and the second terminal 1057 of the converter 105 increases, the piezoelectric transformer, because of its capacitive output due to the capacitance 10508, acts as a class E converter with a predominantly capacitive output load. As a result, the transmitted total power does not decrease to such a degree as if a constant ohmic resistance were operated as a load with the same frequency change. The transmitted total power is divided into the reactive power conducted via the capacitor 10508 and the active power conducted via the load 107. By a decrease in the load current, but a simultaneous increase in the load voltage (burning voltage), the transmitted total power can drop less with a deviation from the resonant frequency less than a constant ohmic load with the same converter, as a larger capacitive reactive power due to larger output voltage on the capacity 10508 is performed.

Further, the piezoelectric transformer 105 is designed so that it generates at about a same or only a slightly different resonant frequency with respect to the load case (ie, burning operation of the gas discharge lamp) a load-less up-transformation of the output voltage, so that an ignition of the gas discharge lamp is made possible. This attribute is achievable in piezoelectric transformers due to an undamped mechanical vibration in a load-free state in a simple and cost-effective manner by narrowing the resonance curve shown in Fig. 3a in the unloaded state, and the broadband resonant curve in the load state encloses the narrowband load-free curve.

4 shows a diagram of the qualitative curves of the switch current I _S , load current I _L and the phase angle Φ _LT to be detected. In addition, a period T, a switch-off time t _off , a reverse time t _rev and the turn-on time t _on are shown. The phase angle Φ _LT , which is determined by the zero crossings of the switch current I _S and the load current I _L is not zero and relatively large in this exemplary diagram, since the load current I _{L has} a larger capacitive component, which is equivalent to the fact that Gas discharge lamp has not yet been converted to its nominal operation (approximately ohmic resistance), where the phase angle Φ _{LT is} smaller and may even become almost zero. At the same time, the reverse time t _rev becomes ever smaller and can become almost zero, so that the negatively flowing reverse current through the diode 1039 disappears.

The frequency-dependent voltage transmission ratio of a piezoelectric transformer is used in the embodiment shown in FIG. 3 according to the invention to realize a frequency-dependent power transmission in response to a variable load, as has already been explained with reference to FIG. 3a. This will be explained in detail below with reference to the voltage transfer ratio of a piezoelectric transformer 105 shown in FIG. 5 depending on a load characteristic of a gas discharge lamp 301.

In most piezotransformers, the resonant frequency in an unloaded operation is higher than the optimum one Frequency under load (for example, for maximum power or for maximum efficiency). To use this property for a control on a gas discharge lamp, the resonant frequency of the electromechanical transducer is realized without load only slightly above the resonant frequency under load, which is technically easily possible by a suitable design of a piezoelectric transformer. The rated frequency for the nominal load combustion mode is intended to coincide approximately with the resonant frequency in a no-load condition. To switch on the gas discharge lamp, the converter is initially driven starting from the resonant frequency with a frequency which is preferably variable around the no-load resonance point, which increases slowly and / or slowly decreases again, and follows a curve of a voltage transfer ratio curve 501 shown in FIG ,

After a detection of a decreasing reverse current flowing through the switch 103 and a sufficiently large gas discharge lamp current, an ignition of the gas discharge lamp can be detected, as shown in Fig. 5 for the überresonanten case. Immediately after the ignition of the gas discharge lamp, the voltage applied to it decreases, the load current increases, with the result that the voltage transmission ratio is lower, as shown by the curve 503 (capacitive load). The course of the voltage transfer ratio in a burning operation of the gas discharge lamp is described by the curve 505. Starting from the courses of the different voltage transfer ratios as a function of the load behavior of the gas discharge lamp and the frequency shown in FIG. 5, it will now be clear that the frequency-dependent voltage transfer ratio can be used to efficiently control a power output to the gas discharge lamp by varying the piezo transformer 105 is stimulated near frequencies lying, for example can be realized by a suitable switching frequency of the switch 103.

According to the invention, a phase angle φ _LT between the load current and the switch current is evaluated for controlling and regulating the converter constructed in this way in order, for example, to realize a superresonant control.

Also illustrated in FIG. 5 is an example waveform of the phase angle φ _LT at rated load versus frequency (curve 507) together with the voltage transfer functions in a no-load condition (ignition) and a load condition (rated load). It can be seen that the phase angle Φ _LT steadily decreases until a maximum power transmission is reached, while it increases in the direction of load-free operation. In this case, the load changes so that below f _OPT the rated load or an even larger load (small voltage transfer ratio ü) occurs, and above f _OPT a smaller load (greater voltage transfer ratio ü) up to the no-load ignition characteristic in association with the function of the phase angle Φ _LT 507 occurs.

For example, the supersorant range above a frequency f _opt can be used for controlling or regulating the gas discharge lamp power. Accordingly, it is therefore not necessary to detect a maximum value of the gas discharge lamp current to perform the control or the regulation of the converter. It is sufficient to sample the phase angle Φ _LT between the switch and the load current and set it to a nominal value. If the frequency becomes smaller, the active power transfer increases in an over-resonant operation up to its maximum at the resonance frequency.

This has the consequence that a capacitive portion of the load current decreases and the gas discharge lamp current, which in the ignited gas discharge lamp (combustion operation) approximately in phase with the lamp voltage flows closer to the phase of the switch current approaches. During switching on, the switch current embodies approximately the input current of the piezo transformer 105, which is distributed via the transformation ratio to the load (gas discharge lamp) and to the output capacitance 10508 of the converter 105.

Furthermore, FIG. 5a shows a dependence of the output power transmission on the input voltage at a constant output impedance. The power can be increased under rated load by increasing the input voltage from a minimum rated input voltage 505 'to a higher input voltage 503' to a maximum load characteristic 501 '. In addition, the output power can not be increased significantly, this being dependent on the volume of the piezoelectric transformer used. A smaller volume allows only a smaller maximum load. It is therefore important to ensure that the piezotransformer is designed at least for a load slightly greater than the rated load, so that the control circuit of FIG. 8 remains functional beyond the rated load.

The course of the phase angle Φ _LT at a constant frequency is shown once again in FIG. 6 as a function of an input voltage U _in applied to the gas discharge lamp. At a rising voltage U _in the phase angle Φ _LT decreases, since in this case more active power is transmitted to the gas discharge lamp, see, for example, Fig. 5a, überresonanter operation. This has the consequence that the active component of the lamp current increases. From this example it is clear that variations in the input voltage U _in reflected also in the size of the phase angle Φ _LT. In addition, such fluctuations of the voltage U _{in can be} compensated by more power is passed to the gas discharge lamp by lowering the frequency at a falling input voltage U _in in the over-resonant mode of operation. With increasing input voltage, less power can be transmitted to the gas discharge lamp be increased by increasing the frequency. If the phase angle is kept approximately constant, the transmitted active power at the load also remains approximately constant. Since the gas discharge lamp keeps the output voltage almost constant due to its weakly negative differential ohmic resistance, an approximately constant phase angle φ _{LT can be} set by setting a different transmission ratio u as a function of the input voltage. This phase angle is at constant output voltage due to the parallel circuit of the approximately constant capacitance 10508 and the load 107 in load operation, a measure of the size of the load current, and thus for the output power.

FIG. 7 shows an exemplary embodiment of a resonance converter according to the invention for low-pressure gas discharge lamps, including switching frequency control. Since this embodiment is based on the embodiment shown in Fig. 3, the functionalities will not be described again with the same reference numerals in the following.

In addition to the embodiment shown in FIG. 3, the embodiment shown in FIG. 7 initially comprises an input rectifier 701 having a first mains terminal 70101 and a second mains terminal 70103. Between an output 7015 and an input 7017 of the input rectifier 701 is a capacitance 703, e.g. a charge capacitor can be coupled. In addition to the capacitor 703, a drive part 705 is further coupled together with a resistor 70501. The output 7015 of the input rectifier 701 is further coupled to the first terminal 2011 of the input inductor 201. The drive part 705 further has a control output 7051, which according to the present invention is coupled to the control input 1035 of the switch 103, which in this embodiment comprises the current-controlled power switch 1037. The drive part 705 further has a first one Input 7053 and a second input 7055 on. The first input 7053 is coupled to the second input 1033 of the switch. Between the first input 7053 of the drive part 705 and the input 7017 of the input rectifier 701, a sense resistor 707 is further arranged. Between the load 107 and the second terminal 1057 of the converter 105, a second sense resistor 709 is arranged. The second input 7055 of the driver 705 is coupled between the load 107 and the second sense resistor 709.

The drive part 705 further has a power supply input 7057, which is coupled to the input 7017 of the input rectifier 701 via a capacitor 70111, which may be designed, for example, as a blocking capacitor. Between the second terminal 1053 of the converter 105 and the power supply input 1057 of the drive part 705, a first diode 70131 is coupled in the flow direction. Between the input 7017 of the input rectifier 701 and the first terminal 1051 of the input port of the converter 105, there is further coupled a parallel circuit consisting of an external capacitance 70151 and a diode 70171 operated in the forward direction. In the following, the operation of the resonance converter shown in Fig. 7 will be explained. However, this does not again deal with the functionalities that have already been discussed with reference to the embodiment shown in FIG.

The task of the control part 705 is to detect the indicated in Fig. 7 with an arrow switch current I _S and the load current I _L suitable to determine a phase difference between the two currents, and so at the control output 7051, a control signal for controlling the Switching frequency of the switch 103 output. For this purpose, a variable dependent on the switch current I _S is initially generated, which can be applied to the first input 7053 of the control part 705. In this embodiment, the switch current I _s at the first sense resistor 707 is converted to a voltage applied to the first input 7053. It should be noted, however, that the size dependent on the switch current can be generated by any functionality, such as a current mirror or a current-controlled voltage source.

On the output side, the piezotransformer 105 with a voltage transfer ratio ü drives a gas discharge lamp with the load resistor 107, through which the load current I _L flows. In order to detect a variable dependent on the load current I _L , a second sense resistor 709 is used in the exemplary embodiment shown in FIG. 7, so that the load current I _L generates a voltage across the resistor 709 which is applied to the second input 7055 of the drive part 705 is applied. On the basis of these two voltages, the phase difference between the switch current I _S and the load current I _{L is first} determined in the drive part 705 and, as has already been described above, a control signal is output which controls the switching frequency of the switch 103.

The power supply of the drive part 701 is realized via a primary-side connection of the piezoelectric transformer 105 via a pumping circuit with the diodes 70131 and 70171, and via the external capacitor 70151, while the capacitor 70111 (blocking capacitor) Supply voltage of the control part 701 smoothes. Thus, a simple power supply device according to the present invention contains no special requirements for electromagnetic compatibility and without further options for dimming or power factor correction, only three capacitances 703, 70111 and 70151, which are designed for example as capacitors, an input rectifier 701 (mains rectifier), an input choke 201, a piezo transformer 105, for example a fast IGBT 1037 with a reverse diode 1039, a possibly integrated control section 705, two diodes 70131 and 70171 as well as some miniature resistors.

The ballast thus obtained can thus be accommodated in a compact design in the smallest space, for example, a height of 10 mm is easily accessible. For example, a size EF 13 up to a power of 18 watts is sufficient for the input choke 201 (choke inductance). For example, for the piezoelectric transformer 105, a cylindrical design with a height of 9mm and a diameter of 20mm may also be considered sufficient for 18W. The transistor 1037, for example, implemented as a fieldstop IGBT, can be housed in a small SOT package, and the drive IC (IC) for the driver 705 can be packaged in a standard 8-pin package. A complete integration of the 1039 reverse diode, such as a Fieldstop IGBT 1037 and a drive IC, is also cost-effective in an 8-pin package as a multi-chip solution.

FIG. 8 shows an exemplary embodiment of the control device 109 according to the invention together with the switch 103 and the load resistor 107.

The control device 109 comprises first a device 801 for detecting a dependent of the switch current I _S size, means 803 for detecting a dependent of the load current I _L size and a phase locked loop 805. The phase locked loop 805 comprises in this embodiment means 807 for determining the Phase shift between switch current and load current from the quantities detected by device 801 and device 803. The device 807 has a first input 8071, a second input 8073 and an output 8075. The output 8075 of the device 807 is via a Resistor 8091 and a capacitor 8093 coupled to a reference potential, such as ground.

Means 805 further includes a voltage controlled oscillator 811 (VCO) and a gate driver 813. An input 81101 of the VCO 811 is coupled between the resistor 8091 and the capacitor 8093. An output 81103 of the VCO is coupled to an input of the gate driver 813 whose output is coupled to the control input 1035 of the switch 103. In this exemplary embodiment, the device 801 has a comparator 8011 with a first input 80111, a second input 80112 and an output 80113. The first input 80111 of the comparator 8011 is coupled to the second input 1033 of the switch 103. The second input 80112 is coupled to the output 80131 of a reference source 8013. The output 80113 of the comparator 8011 is coupled to the first input 8071 of the device 107. The device 803 comprises a comparator 8031 having a first input 80311 and a second input 80312 and an output 80313. The first input 80311 of the comparator 80131 is coupled between the resistors 107 and 709. The second input 80312 of the comparator 8031 is coupled to the output 8031 of the reference source 8013. The output 80313 of the comparator 8031 is further coupled to the second input 8073 of the device 807.

In the following, the operation of the embodiment shown in Fig. 8 will be explained.

In the embodiment shown in FIG. 8, the switch current at the sense resistor 707 is converted into a voltage applied to the first input 80111 of the comparator 8011. At the second input 80112 of the comparator 8011, a reference signal supplied from the reference source 8013 is applied. The comparator 8011 thus samples the zero crossings of the switch current I _S by a comparison between the one falling on the sense resistor 707 Voltage and the reference signal near zero. At the output 80113 of the comparator 8011, an output signal is thus output whose instantaneous phase results from the comparison between the signals present at the inputs 80111 and 80112 and which in this embodiment represents a variable dependent on the switch current I _S. A symmetrical arrangement is located on the load side. At the sense resistor 709, the load current I _{L is converted} into a voltage applied to the first input 80311. At the second input 80312 is also the reference signal 80131, which is supplied by the reference source 8013. By comparison with the reference signal near zero, the load current is sampled via the sense resistor 709 and the comparator 8031 outputs at its output 80313 an output signal representing a magnitude dependent on the switch current I _L. In this embodiment, the two second inputs 80112 and 80312 are coupled to the same output 80131 of the reference source 8013. This reference source is designed in this embodiment as a DC voltage source. It should be noted that reference source 8013 may be any source, such as an AC source, or other arrangement such as a current or voltage controlled voltage source that provides a predetermined, eg, time-dependent reference signal.

In the embodiment shown in FIG. 8, the switch current and the load current at the two sense resistors 707 and 709 are sampled. At this point it should be noted that as the switch current and the load current can be sampled using any functionality, such. B. a current mirror with a resistor as a load or a current-controlled voltage source, or by a separate load or switch current proportional signal source, eg as a tap of the transformer 105 (piezoelectric transformer) in Fig. 7.

Means 807 is for detecting the phase shift between the switch current and the load current from the detected quantities applied to the two inputs 8071 and 8073. In this embodiment, means 807 is implemented as a phase detector which is part of phase locked loop 805. The phase difference signal determined by the phase detector 807, which further depends on its frequency dependency on whether there is an over-resonant or an under-resonant operation, is integrated by the integrator 809, in this embodiment as a filter consisting of a resistor and a capacitor. The filter output signal is applied to the input 81101 of the VCO 811, which generates a suitable frequency f and an associated duty cycle D _f from the filter output signal. This output signal is forwarded to the control input 8035 of the switch 103. It should be noted here that the integrator device 809, which in this example is particularly cost-effective, can also be implemented in other ways, for example by a suitably connected operational amplifier, or another time-delaying circuit.

In this embodiment, the switch 103 includes a voltage controlled power switch 1037. The output of the VCO 813 is supplied to the gate driver 813, the output of which is passed to a gate of, for example, a field stop IGBT or MOSFET as possible embodiments of the voltage controlled power switch. In this embodiment, the device 807 is implemented as a phase detector for detecting the phase difference and generating a difference signal. This has the advantage that for controlling the switch 103, the phase locked loop 805 can be used, which can be realized inexpensively.

The control of Fig. 8 works as follows: When the output load increases (smaller ohmic resistance), Thus, a smaller transmission ratio will occur according to FIG. 5. At the same time, however, the phase angle φ _LT decreases in the over-resonant case, so that the voltage Up which is output at the output of the device 809 increases. Via the VCO 811, a larger frequency is set by the filter 809 at the gate of the fieldstop IGBT 1037, for example, which causes a reduction of the transmitted power. This leads to the over-resonant characteristic of FIG. 5 to a smaller transmission ratio ü. However, since the output load has a negative differential resistance when it is a gas discharge lamp (fluorescent lamp), the output voltage increases and the lamp current decreases disproportionately. Thus, one arrives at a transfer characteristic above the previous characteristic of FIG. 5, so that by changing the output impedance again a larger phase angle φ _{LT is} set, as shown in Fig. 5 also shown (dotted line 507). As the phase angle increases, so will the phase difference voltage U _p again decrease, and the control comes to a static end value.

The same applies to changes in the input voltage: When the input voltage of the converter increases, so the phase angle φ _LT decreases as shown in FIG. 6. Thus, a larger output power is transmitted according to Fig. 5a. The phase difference voltage U _p thereby increases again and, with a time delay, increases the frequency f via the device 809, so that, according to FIG. 5a, the transmitted power is reduced. This in turn results in an increase of the phase angle φ _LT , so that the control comes to a standstill.

If, for some reason, the control enters the sub-resonant mode (Fig. 5, Fig. 5a), it would no longer function. The VCO 811 may therefore have a means, not shown, to store the angelege frequency at the time of ignition. In an embodiment of the piezotransformer 105 according to the invention the ignition frequency in überresonanten branch of the load curve in Fig. 3a. This ignition frequency is then not targeted during control under load, or only by an amount defined by the parameters of the piezo transformer 105 falls below, so that even a disproportionate change in the phase voltage U _p to smaller values no frequency reduction in the sub-resonant operation permits. For this purpose, the VCO 811 can continue to have a device, not shown, to ensure a minimum, lower limit frequency in load operation, which ensures this behavior.

As already mentioned, the VCO 811 is further characterized by a duty cycle D _f , which is adjustable. In order to explain the operation of the oscillator 811, we return again to FIG. The class E amplifier according to the invention is operated so that the switch is switched on after the switch voltage has become zero, possibly with a time delay. This results in turn-on times of typically DE = 0.25 ... 0.45, in order to achieve an optimal limitation of the switch voltage. According to the invention, these switch-on times are supplied by the voltage-controlled oscillator 811 in such a way that the current in the switch only rises during the switch-on time, as illustrated in FIG. 4 by the profile of I _S in an interval marked by t _on . The VCO 811 is therefore designed so that it provides a duty cycle D _f necessary for this purpose. This can be realized, for example, by means not shown in FIG. 8 for setting a predetermined duty cycle of the output signal of the oscillator 811.

From the above-described embodiments according to the present invention, it will be apparent that the improvements over the prior art mean that a resonant inverter consisting of a self-excited or a foreign-excited class E amplifier with a mode tuned to resonance frequency at a high frequency using a high load circuit electromechanical power converter, high efficiency, and limited load variation and input voltage swing, by using a dynamically fast switch with at least about three times reverse bias voltage versus the maximum DC input voltage. Since only one switch and a relatively simple drive circuit are required, the circuit can be realized as a one-chip solution (eg in a SMART-POWER technology) or in a known cost-effective multi-chip design without a need a bridge-capable high-voltage technology for the drive circuit. Using, for example, conventional high-voltage circuit breakers (eg Fieldstop IGBT up to 1700V, Cool-MOS up to 800V), operation at rectified mains voltage is possible. The required drive circuit operates very low loss, in particular when using MOS transistors or fast IGBT because of the capacitive gate behavior, as well as the switch and the electromechanical transducer. Through the use of MOS switches or fast IGBT a high switching speed can be achieved. Because of the resulting possible increase in frequency up to about 200 kHz or even more, the capacitive and the inductive components of the overall arrangement, such as the input choke, decrease in size. A resonance inductance is thus no longer required, just as little as a high-side driver device, which is not the case for comparable half-bridge solutions with narrow-band energy converters, or only with restrictions with regard to the control accuracy. In addition, no reactive components (capacitors, inductors) are required in the load circuit and are completely replaced by the piezoelectric transformer.

Compared to a complex evaluation of the load current for setting the rated power according to document (3) is for the selected class-E circuit only a phase comparison between the zero crossings of load current and switch current required to set the nominal power. This simplifies the control circuit and can be integrated as an analog circuit on a smaller chip area than if complex evaluation circuits for the amplitude of the load current would be used, or this evaluation is made by costly bridge driver ICs.

The inverse voltage transfer ratio 1 / μ (input voltage / output voltage) of the electromechanical transducer is selected to be 1.5: 1 to 5: 1 in terms of sinusoidal transmission at resonant frequency to accommodate typical grid applications for discharge lamps (eg, low pressure lamps). The input mains voltage can be between 80 and, for example, 260 volts AC. From the electrical filter behavior of the electronic-mechanical transducer (eg piezoelectric transformer) then results in a load voltage (burning voltage) in the typical range of low-pressure discharge lamps of z. B. 50 to 160 V. In other gear ratios would be in this circuit topology a load adjustment with optimal switch voltage limit for the described network applications can not be achieved, which is why the correct dimensioned gear ratio in rated load operation is an essential idea of the inventive solution.

Furthermore, the input capacitance of the electromechanical transducer is to be chosen so that in addition to the parallel to the converter input switched semiconductor switch no further parallel capacitance is needed. The value of this input capacitance will be between 100 pF and 1 nF at a frequency of typically 100 kHz and a power of 10 to 20 watts, depending on the input voltage. For low input voltage (80 to 160 V AC), the value of the capacitance should be about 500 pF to 1 nF, for a large input voltage (160 to 260 V AC), this value should be selected as 100 pF to 500 pF.

The parallel acting capacity of the switch is on the order of less than 200 pF. For other power ranges, the value of the input capacitance shifts up (higher power) or down (smaller power). Such an adaptation is possible by a construction of a piezoelectric transformer. Preferably, a circular or a laterally oscillating piezoelectric transformer is used here. On the other hand, a piezoelectric transformer which works on the basis of a thickness vibration or a rose type transformer is less suitable for this application because they do not allow a corresponding transformation ratio in the given power range and the required input capacitance at a sufficient efficiency. It should be noted, however, that these two types of piezotransformers can be used according to the invention.

In addition, the negative differential resistance of a gas discharge lamp in the burning operation helps to stabilize the class E zero-voltage circuit, and as a load in conjunction with a narrow-band electromechanical converter, this is better suited than a constant ohmic resistance. For this purpose, the piezoelectric transformer is designed so that its voltage transfer function has a sufficient bandwidth, which, as already mentioned, follows in frequency about a Gaussian function, and is chosen so wide that at a deviation from the resonance frequency Reduction of the voltage transfer ratio ü occurs, which counteracts the increase in the voltage at the gas discharge lamp. As a result, a control or regulation for gas discharge lamps via detection of the lamp current can be technically reliably implemented if the frequency bandwidth is up to the drop to half a power at least about 5 to 10% of the nominal frequency. In this area, and by the effect of the output capacitance with increasing lamp voltage, the behavior changes The class-E circuit with respect to the zero voltage circuit and the switch current load hardly, so that no significant changes in the switch maximum current, the switch reverse current and the switch maximum voltage occur at approximately constant relative on-time. This is due to the fact that the output capacitance of the piezoelectric transformer is large enough (eg a few nanofarads) to take enough reactive power even in the no-load condition, and resonantly fed back to the input when the load becomes smaller than its nominal value.

The class E converter responds to an increased capacitive or less resistive output load with an increase in reactive current component without violating the zero voltage condition. In this case, the inherent output capacitance of the piezoelectric transformer has a stabilizing effect in this sense. Thus, the switch voltage continues to return to zero even if the gas discharge lamp has been extinguished or removed again. This only increases the proportion of reverse current in the switch. In the event that no real power is transmitted to the load, the maximum reverse current is equal to the maximum inrush current of the switch (ignition mode). Thus, detection of end-of-life effects or load circuit interruption can be done by sampling the reverse current in the switch without having to monitor the lamp voltage. However, the optimum relative on-time changes with frequency and must be tracked with larger frequency changes within the bandwidth. However, such is less or not required for a fixed point load within a PLL control, and will only be applicable to a power position in a wider range, or when an ignition operation with preheating of the gas discharge lamp is provided. However, because of the greater losses during preheating in the piezoelectric transformer, a preheat-free instant start for this solution is preferable.

When the input voltage fluctuates, the transformation ratio of the electromechanical transducer changes little, so that the power changes approximately with the square of the input voltage. If the DC input voltage of the converter decreases, the active current and the reactive current in the load circuit decrease accordingly, and the switch reverse current decreases. If the reverse bias of the switch is large, the input capacitance of the piezoelectric transformer can be reduced to achieve zero voltage switching (ZVS) to lower input voltages. If, however, the reverse voltage reserve of the switch is small, the input voltage must not fall below a certain minimum value. However, because of the sufficiently large output capacitance of a piezoelectric transformer with decreasing ohmic load, this value is small enough to compensate for the usual voltage fluctuations of the grids and in addition to allow a greater voltage fluctuation at the input charging capacitor. If the load is removed or the load-side gas discharge lamp is ignited, the constant or slightly increasing capacitive output load of the piezoelectric transformer on the one hand causes a maintenance of zero voltage switching by the rising reactive current component compensates for the missing load current component. On the other hand, however, the maximum switch voltage does not become much larger, since the effective input current decreases and must be compensated for by a smaller proportion of the load current, while the total relative on-time remains constant.

If the resistive load increases, the proportion of reactive current decreases to smaller values so that the maximum switch voltage is not exceeded even in this case. If the ohmic load is too large, the zero-voltage behavior would no longer be achieved because the transfer characteristic of the piezoelectric transducer only allows a limited power transmission (maximum load). A load increase comes up to the same limit in power transmission as an input voltage increase of Fig. 5a. Thus, the additional power consumed on the switch is converted to heat when it must be turned on at a voltage greater than zero. Thus, the switch maximum voltage is also not exceeded by the transmitted power is no longer increased. In the event that too much load is used, this can be detected by detecting the voltage return to the switch, so that an overload of the switch can be avoided by the converter is turned off. Thus, one can use as a switch, a component whose maximum permitted voltage in any possible operating state of a gas discharge lamp with electromechanical transducer (piezoelectric transformer) is exceeded. Therefore, a non-avalanche-proof switch (MOSFET or IGBT) is well suited for this application, since the output capacitance of the converter, which acts on the input, has a compensating effect with decreasing resistive load, and a maximum transmittable power can not be exceeded. These properties are usually given by the use of inventively dimensioned piezoelectric transformers.

The use of non-avalanche-resistant components, in particular field stop IGBT as a switch, the present application is cheaper by no protective element against overvoltages must be used on the switch, since the output circuit already ensures the protection of the switch by its electromechanical and thus electrical properties.

As has already been mentioned, the phase angle between the load current and the switch current can be evaluated for controlling and regulating a converter constructed in this way. The switch current is superimposed only by the DC component of the input inductor, which changes the phase shift by a fixed amount, and therefore not or only little depends on the power or input voltage. If the input throttle of the converter is chosen so small that the inductor current can decay to zero or less than zero, one can significantly reduce the proportion of superimposed DC flow from the throttle or almost make it to zero, because then the inductor current is typically in the moment switching on the switch reaches about a zero crossing. Even if the input choke is selected to be larger, a phase detection for power control is possible and only slightly adapted to the respective value of the input choke, since the effective input current in this application is much smaller than the load current.

Also, the fluctuation of the input voltage can be compensated via the phase detection and a corresponding frequency change, since the capacitive component of the output current in the converter increases, if due to decreasing input voltage, the active power is smaller.

If too low an input voltage is applied, the target transformation ratio at rated frequency will not be reached if the load impedance becomes too large due to the negative differential resistance of a gas discharge lamp. The electromechanical transducer usually has the property to transmit at a decreasing input voltage with this square decreasing power.

On the other hand, when the ohmic output impedance becomes larger, the converter can only react with an increase in the output voltage even if a small input voltage is applied. As a result, the transformation ratio shifts towards larger values and the transformer internal losses increase slightly. At the same time, however, the inherent output capacitance of the converter is subjected to a larger voltage, whereby the capacitive current component increases and the ohmic current component decreases. The enlargement of the transformation ratio can be adjusted by a design of the electromechanical transducer so that the output voltage from maximum load (minimum load resistance) to smaller loads (larger load resistance) increases so that the resulting equivalent resistance with respect to the input remains approximately constant or changes little.

This allows the class E converter to operate under variable load in a wide input voltage range without violating the zero voltage condition and by allowing the transmitted power to be varied only by small frequency changes. It is therefore also possible to keep the phase angle between load and switch current approximately constant and thereby compensate for both changes in the input voltage and changes in the output load while maintaining the ZVS condition. This possibility is not given in such a wide range of load and input voltage change in a transformer with constant parameters, in particular with a constant gear ratio.

The size of the input choke can also be used to adjust the power within certain limits at a given frequency. If the input choke is made larger, the transmission power increases, because at the same frequency, because of the electrical characteristics of the class E converter, the effective stored energy in the input choke increases, which is passed to the load circuit. The adjustment of the power through the input inductor, however, is possible only within smaller limits because of the limited bandwidth of the electromechanical transducer and will affect the overall performance insignificantly within the usual tolerances of inductive components. On the other hand, the adjustment of the input choke can be used to adjust the operating point, if another adjustment should not be made. An advantage of the finite design of the input choke is thus the possibility to adjust the lamp power. If the input choke is made too large, it can cause an improved smoothing of the current harmonics to the network (noise voltage), but also causes a necessary adjustment of the input capacitance of the converter to smaller values at a power increase and at a constant transformation ratio or a smaller step-down ratio and more consistent or larger input capacitance of the converter.

For the typical down-conversion ratio of 1.5: 1 to 5: 1 in nominal load operation and the other electrical data mentioned, the values for the input choke required for a typical implementation of the invention are to be chosen between 3mH and 20mH at a typical frequency of 100kHz ,

To set the respective desired rated power, a PLL control loop is put into operation in time after the detection of the ignition, in which the zero crossings are sampled by the switch and load current and passed on to a phase detector. Furthermore, this phase difference is passed to a filter which generates a smoothed output voltage. This is switched to a suitable VCO (voltage controlled oscillator), which should be adjusted to a desired value (setpoint comparison) and has a suitable gain. The output signal of the VCO is returned as a frequency signal with associated duty cycle according to the invention (constant or slightly variable within said range) via a driver to the switch (gate of an IGBT or MOSFET). In this case, the duty cycle may increase slightly with decreasing frequency and slightly decrease with increasing frequency or it is kept constant.

In the preferred embodiment described above, the phase difference signal is positively turned on, thereby producing an approximately constant power. When a Gas discharge lamp eg by aging has a higher burning voltage, then flows a disproportionately smaller lamp current, which can be referred to as load change (load reduction). However, the converter will keep the power approximately constant by first "determining" that the phase angle has become larger due to sinking load. The resulting phase voltage Up has become smaller, according to the diagram. If it is given positive to the VCO, then the frequency drops slightly and the power is increased again. As a result, the phase angle decreases again, and the phase difference voltage increases again according to the diagram. Thus, the regulation comes eventually to a new actual value of the lamp power, but not very different from the initial value. It is therefore a regulation that would be only 100% accurate if one could set the parameters so that the gains of the VCO, the phase difference voltage formation, and the controlled system itself (class E with piezo transformer) would allow complete compensation of the power change. This is not the case in practice, but a tolerance of the power of +/- 10 to 15% is quite acceptable. The input voltage is similar. As this increases, the power increases at a constant frequency. As a result, the phase angle decreases, and the phase difference voltage increases. But this also sets a higher frequency, which again sets a smaller transmission ratio over the controlled system (esp. By transmission curve of the piezoelectric transformer). Thus, the power decreases again, and again, it is expected that power fluctuations in the context of Eingangspannun changes will be kept within small limits, since the scheme is also not completely accurate.

Claims (20)

1. Resonance converter for driving variable loads, comprising:

   a piezo-transformer (105) with an input gate and an output gate for providing an output signal for driving the variable load (107);

   a switch (103) for providing an input signal from a source (101) to the input gate of the piezo-transformer (105);

   control means (109) for controlling a switching frequency of the switch;

   characterized in that the piezo-transformer (105) is dimensioned and connected such that a downward transformation ratio between the input signal and the output signal is from 1.5:1 to 5:1 when providing nominal power to the variable load (107); and

   that the control means (109) for controlling the switching frequency of the switch (103) is designed on the basis of a phase shift between the switch current and the load current at variable load and/or variable output voltage.
2. Resonance converter of claim 1 with an input gate connected to a source (101), which provides an input voltage from which the switch (103) generates the input signal for the input gate of the piezo-transformer (105), wherein an input capacity of the piezo-transformer (105) is adjusted depending on the amount of the input voltage and the nominal output power.
3. Resonance converter of claim 2, wherein the input capacity of the piezo-transformer (105) is fixedly adjusted to a value between 100 pF and 1 nF.
4. Resonance converter of claim 2 or 3, wherein the input voltage is between 80 and 160 volts and the input capacity of the piezo-transformer (105) is between 500 pF and 1 nF.
5. Resonance converter of claim 2 or 3, wherein the input voltage is between 160 and 260 volts and the input capacity of the piezo-transformer (105) is between 100 pF and 500 pF.
6. Resonance converter of one of claims 1 to 5, comprising an input choke (201) connected between the source (101) and the switch (103) arranged in parallel to the input gate of the piezo-transformer (105).
7. Resonance converter of claim 6, wherein the inductance of the input choke (201) has a value of 3 mH to 20 mH.
8. Resonance converter of one of claims 1 to 7, wherein the switch (103) includes a voltage-controlled power switch (1037), which may be implemented as a fieldstop IGBT or a cool MOS transistor, with a first input, a second input, and a control input to which a control signal can be applied.
9. Resonance converter of claim 8, wherein a diode (1039) is connected between the second input and the first input of the voltage-controlled power switch.
10. Resonance converter of one of claims 1 to 9, wherein the control means (109) includes means for detecting (801) a quantity dependent on the switch current, means (803) for detecting a quantity dependent on the load current, and means (807) for determining the phase shift between the switch current and the load current from the detected quantities.
11. Resonance converter of claim 10, wherein the control means (109) comprises a phase-locked loop (805) for regulating a mean phase shift to a constant target value.
12. Resonance converter of claim 10 or 11, wherein the control means (109) further includes an oscillator (811), whose frequency is adjustable depending on the determined phase shift and whose output signal serves for controlling the switch (103).
13. Resonance converter of claim 12, further comprising means for adjusting a predetermined duty cycle of the output signal of the oscillator (811).
14. Resonance converter of one of claims 1 to 13, wherein the variable load includes a gas discharge lamp.
15. Method of driving variable loads by a resonance converter including a piezo-transformer (105) with an input gate, wherein the piezo-transformer (105) is dimensioned such that the voltage downward transformation ratio between the input signal and the output signal is from 1.5:1 to 5:1 when providing nominal power to the variable load (107), a switch (103) and control means (109), comprising the steps of:

    controlling a switching frequency of the switch (103) by the control means (109) on the basis of a phase shift between the switch current and the load current at variable load and/or variable input voltage, in order to apply an input signal to the input gate of the piezo-transformer (105) and thereby generate an output signal for driving the variable load.
16. Method of claim 15, comprising the steps of:

    detecting a quantity dependent on the switch current;

    detecting a quantity dependent on the load current;

    determining the phase shift between the switch current and the load current from the detected quantities.
17. Method of claim 16, with a step of regulating a mean phase shift to a nominal value using a phase-locked loop (805).
18. Method of claim 16 or claim 17, comprising the steps of:

    controlling a voltage-controlled oscillator (811) based on the mean phase shift; and

    using the output signal of the oscillator as control signal for the switch (103).
19. Method of claim 18, with a step of adjusting a predetermined duty cycle of the output signal of the oscillator (811).
20. Method of one of claims 14 to 19, wherein a gas discharge lamp is used as variable load.

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