EP1018289A1 - Systeme de ballast electronique - Google Patents

Systeme de ballast electronique

Info

Publication number
EP1018289A1
EP1018289A1 EP98931259A EP98931259A EP1018289A1 EP 1018289 A1 EP1018289 A1 EP 1018289A1 EP 98931259 A EP98931259 A EP 98931259A EP 98931259 A EP98931259 A EP 98931259A EP 1018289 A1 EP1018289 A1 EP 1018289A1
Authority
EP
European Patent Office
Prior art keywords
transformer
voltage
windings
electronic ballast
ballast system
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Withdrawn
Application number
EP98931259A
Other languages
German (de)
English (en)
Inventor
Roger Siao
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
SIAO, ROGER
SIAO, SUSAN
VRIONIS, NICKOLAS G.
Original Assignee
Individual
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority claimed from US08/899,184 external-priority patent/US6005355A/en
Application filed by Individual filed Critical Individual
Publication of EP1018289A1 publication Critical patent/EP1018289A1/fr
Withdrawn legal-status Critical Current

Links

Classifications

    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
    • H05B41/295—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices and specially adapted for lamps with preheating electrodes, e.g. for fluorescent lamps

Definitions

  • This invention pertains generally to fluorescent lighting and, more particularly, to an electronic ballast system for fluorescent lamps.
  • ballasts have been provided for use with fluorescent lamps. Examples of such ballasts are found in U.S. Patents 4,245,178 and 4,631 ,449.
  • Electronic ballasts typically operate at frequencies on the order of 10 KHz to 100 KHz, and are designed to provide high circuit efficiency, high reliability, and low cost. While the physical size and weight of ballasts are dependent upon operating frequency, with higher frequencies permitting ballasts to be smaller in size and lighter in weight, reductions in size and weight have not been easy to achieve.
  • Another object of the invention is to provide a ballast system of the above character which overcomes the limitations and disadvantages of the prior art.
  • an electronic ballast system which has a transformer with primary and secondary windings, a power oscillator connected to the primary winding for operation at a predetermined frequency in the range of 10 KHz to 5 MHz, and a ballasting network connected to the secondary winding and adapted for connection to the fluorescent lamp, with the ballasting network being resonant at a frequency within about ⁇ 10 percent of the predetermined frequency when connected to the lamp.
  • the resonant frequency of the ballasting network remains the same regardless of the number of lamps connected to it. Due to the resonance in the ballasting network, only resistive loading transformation occurs in the power transformer.
  • FIG. 1 is a circuit diagram of one embodiment of an electronic ballast system according to the invention.
  • Figure 2 is an AC equivalent circuit of the primary section of the system of Figure 1.
  • Figure 3 is a set of waveform diagrams of the voltages at certain points in the system of Figure 1.
  • Figures 4 - 7 are circuit diagrams of other embodiments of ballasting networks for use in the system of Figure 1.
  • FIGS 8 - 9 are circuit diagrams of additional embodiments of an electronic ballast system according to the invention.
  • Figure 10 is an AC equivalent circuit of the primary section of the embodiment of Figure 9.
  • FIGS 11 - 13 are circuit diagrams of additional embodiments of an electronic ballast system according to the invention.
  • Figures 14a and 14b are AC equivalent circuits of the primary section of the embodiment of Figure 13.
  • Figure 15 is a set of waveform diagrams of the voltages at certain points in the embodiment of Figure 13.
  • FIG 16 is a circuit diagram of another embodiment of an electronic ballast system according to the invention.
  • Figures 17a and 17b are AC equivalent circuits of the primary section of the embodiment of Figure 16.
  • FIG. 18 is a circuit diagram of another embodiment of an electronic ballast system according to the invention.
  • Figures 19a and 19b are AC equivalent circuits of the primary section of the embodiment of Figure 18.
  • Figure 20 is a set of waveform diagrams of the voltages at certain points in the embodiment of Figure 18.
  • Figures 21 and 22 are circuit diagrams of additional embodiments of an electronic ballast system according to the invention.
  • FIG. 23 is a circuit diagram of another embodiment of an electronic ballast system according to the invention.
  • Figures 24a and 24b are AC equivalent circuits of the primary section of the embodiment of Figure 23.
  • Figure 25 is a set of waveform diagrams of the voltages at certain points in the embodiment of Figure 23.
  • FIG. 26 is a circuit diagram of another embodiment of an electronic ballast system according to the invention.
  • Figures 27a and 27b are AC equivalent circuits of the primary section of the embodiment of Figure 18.
  • FIG. 28 is a circuit diagram of another embodiment of an electronic ballast system according to the invention.
  • Figure 29 is a set of waveform diagrams of the voltages and currents at certain points in the embodiment of Figure 28.
  • FIG. 30 is a circuit diagram of another embodiment of an electronic ballast system according to the invention.
  • Figure 31 is a cross-sectional view, somewhat schematic, of one embodiment of a transformer for use in the ballast system of the preceding figures.
  • Figure 32 is a cross-sectional view taken along line 32-32 of Figure 31 , with a portion of the core structure removed for clarity of illustration.
  • Figure 33 is a cross-sectional view of another embodiment of a shield for use in the transformer of Figure 31.
  • Figures 34 and 35 are isometric views of additional embodiments of transformers for use in the ballast system of Figures 1 - 30.
  • the ballast system includes a transformer 11 , a power oscillator 12 connected to the primary of the transformer, and a ballasting network 13 connected to the secondary of the transformer.
  • the oscillator is doubly tuned, with the frequency determining components of the oscillator and the ballasting network being tuned to substantially the same frequency.
  • Operating power is provided by a power supply 16 which is connected to a standard ⁇ e.g., 120 volt, 60 cycle) AC source.
  • the power supply includes full- wave bridge rectifier 17 which is connected to the source through a low-pass LC filter consisting of an inductor 18 and a capacitor 19.
  • a varistor 21 is connected across the source to absorb transient disturbances from the power lines, and a filter capacitor 22 and an RF bypass capacitor 23 are connected to the output of the rectifier bridge.
  • the supply provides a DC output voltage
  • Transformer 11 has primary windings 26, 27, feedback or drive windings 28, 29, and a secondary winding 31.
  • the two primary windings are connected together in series at a common node or center tap 32, and the two drive windings are connected together at a common node or center tap 33.
  • Power oscillator 12 is a doubly tuned, current switched, transformer coupled,
  • Class-D power oscillator that includes a pair of switching transistors 36, 37, which in the embodiment illustrated are high power MOSFETs. It will be understood, however, that the invention is not limited to a particular type of switching device, and that other switching devices such as bipolar junction transistors (BJTs) or junction field effect transistors (JFETs) can be used.
  • BJTs bipolar junction transistors
  • JFETs junction field effect transistors
  • the drains of the switching transistors are connected to the outer ends of primary windings 26, 27, and the gates are connected to the outer ends of drive windings 28, 29.
  • the sources of the transistors are connected together at a common source node 39.
  • Capacitors 41 , 42 are connected between the outer ends of primary windings 26, 27, and the junction of the two capacitors is connected to ground. These capacitors resonate with the total inductance of the primary windings to determine the operating frequency of the oscillator, and they also serve to protect the switching transistors by providing a low AC impedance between the drains of the transistors and ground when subjected to high frequency transient signals or voltage spikes.
  • the presence of the capacitors also makes the coefficient of coupling of the transformer less critical, and avoids the need for an extremely high coupling factor ⁇ e.g., a factor greater than .98).
  • the values of the inductances and the capacitances are chosen to provide resonance at a frequency in the range 10 KHz to 5 MHz.
  • the voltages at the outer ends of drive windings 28, 29 are in phase with the voltages at the outer ends of primary windings 26, 27, which provides regenerative or positive feedback to establish and maintain self-oscillation in the circuit.
  • the supply voltage V ⁇ JQ is applied to the center tap or common node of the primary windings through an RF choke 43 which prevents AC fluctuations in the switching current of the oscillator.
  • the switching transistors are self-biased, and source degeneration is employed to ensure low power loss and high DC to AC conversion efficiency.
  • a biasing voltage of substantially constant magnitude is provided by a voltage regulator consisting of a Zener diode 46 and a dropping resistor 47 connected between the output of the power supply and ground.
  • the voltage developed across the Zener diode is applied to the center tap or common node of drive windings 28, 29 by a low pass filter consisting of a resistor 48 and a capacitor 49.
  • the filter isolates the Zener diode from AC voltages in the drive windings.
  • An AC bypass capacitor 50 is connected between the common node of the drive windings and ground.
  • a current sensing resistor 51 and a pair of parallel connected, back-to-back diodes 52, 53 are connected in series between the common source node 39 and ground to form a degenerative or negative feedback network which controls the gain and DC bias currents, and enhances the stability of the circuit.
  • the gain of the circuit is decreased by the source voltage feedback, and any abnormal swing in the voltage at the drains of the transistors is automatically reduced, as is any abnormal current flowing through the transistors.
  • a grounding capacitor 54 is connected between the lower end of secondary winding 31 and the metal enclosure of the system to provide an AC path for EMI energy which radiates from electronic components and couples to the enclosure to return to circuit ground.
  • the enclosure is connected to an earth ground, and the grounding capacitor also provides an AC current return path for the lamps during the capacitive discharge mode.
  • ballasting network 13 is specifically intended for use with instant-start fluorescent lamps 56, 57 which are mounted in sockets 58, 59. There are two internally connected connector pins at each end of the lamps, and the sockets have terminals 61-64 and 66-69 for contact with the pins.
  • the ballasting network comprises a differential transformer 71 which has tightly coupled windings 72, 73 with a coefficient of coupling near unity. Each of those windings is connected electrically in series with an inductor 74, with opposite phase ends of the windings being connected to one end of the inductor.
  • the other end of the inductor is connected to the upper end of secondary winding 31 of transformer 11 , and the remaining ends of windings 72, 73 are connected to terminals 61 , 66 at the lower ends of the lamp sockets.
  • the other terminals 62, 67 at the lower ends of the sockets are connected to the lower end of winding 31.
  • Capacitors 76, 77 are connected between the upper end of secondary winding 31 and terminals 63, 68 at the upper ends of the lamp sockets. No connections are made to socket terminals 64, 69 in this embodiment.
  • Ballasting network 13 is thus a tank circuit in which capacitors 76, 77 are connected electrically in parallel with inductor 74 when the lamps are installed in their sockets.
  • the inductances of the two windings 72, 73 of the differential transformer are equal to each other and to the inductance of series inductor 74.
  • Capacitors 76, 77 are also equal in value.
  • the inductances of the two windings of the differential transformer cancel, and the resonant frequency of the tank circuit is determined by L in parallel with 2C, where L is the inductance of the series inductor, and 2C is the capacitance of capacitors 76, 77 in parallel.
  • the resonant frequency of network 13 is the same with either or both of the lamps installed. That frequency is chosen to be substantially equal to the resonant frequency of the circuit on the primary side of transformer 11. The two frequencies do not have to be exactly equal, and the system will work quite well if they are within about ⁇ 10 percent of each other. With resonant circuits on both sides of the transformer, the power oscillator can be said to be doubly tuned.
  • the double tuned power oscillator becomes a single tuned oscillator, and the oscillator frequency is determined solely by the resonant tank circuit on the primary side of the transformer.
  • the natural frequency of the power oscillator remains constant regardless of the number of lamps which are connected.
  • a plurality of resonant ballasting networks can be connected to the transformer secondary to drive any desired number of lamps, and those networks will all resonate at the frequency for which they are designed regardless of the number of lamps connected. Furthermore, the power dissipated by each lamp which is connected remains the same whether one lamp or more is/are connected.
  • FIG 2 shows an AC equivalent circuit of the primary section of the embodiment of Figure 1 , with transistor 37 (Q2) conducting and transistor 36 (Q1) off.
  • R Q -1 represents the AC impedance of the diode 52 (D1) which is biased in the forward direction when transistor Q2 is conducting.
  • the primary current flows around a loop comprising primary windings 26, 27, capacitor 41 (C1), resistor 51 (R3), the AC impedance R ⁇ of diode 52, diode 53 (D2), and transistor 37 (Q2).
  • transistor Q1 When transistor Q1 is conducting, the current flows around a loop comprising the primary windings, capacitor 42 (C2), resistor 51 (R3), the AC impedance R ⁇ ji of diode 52, diode 53 (D2), and transistor 36 (Q1).
  • the AC impedance R ⁇ 0 f diode 52 varies inversely with the current through it, and is approximately equal to 26mV/lDD- where IQD is tne loaded or unloaded DC current of the amplifier.
  • the gain of the amplifier is proportional to the load impedance Z_ and inversely proportional to the AC impedance of diode 52 (D1 ). For values of g m such that 1/g m « Rdi + R 3 ,
  • g m is the transconductance of the transistor
  • R3 is the resistance of the resistor 51 in series with the diode.
  • resistor 51 and diodes 52, 53 are much smaller than those of primary windings 26, 27 and capacitors 41 , 42, and resistor 51 and diodes 52, 53 thus have little effect on the natural frequency of the primary section.
  • the natural frequency of the primary circuit is determined by the relationship
  • l_ ⁇ 1 is the total inductance of primary windings 26, 27 and C is the capacitance of capacitor 41 (C1) or capacitor 42 (C2), depending upon which transistor is conducting, with C1 and C2 typically being equal in value.
  • the relationship between the voltage VQJ at the center tap 32 of the primary winding of the power transformer and the drain-source voltages VQ and VQ2 of transistors 36, 37 is illustrated in Figure 3.
  • the center tap voltage is full- wave rectified and rises above and below the supply voltage VQQ with its peak-to-peak voltage equal to one-half of the peak voltages across the transistors.
  • the drain-source voltages are half-wave rectified and are 180° out of phase with each other.
  • the peak magnitude of the drain-source voltages is ⁇ • V ⁇ JD-
  • FIG 4 illustrates another embodiment of a ballasting network for use with instant-start lamps wherein the resonant frequency and the power dissipated remain the same with one lamp or two.
  • This network is similar to the network of Figure 1 except the differential transformer 71 and series inductor 74 are replaced by two inductors 78, 79 of equal inductance.
  • the network consists of two identical parallel tanks, each of which is tuned to substantially the same frequency as the tank circuit on the primary side of the transformer. With one lamp removed, the network consists of a single tank circuit tuned to that frequency. With both lamps removed, the oscillator frequency is determined solely by the primary tank circuit. In all three cases, the frequency remains the same.
  • inductors 78, 79 of equal inductance are once again connected in series with instant-start lamps.
  • a single resonating capacitor 81 is connected in parallel with the inductors and lamps.
  • the resonant frequency of the combined parallel network is made equal to the resonant frequency of the primary circuit so that when both lamps are connected, the tank circuits on both sides of the transformer will be tuned to the same frequency.
  • the two resonant frequencies will be mismatched by a factor of .707.
  • the ballasting network of Figure 6 is similar to the network of Figure 5 except it has capacitors 82, 83 of equal value in series with the instant-start lamps and an inductor 84 in parallel with the capacitors and lamps.
  • the resonant frequency of the combined parallel network is made equal to the resonant frequency of the primary circuit so that when both lamps are connected, the tank circuits on both sides of the transformer will be tuned to the same frequency. When one of the lamps is removed, the two resonant frequencies will once again be mismatched by a factor of .707.
  • the ballasting network of Figure 7 is similar to that shown in Figure 1 except the bottom end of the secondary winding of transformer 11 is connected to terminals 64, 69 at the upper ends of the tubes, and terminals 62, 67 are left unconnected.
  • This network operates in a manner similar to the network of Figure 1 , and the frequency characteristics of the two are the same.
  • FIG 8 illustrates a system for use with rapid-start fluorescent lamps 86, 87.
  • This embodiment is similar to that of Figure 1 , and like reference numerals designate corresponding elements in the two embodiments.
  • transformer 11 has two filament windings 89, 91 which are connected to the cathode electrodes in the lamps which are connected to the series capacitors 76, 77 in the ballasting network.
  • two of the cathode electrodes are energized by the filament windings, and the other two are energized by the circulating current flowing through the series inductor 74.
  • the addition of the filament windings does not affect the resonant frequency of the ballasting network, and that frequency remains the same whether one or two lamps are connected.
  • Ballasting networks similar to those shown in Figures 4 - 7 can also be used with rapid-start lamps in the system of Figure 6, with filament windings 89, 91 powering the cathode electrodes at one end of the lamps.
  • a third filament winding 92 (shown in Figure 8) is utilized for energizing the cathode electrodes at the other end of the lamps.
  • Figure 9 illustrates another embodiment which is similar to the embodiment of Figure 1 except the junction of capacitors 41 , 42 is connected to the common source node 39 of the switching transistors 36, 37, rather than being connected to ground.
  • this embodiment only a single diode 52 is required rather than the back-to-back pair of Figure 1.
  • the frequency characteristics of this embodiment are identical to those of Figure 1 , and this embodiment can be utilized with any of the ballasting networks shown in Figures 4 - 7, either for instant-start lamps or for rapid-start lamps.
  • FIG 10 shows an AC equivalent circuit of the primary section of the embodiment of Figure 9, with transistor 37 (Q2) conducting and transistor 36 (Q1) off.
  • the loop current does not flow through the diode, and with transistor 37 (Q2) conducting, the loop through which the primary current flows comprises primary windings 26, 27, capacitor 41 (C1) and transistor 37 (Q2).
  • the loop With transistor 36 (Q1) conducting, the loop comprises the primary windings, capacitor 42 (C2) and transistor 36 (Q1).
  • FIG 11 illustrates a system for use with an induction discharge lamp 93.
  • This system is similar to the embodiment of Figure 1 , and like reference numerals designate corresponding elements in the two.
  • the ballasting network consists of two capacitors 94, 96 of equal value connected in series across the secondary of transformer 11 , and an inductor 97 which is connected in parallel with the two capacitors.
  • An AC grounding capacitor 98 is connected between the junction of the capacitors 94, 96 and an earth ground.
  • the tank circuit formed by capacitors 94, 96 and inductor 97 is tuned to substantially the same frequency as the tank circuit on the primary side of the transformer, and the inductor radiates an AC magnetic field which couples to the lamp.
  • the embodiment of Figure 12 includes a power supply 101 which is generally similar to power supply 16 in the embodiment of Figure 1 , and like reference numerals designate corresponding elements in the two embodiments.
  • a differential transformer 102 replaces inductor 18, and a thermostat 103 is connected in series with the fuse.
  • a high frequency bypass capacitor 104 is connected between one of the power lines and the chassis ground. That capacitor is shown as being connected to the neutral conductor, but it can be connected to the line conductor instead, if desired.
  • the amplifier section of the embodiment of Figure 12 is similar to that of Figure 9, and like reference numerals designate corresponding elements in the two embodiments.
  • damping resistors 106, 107 are connected in series with capacitors 41 , 42. These resistors absorb non-linear high frequency noise and thereby enhance the stability of the amplifier during the start-up mode.
  • the combined impedances of resistor 51 and diodes 52, 53 provide sufficient damping for the primary resonant tank circuit, and additional damping resistors are not required.
  • the embodiment of Figure 12 also differs from that of Figure 9 in that capacitor 54 is moved to the primary side of transformer 11 and connected between the primary circuit ground and an earth ground. Having this capacitor on the primary side of the transformer results in a significant reduction in the amount electromagnetic interference (EMI) which is coupled to the power lines.
  • EMI electromagnetic interference
  • the ballasting network 109 in the embodiment of Figure 12 is intended for use with rapid-start lamps operating in an instant-start mode.
  • This network differs from the ballasting network in the embodiment of Figure 1 in that lamp terminals 61 , 66 are connected directly to the lower end of the secondary winding 31 of transformer 11 , and terminals 62, 67 are left unconnected. Since the terminals of rapid-start lamps are not connected together internally like the terminals of instant-start lamps, the lower ends of the two windings 72, 73 of differential transformer 71 are connected directly to the lower end of winding 31 , rather than being connected through the terminals of the lamps as they are in the embodiment of Figure 1. With either one lamp, two lamps or no lamps connected in the circuit, ballasting network 109 has the same electrical characteristics as the network of Figure 1.
  • the embodiment of Figure 13 has a primary system which is similar to that in the embodiment of Figure 1 and a ballast network similar to network 109 in the embodiment of Figure 12, with like reference numerals once again designating corresponding elements in the various embodiments.
  • the embodiment of Figure 13 differs from the others, however, in that RF choke 43 is connected between resistor 51 and the circuit ground, and the lower end of capacitor 50 is connected to the junction of the resistor and the choke.
  • the primary windings 26, 27 are also tightly coupled together to enhance waveshape symmetry and to reduce the leakage flux field between them.
  • Cp is the parasitic capacitance of all of the power transformer windings combined, including the secondary winding 31.
  • the choke voltage V C h is inverted, as compared with the embodiment of Figure 1 , and that voltage is a negative-going full- wave rectified sinusoid which is approximately equal to the voltage across the capacitor C1 or C2 in the active current loop.
  • the drain voltages V ⁇ i and V j 2 of transistors 36 (Q1) and 37 (Q2) are sinusoidal and have a peak-to- peak value of ⁇ • Vpo-
  • and VQ2) is equal to the total voltage across the two primary windings.
  • This embodiment has a significant advantage in that the natural frequency is determined primarily by the series combination of capacitors 41 (C1) and 42 (C2) so that the two capacitors do not have to be made equal in value in order to have waveform symmetry across the primary windings.
  • the high voltage across the primary windings and at the two drain nodes is sinusoidal, which minimizes harmonics and RF radiation to the environment.
  • the embodiment shown in Figure 16 is similar to the embodiment of Figure 13 except tuning capacitors 41 , 42 are connected directly between the drains and the sources of switching transistors 36, 37 in a manner similar to the embodiment of Figure 9, and only a single diode 51 is connected between resistor 53 and the sources of the transistors.
  • the primary current I flows around a loop comprising primary windings 26, 27, the conducting transistor (Q1 or Q2) and the capacitor (C1 or C2) connected across the nonconducting transistor.
  • a parasitic current l p flows around a loop comprising the primary windings the output capacitance (CQ-
  • the effect of the output capacitances CQ ⁇ , CQ2 is relatively small and can be ignored in determining the natural frequency of the primary circuit in accordance with the relationship
  • 1 is the total inductance of the primary windings
  • C Cp + C1
  • C1 C2
  • Cp the parasitic capacitance of all of the power transformer windings combined, including the secondary winding 31.
  • Figure 18 illustrates an embodiment in which two amplifier circuits 111, 112 are stacked in the primary system to provide a power oscillator having a split power supply in which each switching device sees only one-half of the rectified DC voltage.
  • This is advantageous because it enables the system to operate on higher supply voltages ⁇ e.g., 220, 277 or 347 volts AC) without relatively expensive switching transistors with higher breakdown voltages.
  • a 120 VAC system requires 600 volt transistors
  • a 277 VAC system requires 1300 volt transistors, which are substantially more expensive and not always available.
  • One of the stacked circuits is connected between voltage nodes 113 and 114, and the other is connected between voltage node 114 and ground node 116.
  • the voltage at node 113 is the supply voltage Vpp. and tne voltage at node 114 is approximately equal to one-half of the supply voltage.
  • the ground node is connected to the circuit ground.
  • the two stacked amplifier circuits are similar to the amplifier circuit in the embodiment of Figure 1.
  • These circuits include a choke transformer 117 which has a pair of tightly coupled, in-phase windings 117a, 117b of equal inductance.
  • This transformer provides an RF choke impedance for each of the stacked amplifiers and a unity impedance transformation between them.
  • the primary windings 26, 27 of the power transformer are separated but coupled tightly to each other, rather than being connected together to form a center tap as they are in the embodiment of Figure 1.
  • choke winding 117a and primary winding 26 are connected between voltage node 113 (Vr j p) ar >d the drain of a MOSFET switching transistor 119.
  • Back-to-back diodes 121 , 122 and a resistor 123 are connected between the source of this transistor and voltage node 114 (V a ).
  • a capacitor 124 is connected between the drain of the transistor and the ground node.
  • choke winding 117b and primary winding 27 are connected between voltage node 114 (V a ) and the drain of a MOSFET switching transistor 126.
  • Back-to-back diodes 127, 128 and a resistor 129 are connected between the source of transistor 126 and the ground node, and capacitor 131 is connected between the drain of the transistor and the ground node. Resistor 129 and capacitor 131 are equal in value to resistor 123 and capacitor 124.
  • Biasing voltages for the switching transistors are developed across resistors 133, 134 and a Zener diode 136 which are connected in series between voltage node 113 and ground node 116.
  • the resistors serve as a voltage divider which provides a biasing voltage V ) -, for circuit 111 , with the Zener voltage V z being applied to circuit 112. With resistors of equal value, voltage V
  • Voltage V ⁇ is applied to one end of drive winding 28 by a low-pass filter consisting of a resistor 138 and a capacitor 139, and the other end of the drive winding is connected to the gate of transistor 119.
  • Biasing voltage V z is applied to one end of drive winding 29 by a low-pass filter consisting of a resistor 141 and a capacitor 142, and the other end of this drive winding is connected to the gate of transistor 126.
  • AC bypass capacitors 143, 144 are connected between the outputs of the low-pass filters and low voltage nodes 1 14, 116, respectively.
  • the Zener diode In addition to providing to providing substantially constant biasing voltages at the gates of the transistors under steady-state conditions, the Zener diode also serves to stabilize the power oscillator during start-up by providing a soft start.
  • the current vs. voltage (l-V) characteristic of a Zener diode is logarithmic at low current levels, and during start-up, the Zener diode operates in the logarithmic region to provide a voltage which is slightly greater than the voltage required to turn on the transistors.
  • the gate voltages rise slowly and prevent a fast rise of the drain currents. Since the gain of the amplifiers depends on the drain currents, the amplitude of the oscillation increases logarithmically.
  • a resistor 146 is connected in parallel with the Zener diode to reduce the effect of variations in the current- voltage characteristics of Zener diodes from different manufacturers.
  • the resistor desensitizes the system to changes in the dynamic impedance of the Zener diode which varies inversely with the reverse current through the diode.
  • AC bypass capacitors 148, 149 are connected between voltage nodes 113, 114 and the circuit ground.
  • bipolar junction transistors for example, the values of resistors 138, 141 are selected to make the base currents of the transistors much smaller than the DC currents through resistors 133, 134.
  • Diodes should also be connected between the collectors and emitters of the bipolar transistors to provide reverse current paths across the transistors during their OFF states. The anodes of the diodes are connected to the emitters, and the cathodes are connected to the collectors.
  • the AC gains of the two amplifier circuits are substantially identical in the linear operating region.
  • choke transformer 117 acts as a dead short between the primary windings 26, 27, and the active amplifier (amplifier 111 in these figures) is transformed from one choke terminal to the other.
  • the voltage waveforms of the two amplifier circuits are as shown in Figure 20, where V c h is the voltage of either choke winding 117a or choke winding 117b, and VQ-J and VQ2 are the drain-to- source voltages of transistors 119, 126.
  • the natural frequency is given by the relationship
  • C1 and C2 are the capacitances of capacitors 124 and 131
  • Cp is the parasitic capacitance of all of the power transformer windings combined, including the secondary winding 31.
  • the embodiment of Figure 21 is similar to the embodiment of Figure 18, except the lower end of tuning capacitor 124 is connected to voltage node 114 (V a ), rather than to the ground node, and the lower end of the AC bypass capacitor 148 is connected to voltage node 114, rather than to the ground node.
  • the embodiment of Figure 22 is also generally similar to the embodiment of Figure 18.
  • the tuning capacitors 124, 131 are connected in series with resistors 151 , 152 between the drains and sources of the transistors.
  • Figure 23 illustrates an embodiment similar to the embodiment of Figure 18 but with amplifier circuits 153, 154 of the type shown in Figure 13.
  • choke winding 117a is connected between resistor 123 and voltage node
  • bypass capacitor 148 is connected to voltage node 114 (V a ), rather than to the circuit ground, and grounding capacitor 54 is connected directly between the output of the power supply (voltage node 113) and the earth ground.
  • Transistor 126 (Q2) is shown in the conducting state in the AC equivalent circuits of Figures 24a and 24b. In this state, the main primary current I flows around a loop comprising transformer windings 26, 27, capacitor 124 (C1), capacitor 131 (C2), and the parasitic capacitance Cp. In this embodiment, the natural frequency is given by the relationship
  • capacitors 124 and 131 do not have to have equal values.
  • V fj -j and V ⁇ being the drain voltages of transistors 119 (Q1) and 126 (Q2)
  • V C h being the voltage on either choke winding 117a or choke winding 117b
  • VQ-J and VQ2 being the drain-to-source voltages of the transistors.
  • Figure 26 illustrates another embodiment in which the choke windings 117a and 117b are positioned between the source resistors 123, 129 and lower voltage nodes 114, 116 in the stacked circuits.
  • This embodiment differs from the embodiment of Figure 23 in that the tuning capacitors 124, 131 are connected between the drains and the sources of the transistors.
  • choke transformer 117 once again provides a short circuit for the primary current I, and the summation of the choke winding voltages is zero.
  • the natural frequency of this embodiment is
  • Figure 28 illustrates an embodiment which is similar to that shown in Figure 21 but has stacked amplifier circuits 156, 157 with the choke windings connected at different points in the two circuits.
  • choke winding 117a is connected between resistor 123 and voltage node 114 (V a ) in the source circuit of transistor 119, and the lower end of capacitor 143 is connected to the junction of the choke winding and the resistor.
  • choke winding 117b is connected between voltage node 114 (V a ) and primary winding 27 in the drain circuit of transistor 126.
  • an AC circulating current loop is formed during the ON/OFF cycles of the switching transistors.
  • the loop comprises primary winding 26, bypass capacitor 148, choke winding 117b, primary winding 27, tuning capacitor 131 , the output capacitance CQ-
  • the loop comprises primary winding 27, choke winding 117b, bypass capacitor 148, primary winding 26, tuning capacitor 124, the output capacitance CQ2 of transistor 119, diode 122, resistor 123, choke winding 117a, bypass capacitor 149, resistor 129, diode 127, the AC impedance R, j 2 of diode 128, and transistor 126.
  • Capacitors 124, 131 are matched in order to maintain symmetry of the sine wave across the primary windings of the power transformer, and the natural frequency is
  • C1 + Cp + CQI or C1 C2 + Cp + CQ2, depending upon which transistor is conducting.
  • C1 and C2 are the capacitances of capacitors 124 and 131
  • and CQ2 are the output capacitances of transistors 119 and 126
  • Cp is the parasitic capacitance of the transformer windings.
  • and VQ2 are the drain-to-source voltages of transistors 119 and 126
  • IR3 is the current through resistor 129
  • V c h2 is the voltage at the top of choke winding 117a
  • l cn 2 is the current through choke winding 117a.
  • FIG 30 illustrates another embodiment in which the choke windings are connected at different points in two stacked amplifier circuits.
  • choke winding 117a is connected between voltage node 113 (VQQ) and the upper end of primary winding 26, and in circuit 159, choke winding 117b is connected between resistor 129 and ground node 116.
  • the choke transformer With the windings connected directly to the +/- terminals of the DC supply, the choke transformer also functions as a high frequency noise suppressing, differential mode transformer which prevents RF noise generated during the transistor switching from being transmitted to the power lines.
  • the AC circulating current loops and the natural frequency of this embodiment are similar to those of the embodiment of Figure 28.
  • filter capacitor 22 it is also possible to replace filter capacitor 22 with two capacitors of equal value connected in series between the output of the power supply and ground, with the junction of the two capacitors connected to voltage node 114 (V a ), i.e. one capacitor connected between voltage nodes 113, 114, and the other connected between voltage node 114 and ground node 116. Since each of those capacitors would have to handle only one-half of the supply voltage, their capacity can be made larger than that of capacitor 22. That provides a higher AC ⁇ e.g. 120 Hz) ripple current capability, which is important in extending the life of an electronic ballast.
  • Figures 31 - 35 illustrate a transformer construction which is particularly suitable for use in the ballast system of the invention.
  • This construction substantially reduces the radiation of high frequency noise generated by magnetic flux switching in the transformer.
  • the transformer has primary windings 26, 27 and gate drive windings 28, 29 wound on a bobbin 171.
  • Secondary winding 31 is wound over the other windings, with layers of insulation 172 around the windings.
  • An open loop metal shield 173 is positioned between the primary and secondary windings, and core pieces 174 are assembled about the winding structure to form a magnetic core.
  • Shield 173 is fabricated of an electrically conductive metal such as copper or aluminum, and is connected to the circuit ground for radio frequency E-field suppression. It encircles the primary windings and is in the form of an open loop with a gap 176 between confronting ends of the metal which forms the shield. In order to avoid high voltage arcing, the width of the shield is made equal to or less than the width of the windings, with narrower shields being used for higher voltages on the windings.
  • One material which is economic and easy to use for the shield is an aluminum tape or a copper tape.
  • the shield 173 has overlapping end portions 177, 178, with a layer of insulation ⁇ e.g., electrically insulative tape) 179 positioned between the overlapping ends to maintain an open loop configuration.
  • a layer of insulation ⁇ e.g., electrically insulative tape e.g., electrically insulative tape
  • this shield is illustrated as having a generally circular cross-section, it can have any other configuration which is suitable for the transformer in which it is used.
  • Figure 34 illustrates an embodiment in which the transformer has windings 181 on the central leg of a magnetic core 182, with leads 183 extending from one side of the winding layers.
  • a shield 184 is wrapped externally about the core and the windings, with the end portions of the tape being spaced apart to form a gap 186 at one end of the core.
  • the shield is fabricated of an electrically conductive material ⁇ e.g., aluminum or copper tape) and is connected to the circuit ground to suppress radio frequency E-field radiation.
  • the embodiment of Figure 35 is similar to the embodiment of Figure 34, with the shield being extended as indicated at 187 to cover the edges of the windings which project from the magnetic core.
  • the invention has a number of important features and advantages. It provides a simple, low cost, self-starting oscillator circuit which employs self- biased switching devices with emitter or source degeneration for starting and maintaining oscillation with low currents and low Q resonant conditions.
  • the switching devices and power transformer are protected against damage from large voltage spikes and other transient disturbances, and sensitivity to the coefficient of coupling between the primary and secondary windings of the power transformer is also reduced.
  • the leakage flux of a loosely coupled transformer can produce large voltage spikes across the switching devices or across any other semiconductors located within the path.
  • the leakage energy is recirculated through the transformer primary and is absorbed by the circuit loads.
  • the combination of the capacitors, the series RF choke and the inductance of the primary winding also protects the switching transistors against large transient disturbances which can occur in the AC power lines.
  • the oscillator operates at the same resonant frequency with one or two lamps connected, as well as with both lamps disconnected. Because of the resonant operating condition, the resultant impedance of the ballasting network and the lamps is purely resistive, and this permits components of smaller size and lower cost to be used.
  • the double tuning of the oscillator has another significant advantage in that the output of the secondary winding of the power transformer acts as a constant voltage and frequency source, which is an important factor in delivering a fixed power to each lamp.

Landscapes

  • Circuit Arrangements For Discharge Lamps (AREA)

Abstract

Système de ballast électronique pour lampes fluorescentes (56, 57). Un oscillateur (12) de puissance est connecté à l'enroulement primaire d'un transformateur (T1) de puissance de manière à fonctionner à une fréquence prédéterminée de l'ordre de 10 KHz à 5 MHz, et un réseau de ballastage est connecté au second enroulement (31) du transformateur et à une ou plusieurs des lampes. Le réseau de ballastage est résonant à une fréquence supérieure ou inférieure d'environ 10 % à la fréquence prédéterminée et dans certains modes de réalisation, la fréquence de résonance du réseau de ballastage reste la même quel que soit le nombre de lampes connectées au réseau.
EP98931259A 1997-07-23 1998-06-10 Systeme de ballast electronique Withdrawn EP1018289A1 (fr)

Applications Claiming Priority (3)

Application Number Priority Date Filing Date Title
US899184 1997-07-23
US08/899,184 US6005355A (en) 1996-12-27 1997-07-23 Electronic ballast system for fluorescent lamps
PCT/US1998/012321 WO1999005893A1 (fr) 1997-07-23 1998-06-10 Systeme de ballast electronique

Publications (1)

Publication Number Publication Date
EP1018289A1 true EP1018289A1 (fr) 2000-07-12

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EP98931259A Withdrawn EP1018289A1 (fr) 1997-07-23 1998-06-10 Systeme de ballast electronique

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EP (1) EP1018289A1 (fr)
CA (1) CA2297255A1 (fr)
WO (1) WO1999005893A1 (fr)

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Publication number Priority date Publication date Assignee Title
AU2020209726B2 (en) 2019-01-16 2023-01-19 Rhythmlink International, Llc Neurological monitoring cable for magnetic resonance environments

Family Cites Families (2)

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Publication number Priority date Publication date Assignee Title
US3963975A (en) * 1975-03-05 1976-06-15 General Electric Company Electromagnetically shielded electrical power supply with reduced common mode electromagnetic interference output
US4873471A (en) * 1986-03-28 1989-10-10 Thomas Industries Inc. High frequency ballast for gaseous discharge lamps

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
See references of WO9905893A1 *

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WO1999005893A1 (fr) 1999-02-04
CA2297255A1 (fr) 1999-02-04

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