EP0729223B1 - Offsetspannungskompensationsschaltung - Google Patents

Offsetspannungskompensationsschaltung Download PDF

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Publication number
EP0729223B1
EP0729223B1 EP96300822A EP96300822A EP0729223B1 EP 0729223 B1 EP0729223 B1 EP 0729223B1 EP 96300822 A EP96300822 A EP 96300822A EP 96300822 A EP96300822 A EP 96300822A EP 0729223 B1 EP0729223 B1 EP 0729223B1
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European Patent Office
Prior art keywords
amplifier
input
voltage
high gain
circuit
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EP96300822A
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English (en)
French (fr)
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EP0729223A2 (de
EP0729223A3 (de
Inventor
Keith L. Jones
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Microsemi Semiconductor Ltd
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Zarlink Semiconductor Ltd
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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/30—Modifications of amplifiers to reduce influence of variations of temperature or supply voltage or other physical parameters
    • H03F1/303—Modifications of amplifiers to reduce influence of variations of temperature or supply voltage or other physical parameters using a switching device

Definitions

  • the invention relates to a voltage offset compensation circuit having a particular, but not necessarily an exclusive, application in either a sampled-data integrator, or a continuous-time circuit.
  • Compensation techniques for reducing input voltage offset in electronic circuits are known (US-A-4 546 324, US-A-4 962 325) and find application in sampled-data circuits and continuous-time circuits. Compensation techniques which effect a reduction in input voltage offset have the added advantage of effecting a reduction in low frequency noise. Some of the known techniques also compensate for the finite gain of the amplifier which otherwise manifests itself as a gain error.
  • the circuit can be offset-compensated during the other phase of the clock signal and the known techniques for effecting such compensation give good results.
  • an offset-compensated output is required during both phases of the clock signal. The known compensation techniques are, therefore, unsuitable for such applications.
  • the Cohn compensating arrangement involves the use of an auxiliary amplifier in a feed-forward approach, but this arrangement requires a large, and therefore external, storage capacitors to avoid circuit stability problems. Also, the recovery time of such a circuit is very long. Since, with this arrangement, the voltage offset cancellation loop includes part of the main amplifier, the design/performance of the main amplifier is adversely affected by the need to maintain circuit stability.
  • the shortcomings of known offset cancellation techniques include the need to use external components, problems associated with clock related noise, performance compromises in the amplifier design and constraints in application of the known techniques. Furthermore, the fact that the generally preferred chopper-stabilisation technique requires post-filtering to remove signal error components, places demands on the amplifier by requiring its output to slew from one polarity of signal to another between clock cycles, and causes glitches because the amplifier is momentarily out of circuit.
  • the invention provides a voltage offset compensation circuit for a high gain amplifier having a fixed input voltage offset, the said circuit including sample and hold means for periodically sampling the offset voltage and the gain error voltage of the amplifier, and for holding the sampled voltage; storage means operable between sampling periods to store the sampled and held voltage; and further means operable during the sampling periods to continuously maintain the output of said high gain amplifier at a value that is gain error and voltage offset compensated.
  • the sample and hold means are adapted to periodically sample the offset voltage and gain error voltage at either, or both, of the inputs of the high gain amplifier.
  • the storage means are adapted to store the sampled and held voltage of either, or both, of the inputs of the high gain amplifier.
  • the sample and hold means include a further amplifier and, for each input of the high gain amplifier, coupling means for capacitively coupling an input of said high gain amplifier to a respective input of the further amplifier during the sampling periods, and connection means for connecting the sampled and held voltage to said storage means between sampling periods.
  • the voltage offset compensation circuit may be used in sampled-data circuits, or continuous-time amplifier circuits utilising either single-ended, or differential, inputs and either single-ended, or differential, outputs.
  • the other inputs of the high gain amplifier and further amplifier are connected to a voltage source.
  • the sample and hold means include coupling means for capacitively coupling each input of the further amplifier to a respective input of the high gain amplifier during the sampling periods, and connection means for connecting, between sampling periods, the said sampled and held voltage for each input of said high gain amplifier to respective storage means.
  • the storage means are adapted to store the sampled and held voltage at each input of said high gain amplifier, and the further means are adapted to continuously maintain each output of the high gain amplifier at a value that is gain error and voltage offset compensated.
  • Figure 1 of the accompanying drawings shows this known sampling technique applied to a standard configuration of integrated circuit.
  • one input (-ve) of a high gain amplifier 1 is connected to an input terminal 2 via a capacitor C1 and a field effect transistor T1, the junction of the capacitor C1 and transistor T1 being connected to a voltage source Vcm via a field effect transistor T2.
  • the amplifier input (-ve) is also connected to the voltage source Vcm via a capacitor C2 and a field effect transistor T3.
  • the junction of the transistor T3 and capacitor C2 is connected via a field effect transistor T4 to an output terminal 3 of the high gain amplifier 1.
  • a capacitor C3 is connected in series with a field effect transistor T5 and in parallel with a field effect transistor T6 between the output terminal 3 and the amplifier input (-ve).
  • the other input of the amplifier 1 is connected to the voltage source Vcm.
  • the transistors T1 to T6 are preferably provided by MOS (metal-oxide-silicon) transistors.
  • the voltage source Vcm is the common-mode, or reference potential, for the circuit and would typically be the midpoint of the voltage supply.
  • the gate electrodes G1 and G2 of the transistors T1 to T6 are connected to a two-phase output of a clock pulse generator (not illustrated), the clock pulses being applied to gates G1 during one phase, i.e. the first phase, of the two-phase output of the clock pulse generator to effect operation of the transistors T1, T4 and T5, i.e. the transistors are rendered conductive, whereas the clock pulses applied to the gates G2 during the other phase, i.e. the second phase, of the two-phase output of the clock pulse generator effect operation of the transistors T2, T3 and T6.
  • the transistors are non-conductive, i.e. effectively an open-circuit.
  • the transistors T1, T4 and T5 are open-circuit, the transistors T2, T3 and T6 are switched to the closed-circuit mode, and the output of the amplifier 1 is connected directly to the input via the transistor T6, i.e. a short-circuit connection is established between the input and the output of the amplifier 1.
  • This causes the fixed voltage offset at the input of the amplifier 1 to be stored on the sampling capacitors C1 and C2 which are each connected on one side thereof to the voltage source Vcm respectively via the transistors T2 and T3.
  • the next phase of the clock pulses i.e.
  • the transistors T1, T4 and T5 are in the closed-circuit mode and the transistors T2, T3 and T6 are in the open-circuit mode.
  • This causes said one sides of the capacitors C1 and C2 to be respectively connected to the input terminal 2 via the transistor T1, and the output terminal 3 via the transistor T4.
  • the result of this is to effectively subtract the voltage offset of the amplifier 1 from the input signal so that no errors, due to amplifier offset, appear at the output terminal 3.
  • This technique does not, however, compensate for errors due to finite amplifier gain and, in addition, the output of the amplifier 1 has to slew from its offset voltage to the output signal voltage, and conversely between each phase of the clock pulses.
  • That part of the circuit which comprises the components C1, T1 and T2 is the same as the corresponding circuit in Figure 1 of the drawings.
  • the capacitors C2 and C3 are connected in parallel via field effect transistors T8 and T9 across the input (-ve) and output of the high gain amplifier 1.
  • a field effect transistor T7 connects the capacitor C2 to the voltage source Vcm.
  • An auxiliary amplifier 4 has its input (-ve) capacitively coupled to the input (-ve) of the amplifier 1 via a field effect transistor T12, the capacitive coupling being effected by a capacitor C4.
  • the other inputs of the amplifiers 1 and 4 are connected to the voltage source Vcm.
  • the output of the amplifier 4 is connected directly to its input via a field effect transistor T13 and to the junction of two field effect transistors T10 and T11 which are connected in series between the junction of the capacitors C1 and C2 and the junction of the capacitor C4 and transistor T12.
  • clock pulses will be applied to the transistors T1, T7, T10 and T11 via the gates G1 during the first phase of the clock pulses and to the transistors T2, T8, T9, T12 and T13 via gates G2 during the second phase of the clock pulses.
  • auxiliary amplifier 4 in which the signal being processed is substantially constant between each phase of the clock pulses, enables the amplifier bandwidth to be less than that of the high gain amplifier 1 thereby giving rise to lower noise performance than known voltage offset and gain compensation techniques.
  • the voltage offset at the input of the amplifier 1 is sampled during the second phase of the clock pulses by the offset-cancelled sample and hold circuit enclosed by the dashed lines 5.
  • transistors T2, T8, T9, T12 and T13 are in closed-circuit mode thereby causing the input (-ve) of amplifier 1 to be connected to the input of amplifier 4 via the transistor T12 and capacitor C4, and the output of the amplifier 4 to be connected directly to its input via the transistor T13.
  • the effect of this is that the voltage offset at the input of the amplifier 1 charges the capacitor C4, the offset voltage being held on that side of the capacitor C4 which is connected to the transistor T12.
  • the circuit enclosed by the dashed line 5 samples and holds the offset voltage during the second phase of the clock pulse.
  • the capacitors C2 and C3 are connected in parallel across the input and output of the amplifier 1 and the capacitor C3, which is a voltage offset integration capacitor, holds the output of the amplifier 1 at a value which is gain and offset corrected.
  • the transistors T1, T7, T10 and T11 are in the closed-circuit mode and the transistors T2, T8, T9, T12 and T13 are open circuit.
  • the effect of this is that the side of the capacitor C4, on which the offset voltage is held, is connected directly to the junction of the capacitors C1 and C2 via the transistors T10 and T11, the capacitors C1 and C2 are connected in series between the input terminal 2 and the voltage source Vcm by means of the transistors T1 and T7, and the capacitor C2 is disconnected from the capacitor C3.
  • the charge transfer capacitors C1 and C2 are charged to the output voltage of the sample and hold circuit, which has a value equal to the sum of the offset voltage of the amplifier 1 and the signal gain error term.
  • the input voltage is scaled by the capacitor ratios and transferred to the output of the amplifier 1 without any error due to amplifier offset, or finite gain error.
  • the sample and hold circuit of Figure 2 of the drawings can optionally be clocked at a sub-multiple of the clock frequency used in the main circuit. These considerations allow a relatively low-noise, low-bandwidth auxiliary amplifier 4 to be used.
  • the sample and hold circuit of Figure 2 of the drawings is also sampling the low frequency noise of the amplifier 1 and this additionally provides low frequency noise reduction.
  • the sample and hold circuit of Figure 2 of the drawings is adapted to be fully differential, irrespective of the type of signal traversing the high gain amplifier 1.
  • the high gain amplifier 1 could have single-ended, or differential, input signals and single-ended, or differential, output signals, giving four possible combinations for the circuit. Each of these combinations are encompassed by the circuit diagram diagrammatically illustrated in Figure 3 of the accompanying drawings.
  • This circuit is a general-purpose circuit. Clearly, specific high gain amplifier 1 circuits could be implemented for each type of application. However, the area penalty in having such a multi-purpose circuit is small.
  • Both p-type and n-type transistors are used in the circuit of Figure 3 for the switching function provided by transistors T1, T2, T7 and T8 which are assumed to handle bipolar signals, and the amplifier 1 is of a type which can be simply modified to give either single-ended, or differential outputs.
  • capacitors match better than resistors. Furthermore, the impedance of such capacitors is much higher than resistors of the same area, over the usual signal ranges of interest, and this makes the bypassing of such components by transmission gates, i.e. the switchable field effect transistors of Figures 2 and 3 of the accompanying drawings, much easier.
  • the bypassing of passive components is used, for example, in variable gain amplifiers.
  • a voltage offset-compensated, finite-gain-compensated, precision amplifier including the sample and hold circuit of Figure 2 of the accompanying drawings, is diagrammatically illustrated in Figure 4 of the accompanying drawings.
  • the continuous-time precision amplifier includes an offset-cancelled, fully-differential, sample and hold circuit which is enclosed by the dashed line 7.
  • the Figure 4 sample and hold circuit includes an auxiliary amplifier 4 and associated circuitry provided by field effect transistors T11a/T11b, T12a/T12b and T13a/T13b and capacitors C4a/C4b.
  • both inputs of the high gain amplifier 1 are capacitively coupled to respective ones of the inputs of the auxiliary amplifier 4 respectively via the transistor T12a and the capacitor C4a, and the transistor T12b and the capacitor C4b.
  • the sample and hold circuit of Figure 4 of the accompanying drawings operates, during each phase of the two-phase clock pulses, in a manner similar to that described in relation to Figure 2 of the drawings, i.e. during the sampling phase, the capacitors C4a and C4b are respectively charged via the transistors T12a and T12b to a level whereby a voltage, corresponding to the voltage offset and gain error of the amplifier 1, is held on the plate of each capacitor, remote from the input of the auxiliary amplifier 4, during the hold phase.
  • This voltage charges floating capacitors C1a/C1b, C2a/C2b, i.e.
  • the transistors T1a/T1b, T7a/T7b, T10a/T10b, and T11a/T11b into the closed-circuit mode, during the first phase of the two-phase clock pulses, the transistors T2a/T2a, T8a/T8b, T9a/T9b and T13a/T13b being in the open-circuit mode and cause the voltage, corresponding to the voltage offset and gain error of the amplifier 1, to be applied to the junction of the capacitors C1a and C2a which are connected in series between the terminals 3a and 4a, and to the junction of the capacitors C1b and C2b which are connected in series between the terminals 3b and 4b. This results in charging of the capacitors C1a, C1b, C2a, C2b.
  • the voltage at the inputs of the high gain amplifier 1 is sampled by the sample and hold circuit, enclosed by the dashed line 7, during the second phase of the two-phase clock pulses, and is then transferred onto the floating capacitors C1a/C1b and C2a/C2b during the first phase of the two-phase clock pulses.
  • the floating capacitors C1a, C1b, C2a and C2b are respectively connected in parallel with gain setting capacitors C5a, C5b, C6a and C6b by the transistors T2a, T2b, T8a, T8b, T9a and T9b, which are switched, during this phase, to the closed-circuit mode.
  • This procedure effectively drives the junctions of the capacitors C5a/C6a and C5b/C6b to the input offset voltage of the high gain amplifier 1 and the output of amplifier 1 to a value which is gain and offset compensated.
  • a filtering function can be effected by making the value of the gain setting capacitors much higher than the value of the floating capacitors.
  • the sample and hold amplifier output need only change by an amount equal to the difference of the small input offset voltages of the two amplifiers.
  • the clock frequency need not be very high.
  • circuit of Figure 4 of the drawings also acts to reduce the low frequency noise of the high gain amplifier I and, therefore, that of the whole circuit.
  • the use of separate common-mode reference voltages Vicm (terminals 3a, 3b) and Vocm (terminals 4a, 4b) will allow precision level shifting, if this is required, between these potentials, otherwise they can be of the same value.
  • circuit diagrammatically illustrated in Figure 4 of the drawings is multi-purpose in that it can be used for applications having signals which provide either single-ended, or differential, inputs and require either single-ended, or differential, outputs.
  • the main amplifier is of a type that can be easily configured to have single-ended, or differential, outputs. Specific circuits could be used for each type of application but the area penalty of the described approach is small. With single-ended inputs either Vin or Vip is connected to Vcm.
  • FIG. 5 Another voltage offset-compensated, finite-gain-compensated, continuous-time amplifier, including the sample and hold circuit of Figure 2 of the accompanying drawings, is diagrammatically illustrated in Figure 5 of the accompanying drawings. It will be seen from Figure 5 that only the basic high gain amplifier 1 is shown together with the sample and hold circuit, enclosed by the dashed line 8, and the additional components for storing the sampled and held voltage between sampling periods, and for continuously maintaining the output of the high gain amplifier 1 at a value that is gain error and voltage offset compensated.
  • the circuit behaves, in total, as a near ideal gain and offset compensated amplifier to which additional components are added to define the required circuit function.
  • the additional components, associated with each input of the high gain amplifier 1, include field effect transistors T20 and T21 connected in series between a voltage source Vcm and an input terminal (Vip/Vin), field effect transistors T22 and T23 connected in series between an input of the high gain amplifier 1 and a respective output of the auxiliary amplifier 4, a capacitor C7 connected between the junction of the transistors T20 and T21, and the junction of transistors T22 and T23, and a capacitor C8 connected between the input of the amplifier 1 and the input terminal (Vip/Vin).
  • the signal at the terminals of the high gain amplifier 1 is periodically sampled, in a manner as previously outlined, by the sample and hold circuit 8, i.e. during the second phase of the two-phase clock pulses, and thereby results in the capacitors C4a and C4b being respectively charged via the transistors T12a and T12b to a level whereby a voltage, corresponding to the voltage offset and gain error of the amplifier 1, is held on the plate of each capacitor, remote from the input of the auxiliary amplifier 4, during the hold phase.
  • the clock pulses applied to the gates G1 of transistors T20a/T20b, T22a/T22b and T11a/T11b will cause these transistors to be conductive, i.e. switched to the closed circuit mode, and thereby cause the capacitors C7a and C7b to be connected in series between the voltage source Vcm and the output of the sample and hold circuit 8. This will result in storage of the sampled and held voltage on floating capacitors C7a and C7b.
  • the next phase of the clock pulses i.e.
  • the clock pulses applied to the gates G2 of the transistors T21a/T21b, T23a/T23b will cause these transistors to be conductive and the capacitors C7a and C7b will be respectively connected in parallel with capacitors C8a and C8b.
  • This will result in the potential on the plates of the capacitors C8a and C8b which are respectively connected to the input terminals Vip and Vin, to be forced to the potential of the voltage source Vcm and the potential on the plates of the capacitors C8a, C8b which are connected to high gain amplifier 1 to be forced to the input offset voltage of amplifier 1.
  • the total amplifier circuit will appear as an ideal amplifier with no voltage offset and without any finite gain errors.
  • the voltage offset of the high gain amplifier 1 is constant between clock phases and the gain error voltage is very small, it follows that the input of the auxiliary amplifier 4 and the charge on the sample and hold capacitors C4a and C4b are practically constant between clock phases. Also, the output of the auxiliary amplifier 4 need only change by an amount equal to the difference of the small input offset voltages of the two amplifiers 1 and 4. In addition, the clock frequency need not be very high. These considerations allow a relatively low noise, low-bandwidth auxiliary amplifier 4 to be used in combination with a relatively low clock rate. This acts to reduce sampled noise and other clock related effects. Making C7a and C7b much less than C8a and C8b provides an in-built filtering function which will further reduce the effect of auxiliary circuit noise to very low levels.
  • circuit of Figure 5 also acts to reduce the effect of low frequency noise in the main amplifier and therefore that of the whole circuit.
  • the offset-compensated continuous-time amplifier circuit of Figure 5 includes a high gain amplifier 1 with a differential output, the offset compensation techniques would work equally well with a single-ended output amplifier, changes only being necessary to the output stage of the amplifier 1.
  • the field effect transistors of Figures 3 to 5 of the accompanying drawings are preferably provided by MOS transistors.

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Claims (20)

  1. Versatzspannungskompensationsschaltung für einen hochverstärkenden Verstärker (1) mit einem festen Eingangsspannungsversatz, derart daß die Schaltung Abtast- und Haltemittel (5, 7) zum periodischen Abtasten der Versatzspannung und der Verstärkungsfehlerspannung des Verstärkers (1) und zum Halten der abgetasteten Spannung aufweist; Speichermittel (C1,C2), die zwischen Abtastperioden dazu dienen, die abgetastete und gehaltene Spannung zu speichern; und weitere Mittel (C3), die während den Abtastperioden dazu dienen, den Ausgang des hochverstärkenden Verstärkers (1) kontinuierlich bei einem Wert beizubehalten, welcher verstärkungsfehler- und spannungsversatzkompensiert ist.
  2. Kompenstionsschaltung nach Anspruch 1, dadurch gekennzeichnet, daß die Schaltung außerdem Mittel zur Erzeugung eines Taktsignals aufweist, um zu ermöglichen, daß die periodische Abtastung der Versatzspannung und der Verstärkungsfehlerspannung bewirkt wird.
  3. Kompensationsschaltung nach Anspruch 1 oder Anspruch 2, dadurch gekennzeichnet, daß die Abtast- und Haltemittel (5, 7) angepaßt sind, um die Versatzspannung und die Verstärkungsfehlerspannung periodisch an einem oder beiden der Eingänge des hochverstärkenden Verstärkers (1) abzutasten.
  4. Kompensationsschaltung nach Anspruch 3, dadurch gekennzeichnet, daß die Speichermittel (C1, C2) angepaßt sind, um die abgetastete und gehaltene Spannung des einen oder beider der Eingänge des hochverstärkenden Verstärkers (1) zu speichern.
  5. Kompensationsschaltung nach Anspruch 3 oder Anspruch 4, dadurch gekennzeichnet, daß die Abtast- und Haltemittel (5) einen weiteren Verstärker (4) und für jeden Eingang des hochverstärkenden Verstärkers (1) Kopplungmittel zum kapazitiven Koppeln eines Eingangs des hochverstärkenden Verstärkers (1) mit einem entsprechenden Eingang des weiteren Verstärkers (4) während der Abtastperioden und Verbindungsmittel zum Verbinden der abgetasteten und gehaltenen Spannung mit den Speichermitteln (C1, C2) zwischen den Abtastperioden aufweisen.
  6. Kompensationsschaltung nach Anspruch 5, dadurch gekennzeichnet, daß die Kopplungsmittel für jeden Eingang des hochverstärkenden Verstärkers (1) eine erste Kapazität (C4) aufweisen, die mit einem ersten Schaltmittel (T12) zwischen dem Eingang des hochverstärkenden Verstärkers (1) und dem entsprechendem Eingang des weiteren Verstärkers (4) in Reihe geschaltet sind, daß das erste Schaltmittel (T12) sich zwischen der ersten Kapazität (C4) und dem Eingang des hochverstärkenden Verstärkers (1) befindet, und angepaßt ist, um die Kapazität (C4) mit dem Eingang des hochverstärkenden Verstärkers nur während der Abtastperioden zu verbinden, und daß der Eingang des weiteren Verstärkers (4) mit einem entsprechenden seiner Eingänge mit einem zweiten Schaltmittel (T13) verbunden ist, das angepaßt ist, um eine direkte Verbindung zwischen dem Eingang und dem Ausgang des weiteren Verstärkers (4) nur während der Abtastperioden zu bewirken.
  7. Kompensationsschaltung nach Anspruch 6, dadurch gekennzeichnet, daß die Verbindungsmittel ein drittes Schaltmittel (T10, T11) aufweisen, das angepaßt ist, daß es die Verbindung der ersten Kapazität (C4) und das erste Schaltmittel (T12) und den Ausgang des weiteren Verstärkers (4) mit dem Speichermittel nur zwischen den Abtastperioden verbindet.
  8. Kompensationsschaltung nach Anspruch 7 dadurch gekennzeichnet, daß die Speichermittel für jeden Eingang des hochverstärkenden Verstärkers (1) eine zweite (C1) und einen dritte (C2) Kapazität aufweisen, die in Reihe geschaltet sind, wobei jede der Kapazitäten auf einer deren Seiten mit den dritten Schaltmitteln (T10, T11) verbunden ist, daß die anderen Seiten der zweiten und dritten (C2) Kapazitäten jeweils mit einem Eingangsanschluß (2) der Kompensationsschaltung und einer Spannungsquelle (Vcm) durch vierte Schaltmittel (T7) verbunden sind, die angepaßt sind, um nur zwischen dem Abtastperioden betrieben zu werden, daß die andere Seite des zweiten Kapazität (C1) mit der Spannungsquelle (Vcm) durch fünfte Schaltmittel (T2) verbunden ist, die angepaßt sind, daß sie nur zwischen den Abtastperioden betrieben werden, und daß die dritte Kapazität (C2) parallel zu den weiteren Mittel (C3) über sechste Schaltmittel (T8, T9) verbunden ist, die angepaßt sind, um nur während der Abtastperioden betrieben zu werden.
  9. Kompensationsschaltung nach Anspruch 8, dadurch gekennzeichnet, dass die weiteren Mittel für jeden Eingang des hochverstärkenden Verstärkers (1) eine vierte Kapazität (C3) aufweisen, die zwischen dem Eingang und einem entsprechenden Ausgang des hochverstärkenden Verstärkers (1) verbunden ist.
  10. Kompensationsschaltung nach Anspruch 7, dadurch gekennzeichnet, daß die Speichermittel für jeden Eingang des hochverstärkenden Verstärkers (1) eine fünfte Kapazität (C7) aufweisen, die an deren einen Seite mit den dritten Schaltmitteln verbunden ist, daß die fünfte Kapazität (C7) an deren anderer Seite mit der Spannungsquelle (Vcm) durch siebte Schaltmittel (T20) verbunden ist, die angepaßt sind, um nur zwischen den Abtastperioden betrieben zu werden, und daß die fünfte Kapazität (C7) parallel zu den weiteren Mitteln (C8) über achte Schaltmittel (T21, T23) verbunden ist, die angepaßt sind, um nur während der Abtastperioden betrieben zu werden.
  11. Kompensationsschaltung nach Anspruch 10, dadurch gekennzeichnet, daß die weiteren Mittel für jeden Eingang des hochverstärkenden Verstärkers (1) eine sechste Kapazität (C8) aufweisen, die zwischen einem Eingang des hochverstärkenden Verstärkers (1) und einem entsprechenden Eingang (Vip, Vin) der Kompensationsschaltung verbunden ist, und daß das die fünfte (C7) und sechste (C8) Kapazität parallel mit den achten Schaltmitteln (T21, T23) während der Abtastperioden verbunden sind.
  12. Kompensationsschaltung nach Anspruch 8, dadurch gekennzeichnet, daß die Schaltung Verstärkungseinstellmittel einschließlich in Reihe geschaltete Kapazitäten (C5, C6) für jeden Eingang des hochverstärkenden Verstärkers (1) aufweist, von denen jede parallel zu der entsprechenden der zweiten (C1) und dritten (C2) Kapazitäten nur während der Abtastperioden verbindbar ist, und daß eine der in Reihe geschalteten Kapazitäten (C6) der Verstärkungseinstellmittel, die parallel zu den dritten Kapazitäten (C6) verbunden ist, das die weiteren Mittel bildet.
  13. Kompensationsschaltung nach einem den Ansprüche 3 bis 12, dadurch gekennzeichnet, daß die Schaltung einpolige oder differentielle Eingänge und einpolige oder differentielle Ausgänge verwendet und daß für den Fall einpoliger Eingänge die anderen Eingänge des hochverstärkenden Verstärkers (1) und des weiteren Verstärkers (4) mit einer Spannungsquelle (Vcm) verbunden sind.
  14. Kompenstionschaltung nach einem der Ansprüche 5 bis 12, dadurch gekennzeichnet, daß die Schaltung differentielle Eingänge und differentielle Ausgänge verwendet, und daß die Abtast- und Haltemittel (5,7) Kopplungmittel (C4a, C4b) zur kapazitiven Kopplung jedes Eingangs des weiteren Verstärkers (4) mit einem entsprechenden Eingang des hochverstärkenden Verstärkers (1) während der Abtastperioden und Verbindungsmittel, um die abgetastete und gehaltene Spannung jedes Eingangs des hochverstärkenden Verstärkers (1) mit den entsprechenden Speichermitteln zwischen den Abtastperioden zu verbinden, aufweisen.
  15. Kompensationsschaltung nach Anspruch 14, dadurch gekennzeichnet, daß die Speichermittel angepaßt sind, um die abgetastete und gehaltene Spannung bei jedem Eingang des hochverstärkenden Verstärkers (1) zu speichern, und daß die weiteren Mittel angepaßt sind, um jeden Ausgang des hochverstärkenden Verstärkers (1) bei einem Wert kontinuierlich beizubehalten, der verstärkungsfehler- und spannungsversatzkompensiert ist.
  16. Kompensationsschaltung nach einem der Ansprüche 6 bis 15, dadurch gekennzeichnet, daß die Schaltmittel Feldeffekttransistoren aufweisen.
  17. Kompensationsschaltung nach Anspruch 16, dadurch gekennzeichnet, daß die Feldeffekttransistoren MOS-Transistoren sind.
  18. Hochverstärkende Verstärkerschaltung mit einer Spannungskompensationsschaltung nach einem der vorhergehenden Ansprüche.
  19. Hochverstärkende Verstärkerschaltung nach Anspruch 18, dadurch gekennzeichnet, daß die Schaltung eine Abtastdatenschaltung ist.
  20. Hochverstärkende Verstärkerschaltung nach Anspruch 18, dadurch gekennzeichnet, daß die Schaltung eine Dauerzeitschaltung ist.
EP96300822A 1995-02-21 1996-02-07 Offsetspannungskompensationsschaltung Expired - Lifetime EP0729223B1 (de)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
GB9503425A GB2298329B (en) 1995-02-21 1995-02-21 Voltage offset compensation circuit
GB9503425 1995-02-21

Publications (3)

Publication Number Publication Date
EP0729223A2 EP0729223A2 (de) 1996-08-28
EP0729223A3 EP0729223A3 (de) 1998-09-16
EP0729223B1 true EP0729223B1 (de) 2003-04-23

Family

ID=10769987

Family Applications (1)

Application Number Title Priority Date Filing Date
EP96300822A Expired - Lifetime EP0729223B1 (de) 1995-02-21 1996-02-07 Offsetspannungskompensationsschaltung

Country Status (6)

Country Link
US (1) US5812023A (de)
EP (1) EP0729223B1 (de)
JP (1) JPH08256028A (de)
AT (1) ATE238629T1 (de)
DE (1) DE69627567T2 (de)
GB (1) GB2298329B (de)

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* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
FR2754405B1 (fr) * 1996-10-08 1998-12-18 Dolphin Integration Sa Montage a faible bruit d'un amplificateur
US6380790B1 (en) * 2000-02-11 2002-04-30 President And Fellows Of Harvard College Integrator topplogy for continuous integration
US6583660B2 (en) * 2001-05-25 2003-06-24 Infineon Technologies Ag Active auto zero circuit for time continuous open loop amplifiers
US6661283B1 (en) * 2002-10-03 2003-12-09 National Semiconductor Corporation Wide gain range and fine gain step programmable gain amplifier with single stage switched capacitor circuit
US7302246B2 (en) * 2002-12-23 2007-11-27 Intel Corporation Programmable gain amplifier with self-adjusting offset correction
DE102004022991B3 (de) * 2004-05-10 2005-12-08 Infineon Technologies Ag Abtast-Differenzverstärker und Abtast-Verstärker
US9355693B2 (en) * 2013-03-14 2016-05-31 Intel Corporation Memory receiver circuit for use with memory of different characteristics
US9960782B2 (en) * 2015-09-11 2018-05-01 Texas Instruments Incorporated Precharge switch-capacitor circuit and method
US10938362B2 (en) 2017-07-31 2021-03-02 Renesas Electronics Corporation Offset cancellation

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Publication number Priority date Publication date Assignee Title
US4282515A (en) * 1979-07-20 1981-08-04 Harris Corporation Analog to digital encoding system with an encoder structure incorporating instrumentation amplifier, sample and hold, offset correction and gain correction functions
US4322687A (en) * 1980-05-19 1982-03-30 Bell Telephone Laboratories, Incorporated Operational amplifier with improved offset correction
US4438354A (en) * 1981-08-14 1984-03-20 American Microsystems, Incorporated Monolithic programmable gain-integrator stage
US4439693A (en) * 1981-10-30 1984-03-27 Hughes Aircraft Co. Sample and hold circuit with improved offset compensation
US4546324A (en) * 1982-12-27 1985-10-08 Intersil, Inc. Digitally switched analog signal conditioner
US4962325A (en) * 1988-09-09 1990-10-09 Analog Devices, Inc. Sample-hold amplifier circuit
US5258664A (en) * 1991-07-05 1993-11-02 Silicon Systems, Inc. Operational amplifier with self contained sample and hold and auto zero
NL9200327A (nl) * 1992-02-21 1993-09-16 Sierra Semiconductor Bv Offset-gecompenseerde bemonsterinrichting en werkwijze voor de bediening daarvan.
JP3222276B2 (ja) * 1993-07-30 2001-10-22 セイコーインスツルメンツ株式会社 コンパレータ回路およびコンパレータ回路の制御方法
GB2296399B (en) * 1994-11-23 1998-10-14 Motorola Inc Compensating circuit

Also Published As

Publication number Publication date
JPH08256028A (ja) 1996-10-01
US5812023A (en) 1998-09-22
GB2298329B (en) 2000-02-16
DE69627567D1 (de) 2003-05-28
ATE238629T1 (de) 2003-05-15
EP0729223A2 (de) 1996-08-28
EP0729223A3 (de) 1998-09-16
DE69627567T2 (de) 2003-11-27
GB2298329A (en) 1996-08-28
GB9503425D0 (en) 1995-04-12

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