EP0600799B1 - An active antenna with variable polarisation synthesis - Google Patents

Description

The invention relates to active antennas, which are made up of a large number of radiating sources excited by microwave power amplifiers during transmission, or whose received signals are amplified by low noise amplifiers at reception. Such antennas are used in various applications such as telecommunications or radars; the invention will be particularly advantageous for radars. In the radar field indeed, the usual architecture of a monostatic radar involves the use of a lane transmission and reception channel which lead to the same radiant source. Usually a switch allows the selection of the transmission channel to transmit a radar signal in pulses, the time space between the emission pulses being used to listen, in selecting the reception channel, the radar echoes which come back from the environment.

In the telecommunications sector, the increase demand makes us look for a better use of the radio spectrum. This concern translates by the use of fine, orientable beams, and sometimes even polarized to allow the reuse of frequencies. These characteristics can be combined advantageously in realizations of array antennas. The invention will find an application in such antennas telecommunications, more specifically designed but not exclusively for the broadcast.

In the field of multistatic radars, antennas transmission and reception are spaced from each other others, sometimes by the tens or even hundreds of kilometers. Network antennas can be designed to fulfill the two missions, emission and reception, or they can be designed to fill only one of these two missions. A variant of the invention will be applicable for each of these possibilities.

The design of active antennas for radars monostatic to have evolved considerably in recent years, and in the current state of the art, the radiating sources are connected to active transmission / reception modules (MAER or T \ R in English), made in MMIC technology (Monolithic Microwave Integrated Circuits) or in hybrid technology. Transmission / reception switching is generally included in the active module, a schematic diagram of which is given in Figure 1, with its location within the antenna.

Figure 1 schematically shows an antenna active radar, operating alternately in transmission and in reception. The alternation of transmission / reception is ensured by switches 25, 52 controlled by a synchronization clock 24. On this figure 1, orthogonal polarizations can be selected by switch 26, for the reception as for the emission. Phase and gain are controllable by control means 23, also both in transmission and in reception. The values of order which will be provided for ordering a track of reception given, are not necessarily the same as for the same channel used for transmission.

In Figure 1, a single active module transmission / reception is shown, including phase shifter controllable 27 and a controllable attenuator 28, for adjust the gain of the module. However, you need a module active per channel and, in this example, there are m · m 'channels, each channel being connected to a compound radiating source from K elementary radiators S 1 / ij to S k / ij, m 'being the number of source columns, of which only the first and the second are (partially) represented.

In transmission mode, the transmitter 21 supplies its signals to a distributor / combiner 22, which supplies the active I / O modules. The phase and the attenuation of the signal will be determined by the controllable phase shifter 27 and the controllable attenuator 28, according to the instructions given by the control computer 23. Then, the switches 25 and 52 will be controlled by the clock 24 to engage the power channel, and the signal will be amplified by the power amplifier 29, before being sent to the radiating sources S _ij .

In reception mode, the receiver 31 receives the signals from the combiner / distributor 22, which are routed by the active E / R modules. In the E / R modules, the signals coming from the radiating sources S _ij are switched by the switches 25, 52 on the reception channel and pass through a low noise amplifier 30. Then, the phase shift and the attenuation are applied by the controllable phase shifter 27 and the controllable attenuator 28, controlled by the control computer 23.

This configuration allows electronic scanning of the beam, in transmission and reception thanks to phase shifters controllable. Beams can also be formed, for example with stiff sides and side lobes low, to improve antenna performance when ambiguities of echoes, and in the presence of noise sources, thanks to controllable phase shifters and attenuators. Finally, thanks to the switches 26, one of the two orthogonal polarizations can be selected, interesting for the radar because the useful signal and the noise have different variations depending on the polarization, which allows an optimization of the signal ratio on noise by acting on the polarization.

M est le facteur de mérite de l'antenne ;

N est le nombre d'amplificateurs de puissance ;

P_e représente la puissance émise par chaque amplificateur

L_e représente les pertes après le (ou les) amplificateur(s) de puissance ;

D_e et D_r représentent les directivités des diagrammes générés par le réseau des éléments rayonnants, respectivement en émission et en réception ;

L_r représente les pertes avant le (ou les) amplificateur(s) faible bruit en réception ;

FB représente le facteur de bruit de la chaíne réception. Si le gain du premier amplificateur faible bruit est suffisant, le facteur du bruit de la chaíne de réception est pratiquemet égal au facteur du bruit de l'amplificateur faible bruit.

In fact, the performance of a radar is essentially characterized by a link budget which determines the ratio of useful signal to undesirable noise. The terms which depend on the microwave part of the radar are as follows: M = NP e - L e + D e + D r - L r - FB; or :

M is the antenna merit factor;

N is the number of power amplifiers;

P _e represents the power emitted by each amplifier

L _e represents the losses after the power amplifier (s);

D _e and D _r represent the directivities of the diagrams generated by the network of radiating elements, respectively in transmission and in reception;

L _r represents the losses before the low noise amplifier (s) on reception;

FB represents the noise factor of the reception chain. If the gain of the first low noise amplifier is sufficient, the noise factor of the reception chain is practically equal to the noise factor of the low noise amplifier.

It appears that, to have the best factor of merit, we seek to minimize the losses and the noise factor of the reception chain, while optimizing the transmitted power and the directivity of the radiation patterns. For a given elementary power P _e from a source of the network, the directivities as well as the term in N · P _e can be optimized with a larger number of sources.

Elementary sources, in the context of this invention, are radiant elements capable of providing polarized radiation, the polarization of which can take at least two orthogonal values, for example horizontal (H) and vertical (V), or circular polarizations right and left (R, L).

Cornets with square, circular or hexagonal are radiating elements that can generate H or V polarizations. They are particularly suitable to high power antennas or ground is not a element criticized.

The printed radiant elements (known to the man of craft by name in English "patch") are paving stones metallic photo-etched on a thin dielectric substrate with low microwave losses. They allow the production of radiant panels comprising a lot of elementary sources, these panels can be thin, light, and even conformable. For generate orthogonal polarizations with patches it just excite them by two points offset by 90 ° by relative to the center of the patch, as shown in Figure 1. The link between the MAER and the patch can be online coaxial or microstrip, for example. If only one MAER must order several patches, they can be grouped in sub-networks, connected to MAER by distributors in microstrip line for each H and V polarization

To generate circular polarizations radiated by patches, just the same excitations as those that are necessary to generate linear polarizations orthogonal; only orthogonal excitations linear must be offset by 90 ° in phase, in addition to their physical offset of 90 ° around the center of the patch. This is easily achieved using a hybrid coupler 90 °, placed between the MAER and the patch, whose excitation on one of its inputs gives us circular polarization right, and on the other of its entries gives us the left circular polarization.

To excite cones in circular polarizations or ellipticals it is known the use of excitation by means of such orthomodes and polarizers. These techniques are well known to those skilled in the art, and exposed for example in the document D2 = US-A-3,357,013, and it will not be necessary to explain them further for a good understanding of the present invention.

Compared to the configuration of FIG. 1, it is known to make some modifications making it possible to improve the merit factor of the antenna by reducing the losses of the circuit between the sources of the antenna and the amplifiers. On the one hand, the E / R (transmit / receive) switches closest to the sources (item 52 in Figure 1) can be replaced by circulators, which are more bulky and heavier, but with lower losses and better power handling. On the other hand, the switching between H and V polarizations can be performed upstream of the power amplifier, as shown in Figure 2. The losses L _e and L _r are thus reduced, but these chains must then be doubled amplification, as we see in Figure 2. Such a solution is taught for example by the document D1 = DE-U-91 13 444.

In Figure 2, we see that the same elements have the same references as in Figure 1; only the landmarks associated with one or other of the chains amplifiers carry clues that signal their membership: the index "a" for the chain H, and the index "b" for chain V. Phase and attenuation control amplitude is always carried out by means of command 23, and the H or V polarization command, as well that the selection of transmission or reception from clock 24, the output of which is connected to the bias 26 as well as the two E / R switches 25a, 25b.

The circuit of Figure 2 further includes, a additional protection for low noise amplifiers against any unwanted reflections coming back radiating sources S k / ij when they are powered by power amplifiers, in case of mismatch of the antenna. These protection means are switches 32a, 32b placed in shunt towards ground, on the inputs of low noise amplifiers. These switches are also controlled by the clock 24, at the same time and in synchronization with E / R switches 25a, 25b. The optional isolators 33a, 33b, allow evacuation to the mass the power reflected (a second time) by these protection means 32a, 32b.

In the transmission position of these switches 25a, 25b, either of the microwave amplifiers power, 29a or 29b, will be requested, depending on the position of the polarization switch 26.

The microwave circulators 52a, 52b replace advantageously, on the amplifier chains H, V respectively, the switch 52 of the MAER of FIG. 1. The insertion losses of the circulators are lower than losses caused by conventional switches used in MAER.

In the reception position of these switches 25a, 25b, and possibly 32a, 32b, one or other of the low noise microwave amplifiers, 30a or 30b, will be requested, depending on the position of the polarization 26.

In these systems known from the prior art, such as we presented them, there are disadvantages important due to the different switches required to perform either transmission, reception or H polarization, either in V or else in polarization circular D, i.e. in G.

If, according to a first solution of the prior art, the switches are located between the amplifiers and the radiating elements (FIG. 1), they severely strain the radar link budget (or the merit factor of the active antenna) , by their losses L _e and L _r which occur twice, in transmission and reception.

If, on the contrary and according to a second solution of prior art, the switches are placed upstream of the power amplifiers, and downstream of the amplifiers low noise (fig. 2), this first problem is avoided, but in this case two amplifiers of each type are required to each radiating source, only one of the four operates at a given time, in turn depending on the polarization and according to the emission or the reception. The size and mass of the MAER are increased, without power is not. Merit factor increased by reducing losses, because there is no longer any high level polarization switch. this is particularly penalizing for on-board missions, all the more so for satellites than for aircraft. In in addition, the MAERs in Figure 2 are likely to cost almost twice as expensive as those in Figure 1.

The invention overcomes these drawbacks of prior art. According to the invention, a configuration of MAER is proposed which avoid the losses of the switches of the first solution, without increasing the mass and the overall dimensions of the assembly, at equal power, as in the second solution.

For these purposes, the invention provides an alternating transmission and reception (E / R) microwave circuit for synthesis antenna with variable polarization synthesis, this E / R circuit able to supply excitation signals for at least two polarizations. orthogonal to radiating sources via two respective channels, these channels supplied respectively by two amplifying power transmission chains; this E / R circuit able to receive at least two signals having orthogonal polarizations detected by these same sources and supplying two low noise amplifier chains in reception; said E / R circuit comprising, in addition to the phase shifter located on the common channel intended to point or form the beam, at least one phase shifter controllable on a transmission channel, and at least one phase shifter controllable on a reception channel intended to choose the polarization said circuit characterized in that the two power amplifier chains operate simultaneously during transmission, and in that the two low noise amplifier chains operate simultaneously during reception.

According to a preferred characteristic, the polarizations H (horizontal) and V (vertical) are obtained by sum or difference of two orthogonal polarizations, inclined by 45 ° to the horizontal: each of them is connected directly to one of the two paths of the E / R circuit.

According to an advantageous embodiment, the two chains power amplifiers are powered from a power divider in phase, allowing easy synthesis linear orthogonal polarizations; according to another advantageous realization, the two amplifying chains of power are supplied from a hybrid coupler to two 90 ° phase-shifted outputs, allowing easy synthesis circular polarizations.

According to one characteristic, said phase shifters are one bit digital controllable phase shifters, this bit corresponding to its value, either at 0 ° or 180 °.

According to a variant, said phase shifters are one bit digital controllable phase shifters, this bit depending on its value, either 0 ° or 90 °.

According to another characteristic, said phase shifters are two-bit digital controllable phase shifters, of which a first bit corresponding according to its value is 0 either 180 ° and a second corresponding bit according to its value either at 0 ° or 90 °.

In this case, we synthesize one of the four following polarizations: linear H or V, right circular or left.

According to an advantageous embodiment, an attenuator variable allows to adjust the gain of at least one chain power amplifier.

According to another advantageous embodiment, a variable attenuator adjusts the gain of at least one low noise amplifier chain.

According to a high-performance embodiment, said E / R circuit further includes at least two phase shifters and at least two quasi-continuously controllable attenuators, allowing the synthesis of any polarization, linear, circular, or elliptical.

According to a particular variant, said command quasi-continuous phase shifters and attenuators is analog design.

According to a complementary variant, said command quasi-continuous phase shifters and attenuators is digital design, with a high number of bits, allowing the synthesis of any polarization, linear, circular, or elliptical.

The invention also relates to an antenna comprising E / R circuits according to one of the embodiments or previous variants.

According to a variant of the antenna of the invention, the radiant sources are of printed type (or patch in English).

According to another variant of the antenna, the sources radiant consist of annular slots photo-etched on one side of a dielectric substrate with low microwave losses, these slots being excited by photo-etched lines on the face opposite.

According to another variant of the above, the slots ring fingers are excited by photo-etched lines on a hanging substrate.

According to one characteristic, the E / R circuits are made in MMIC technology.

Alternatively, miniature circulators are added to MMIC circuits to increase power maximum allowed.

According to a preferred embodiment, duplexers miniatures comprising a circulator and an isolator are added to the circuits of the invention, in order to better isolate the transmission channels of the reception channels.

According to a last embodiment, the antenna according to the invention is self-adaptive in polarization, for ability to extract useful radar signal in the presence of interference of any fixed polarization; to do, the antenna detects the polarization of the jammer, and adapts the phases and possibly the amplitudes of the signals emitted to work in a polarization orthogonal to that of the jammer.

La figure 1, déjà décrite, représente schématiquement un exemple d'architecture d'une antenne active pour radar de l'art antérieur travaillant en polarisations linéaires orthogonales, avec ses MAER et ses sources rayonnantes ;

La figure 2, déjà décrite, représente schématiquement un exemple d'un circuit E/R de l'art antérieur, ayant des pertes plus faibles que le circuit de la figure 1 ;

La figure 3 représente schématiquement un exemple de circuit MAER selon l'invention ;

La figure 4 représente schématiquement un deuxième exemple de circuit MAER selon l'invention, comprenant en plus les caractéristiques de la figure 2 ;

La figure 5 représente schématiquement une variante de l'invention, capable de synthétiser une polarisation quelconque.

La figure 6 représente schématiquement comme une variante de l'invention, un circuit hyperfréquence d'émission ou de réception, à synthèse de polarisation variable.

Other characteristics and advantages of the present invention will emerge from the detailed description which follows, with its annexed figures including:

FIG. 1, already described, schematically represents an example of architecture of an active antenna for radar of the prior art working in orthogonal linear polarizations, with its MAERs and its radiating sources;

FIG. 2, already described, schematically represents an example of an E / R circuit of the prior art, having lower losses than the circuit of FIG. 1;

FIG. 3 schematically represents an example of a MAER circuit according to the invention;

FIG. 4 schematically represents a second example of a MAER circuit according to the invention, further comprising the characteristics of FIG. 2;

FIG. 5 schematically represents a variant of the invention, capable of synthesizing any polarization.

FIG. 6 diagrammatically represents, as a variant of the invention, a transmission or reception microwave circuit, with variable polarization synthesis.

All figures are given as examples non-limiting; the skilled person will learn the education it takes to generalize these examples many other achievements, without leaving the part of the invention.

In all the figures, the same references designate the same elements, these elements being functions microwave, organized in a schematic flowchart of the circuit. In case a function can be performed by one, among several components for a result similar, this is indicated in the description which follows. Likewise, the figures represent general diagrams, therefore other variants on these diagrams can be made without departing from the scope of the invention.

In FIG. 3, we see a first schematic example of a MAER circuit according to the invention. Compared to Figures 1 and 2 already described, we have simplified the diagram by ignoring the environment of the circuit shown; nevertheless, this circuit is intended to be installed in the same way as the circuits of the prior art, between a distributor / combiner (22 in FIG. 1) and a network of radiating sources S _ij . As in the previous figures, the variable attenuator 28 and phase shifter 27 are controlled by instructions given by the control computer (not shown), and the E / R switch 6 is controlled by a clock (not shown). The elements 35a, 35b correspond either to the E / R switches (52 in FIG. 1), or to the circulators (52a, 52b in FIG. 2), the function of which in both cases is to pass either the transmission power between the power amplifiers 29a, 29b and the corresponding radiating source S _i j, ie the reception signal between the radiating source S _ij and the low noise amplifiers 30a, 30b.

The elements 5a are distributors and the elements 5b are combiners, the nature of which will be discussed below.

Elements 1, 2, 3, 4 are phase shifters, including at least one phase shifter controllable on a transmission channel (s ₁ or s ₃ ), and at least one phase shifter controllable on a reception channel (s ₂ or s ₄ ). According to the invention, it is therefore possible that there are only two controllable phase shifters, for example 3 and 4, and that the elements 1, 2 can be removed from this diagram. Several variants of the invention can be built around this general diagram, in particular by playing on the different possibilities for these elements 1, 2, 3, 4; a number of these possibilities will be described later.

In a first variant of the invention, the element 5a is a power distributor and the element 5b is a power combiner, both operating in phase, that is to say that the phase of the signals s ₁ and s ₃ is the same, and that the signals s ₂ and s ₄ are also combined in phase. In this first variant, the simplest embodiment according to the invention, the elements 1 and 2 does not exist; and elements 3 and 4 are single-bit phase shifters, which introduce a phase shift of either 0 ° or 180 °, depending on the value of the control bit, supplied by control means not shown.

The radiating source S k / ij shown in FIG. 3 is an engraved "patch" of square shape, whose orientation is schematically significant. Indeed, the square is oriented with its diagonals respectively to the horizontal and vertically. The propagation lines from switches or circulators 35a, 35b to the patch are mutually perpendicular and oriented at 45 ° from diagonals of the patch.

In an ideal circuit according to FIG. 3, ignoring the insertion losses and propagation delays in the phase shifters 3 and 4, the amplitude of the signals s ₁ and s ₃ is the same, and the signals s ₂ and s ₄ have the same amplitudes too. If the phase shifter 3 is controlled, according to its control bit, at a value of 0 °, the two accesses are excited in phase by the two power amplifiers 29a, 29b, which results in a wave having a horizontal linear polarization. If on the other hand the phase shifter 3 is controlled, according to its control bit, at a value of 180 °, the two accesses are excited in phase opposition by the two power amplifiers 29a, 29b, which results in a wave having a polarization vertical linear.

In the same way for reception, if the phase shifter 4 is controlled, according to its control bit, at a value of 0 °, the two accesses which are excited in phase, and after amplification by the two low noise amplifiers 30a, 30b, are combined in phase by the combiner 5b, which corresponds to a wave having a horizontal linear polarization on reception. If on the other hand the phase shifter 4 is controlled, according to its control bit, at a value of 180 °, the combiner 5b will have on its inputs the two signals s ₂ and s ₄ whose signal s ₄ will have been phase shifted by 180 °, which means that, only if the two accesses are excited in phase opposition, and after amplification by the two low noise amplifiers 30a, 30b, the result is obtained which corresponds to a wave having a vertical linear polarization.

In practice, to achieve exactly the synthesis vector of the desired polarizations such as we described it in the preceding paragraphs, it is necessary take into account the insertion losses of the phase shifters 3, 4, as well as the gains and insertion phases of the amplifiers 29a, 29b and 30a, 30b. For example, real amplifiers will be paired (29a, b and 30a, b) so as to have the same gain and the same insertion phase, and the loss of 3.4 phase shifters is compensated by a slight imbalance of dividers 5a / combiners 5b: for example if the phase shifters lose 1 dB, dividers / combiners are designed to present the same gap between the amplitude of their two outputs / inputs (respectively). He is also at note that the two states 0 and 180 ° of the phase shifters must have the same insertion loss, which is commonly made by those skilled in the art, whatever the technology used for these phase shifters. In case the pairing of two amplifiers with the same gain and insertion phase would be difficult (MMIC technology, for example), balancing of the two channels must be done using the devices for adjusting these parameters, which will be added to the schematic circuit of FIG. 3.

The schematic circuit of FIG. 3 can also provide orthogonal circular polarizations, with hybrid couplers at 90 ° 5a, 5b in place of the phase dividers / combiners considered above. With a hybrid coupler 5a on the transmission channel, for example, the two power amplification chains will carry the same signal, except that the signal s ₃ will be offset by + 90 ° in phase, compared to the signal s ₁ (when the phase shifter 3 has a value of 0 °). Excitation of the patch by two orthogonal accesses, with a signal s ₃ on the first access, offset by + 90 ° in phase with respect to the signal s ₁ on the second orthogonal access, results in a wave having a right circular polarization, for example. By switching the control bit of the phase shifter 3, a phase shift of 180 ° is obtained on the signal s ₃ , which is equivalent to an offset of -90 ° with respect to the signal s ₁ . The result will be a radiated wave with a left circular polarization.

The reception channel can synthesize waves with right and left circular polarization in the same way, the design being perfectly symmetrical between the tracks transmission and reception.

We saw in this first example, the characteristics of the circuit according to the invention which confer its advantages over the prior art: two parallel amplification channels operate simultaneously, in transmission as in reception. this allows get twice the power of the configuration of prior art. In addition, the losses of the dividers and phase shifters do not intervene, nor in the link budget radar nor in the antenna figure of merit because they are located upstream of the power amplifiers in emission, and downstream of the low noise amplifiers in reception.

Still with respect to this FIG. 3, we will discuss other possible variants of embodiments according to the invention. For example, it is obvious that the elements 35a, 35b can be E / R switches controlled by the clock (not shown), or they can be circulators, which allow the signal to pass through power amplifiers 29a, 29b to the radiating source S _ij , or vice versa, from the source S _ij to the low noise amplifiers 30a, 30b, but in no case will the signal be allowed to pass from the power amplifiers 29a, 29b to the low noise amplifiers 30a, 30b.

Still according to another variant of FIG. 3, the circuit can have the ability to synthesize either linear orthogonal polarizations or orthogonal circular polarizations. To do so, just put phase shifters with two control bits in blocks 3, 4 on the diagram, with dividers / combiners in phase for elements 5a, 5b. The first command bit, depending on its value, will give a phase from 0 ° or 180 ° as before, and to this phase will be added the phase controlled by the second control bit, from 0 ° or 90 °. If the second bit commands a phase shift of 0 °, we we find in the previous case of polarizations orthogonal linear; if on the other hand the second bit of command designates a 90 ° phase shift, the circuit is equivalent to that discussed before where elements 5a, 5b were 90 ° hybrid couplers, that is, we we find ourselves in a configuration that allows us to synthesize right and left circular polarizations.

You can get the same performance with a circuit alternative, in which the elements 5a, 5b are 90 ° hybrid couplers, and we put phase shifters 0 or 90 ° to a command bit in boxes 1, 2 of the Figure 3, and 0 or 180 ° phase shifters with one control bit in boxes 3, 4 of the same figure. The result is identical to the result of the previous paragraph. This configuration provides an additional benefit of realization, if the losses of the phase shifters 1, 2, 3, 4 are the same, because the hybrid couplers 5a, 5b can be in this case balanced couplers, more components currents than unbalanced couplers.

This last observation leads us to give two more variants of the invention, still according to the figure 3. A variant allowing the synthesis of linear orthogonal polarizations, includes two 90 ° hybrid couplers 5a, 5b, and four 0 or 90 ° phase shifters to a control bit 1, 2, 3, 4. As in the first variant described, a horizontal polarization of emission is obtained if the phase shifter 1 has a phase shift of 0 ° and the phase shifter 3 has a 90 ° phase shift (which is added to the 90 ° phase shift of the hybrid coupler); with the same to reception, with 0 ° on the phase shifter 2, and 90 ° on the phase shifter 4. Vertical polarization is obtained by reversing the phase shifts of the four phase shifters. As in the previous case, the realization can be simplified by the fact that with identical phase shifters on all four channels, just select components with the same losses and insertion phases to approach the circuit ideal for polarization synthesis.

A final variant of Figure 3 will be for synthesize orthogonal circular polarizations, using the same trick as in the previous case: four phase shifters 1, 2, 3, 4 at 0 or 90 ° and one control bit, but with phase dividers / combiners 5a, 5b. In this case, a right circular polarization is obtained with a 90 ° phase shift on phase shifters 1, 3 and 0 ° on phase shifters 2, 4; and conversely, a polarization left circular is obtained with a 90 ° phase shift on phase shifters 2, 4 and 0 ° on phase shifters 1 and 3.

In Figure 4 we see another variant of E / R circuit according to the invention, or we have added to the circuit of the invention according to FIG. 3, the characteristics of the prior art according to Figure 2. More specifically, to decrease losses associated with switches in position 35a, 35b in figure 3, we have inserted circulators 52a, 52b in their place. This corresponds to one of the variants already discussed in the context of Figure 3. But on top of that, we inserted, in the reception channel, protection against possible reflections from the mismatches of the antenna. This protection is provided by switches 32a, 32b which are controlled by the clock for connect the inputs of the low noise amplifiers to the mass during transmission. An additional benefit is obtained by inserting a second circulator on each reception channel 33a, 33b for evacuating reflections possible from these switches 32a, 32b when they are closed because any reflections there would reduce the transmission power, especially if the direct and reflected signals combine in opposition to phase.

In Figure 5, we see schematically the most general realization of a synthesis circuit of polarization according to the invention. Indeed, this circuit is able to synthesize any polarization: linear, circular or elliptical with axes arbitrary, and can easily pass among these possibilities by means of commands supplied to its phase shifters 27a, 27b and its attenuators 28a, 28b almost continuously controllable. In this way, the instant orientation of the polarization vector is given by the relative phases coming from the phase shifters 27a, 27b which can take arbitrary and variable values over time, and the relative amplitude of the signals passing through variable attenuators can also take arbitrary and time-varying values, for determine the length of each of the two projections of the vector of the electric field, on the two orthogonal axes, corresponding to the polarizations generated on each of the access to radiant elements. The polarization will be linear when this phase shift is 180 °; it will be circular if it is +/- 90 °, and that the attenuations of the two channels are equal; we have an elliptical polarization in the case where the phase shift takes a different value, or linear, or good if the attenuations of the two channels are different.

In a practical example of this variant, as shown in figure 5, we have chosen a design using two variable attenuators 28a, 28b and two variable phase shifters 27a, 27b. We could have, of course use four of each with the configuration of FIG. 3 or FIG. 4, placing them, one of each, in boxes 1, 2, 3, 4 of these figures 3 and 4. In the configuration of figure 5 so we added some circulators 7a, 7b for separating the emission signals from the reception signals according to their direction of propagation in the circuit.

The transmission signal arrives at a power divider, and is sent to the two circulators 7a, 7b after division in phase. Then we have two E / R circuits in parallel, the description of which conforms to the descriptions in the previous figures, with the same references representing the same elements in all the figures. These two circuits deliver signals on two orthogonal accesses to the radiating source S _ij , with a relative phase and a relative amplitude which are determined by the controllable attenuators and phase shifters 28a, 28b, and 27a, 27b respectively.

Conversely, the reception signal from the source S _ij is probed by the two orthogonal ports, and the two received signals are amplified separately by the low noise amplifiers 30a, 30b. Their relative amplitude and their relative phase are adjusted by the controllable attenuators and phase shifters 28a, 28b and 27a, 27b respectively, according to the polarization of the received wave that one wishes to look at. These signals are then transmitted, via the circulators 7a, 7b to separate reception channels for signal processing in an appropriate computer (not shown).

This possibility of synthesis of a polarization arbitrary allows to obtain a self-adapting antenna, that is to say who can reconfigure to take into account a environment polluted by parasitic emissions intentional or not. The principle consists in measuring the dominant polarization of the radio environment at the operating frequency band of the equipment, putting the attenuators and phase shifters in a state of reference. The polarization of the emission is then chosen orthogonal to this dominant polarization. This mode may allow operation significantly improved in the presence of interference intentional with stationary polarization, or in the case where unwanted specular reflections mask a target radar of small equivalent surface, but not presenting a specularity.

In Figure 6, we have shown schematic a simplest configuration of a circuit microwave according to the invention. Following the characteristics of components used, this circuit will suitable for either transmitting or receiving signals polarized synthesis microwave. Such circuits find applications for radar antennas multistatic, for example, or in the antennas of telecommunications.

According to a first example of implementation of the circuit shown in figure 6, this circuit would be intended for amplification of signals for transmission. Low signal level arriving at the input of the variable attenuator 28, and attenuated by the latter, then propagated through a controllable phase shifter 27, so as to adjust the phase and the signal amplitude of this circuit with respect to its position in the array of radiating elements of the antenna network (not shown). As in the previous figures (and in particular Figure 3), element 5 is a divider of power, either a phase divider or a coupler hybrid with a 90 ° phase shift.

Blocks 1, 3 represent phase shifters controllable at 0, 1, or 2 bits, having values of phase shift of 0-0 °, 0-90 °, or 0-180 °, as in the Description of Figure 3. The construction of the circuit is strictly analogous to the description given for this Figure 3, with regard to the transmission channel. The components 20a, 20b are then amplifiers of power, which feed through channels inclined at 45 ° from the horizontal, the patches S k / ij.

In a second example of implementation of the circuit shown in figure 6, this circuit would be intended for amplification of signals for reception. A signal from very low level arriving at the radiating element S k / ij is routed by access roads inclined at 45 ° to horizontally, towards the low noise amplifiers 20a, 20b. Then the amplified signals will be phase shifted by 0, 90, 180, or 270 ° (= -90 °) by controllable phase shifters 1, 3 to 0, 1, or 2 command bits. After phase shift, the signals will be combined either in phase or with a 90 ° phase shift, thanks to the means 5 which are either a phase combiner, i.e. a hybrid coupler with a 90 ° phase shift between its two inputs. The signals will then routed through a phase shift and attenuation controllable according to the position of the radiating element in the network antenna network.

Of course, the controllable phase shifters can to be almost continuous, as in the case of previous figure 5, to allow greater flexibility in the synthesis of polarizations, if necessary.

In the examples shown in the figures, we have used to simplify the presentation, only the square patch as a radiant source, this patch being oriented with its horizontal and vertical diagonals. But it is good understood that the invention is situated at the level of the E / R circuit, and that the radiating sources can be of different types or different orientations. For example, patches can be oriented with the horizontal sides and vertical, and fed by orthogonal accesses according to their diagonals. As already mentioned, the lines of propagation leading to accesses can also be different types, for example coaxial, microstrip, triplate, ...

The radiating sources can also be annular slots photo-etched in an upper ground plane, excited by lines directed at 45 ° relative to the directions H and V, located in a lower plane, ie on the other face of the substrate comprising the plane mass and slots; either on a second suspended substrate, the two substrates kept spaced from each other by spacers or by a material having low microwave losses, such as foam or honeycomb. Such constructions of networks of radiating sources and their power supplies are well known to those skilled in the art, and are described for example in the Proceedings of Military Microwaves 1992 , "Antennas for space scatteromertes and SARS", by R. Petersson , the description of the prior art of which forms an integral part of the present application.

Other more conventional elements can also be used, such as horns with square, circular or hexagonal opening, which will be excited in two directions inclined by 45 ° with respect to the H and V polarizations. Another example of a radiating element, for to obtain a wider bandwidth, is the flared slot (in English: "notch antenna"), described in detail in the Proceedings of Antenna and Propagation Symposium, 1974, IEEE, "A broadband stripline array element", by LR Lewis et al ., of which the description of the prior art forms an integral part of the present application.

The circuits presented as an example in the figures can also be made according to different technologies without departing from the scope of the invention: if MMIC is a preferred technology for its low mass and congestion, as well as these remaining production costs reasonable for large series production, a higher transmitting power can be tolerated in using circulators instead of switches integrated downstream of the power amplifiers. The size and mass of these circulators are higher but the losses lower than the losses of MMIC switches.

However, some performances can be optimized by a hybrid technology implementation: discrete amplifiers can provide more power higher for emission, and better noise factors for reception, that the integrated amplifiers of the MMIC technology. According to the realization technology chosen, different options discussed in the description of Figure 3 will be preferred to others. The ultimate performance can be optimized depending on the mission of the antenna, according to the numerous criteria mentioned. In all cases, the use of the E / R circuit according to the invention provides a significant improvement in performances obtained, in particular in the signal ratio useful on noise.

Claims (18)

1. Alternate transmit/receive (T/R) microwave circuit for variable synthesized polarization array antennas adapted to supply excitation signals for at least two orthogonal polarizations to array elements (S^k _ij) via two respective input/output channels fed by two respective transmit power amplifiers (29a, 29b) and to receive at least two signals having orthogonal polarizations detected by the same array elements and feeding two low-noise receive amplifiers (30a, 30b); characterized in that said transmit/receive (T/R) microwave circuit further comprises at least one controllable phase shifter (1, 3) on one of said transmit channels and at least one controllable phase shifter (2, 4) on one of said receive channels, and in that the two power amplifiers (29a, 29b) operate simultaneously during transmission and in that the two low-noise amplifiers (30a, 30b) operate simultaneously during reception.
2. Circuit according to claim 1 characterized in that said two input/output channels are connected to array elements (S^k _ij) to generate polarizations inclined at 45° to the horizontal so that by adjusting the phase-shifters (1, 2, 3, 4) it is possible to synthesize the standard horizontal H or vertical V polarization.
3. T/R circuit according to claim 1 or claim 2 characterized in that said two power amplifiers (29a, 29b) are fed by an in-phase power divider (5a) to facilitate synthesis of orthogonal linear polarizations.
4. T/R circuit according to any one of claims 1 to 3 characterized in that said two power amplifiers (29a, 29b) are fed by a hybrid coupler (5a) having two outputs with a relative phase difference of 90° to facilitate synthesis of circular polarizations.
5. T/R circuit according to any one of claims 1 to 4 characterized in that said phase-shifters are one-bit digital controllable phase-shifters and the value of said one bit represents either 0° or 180°.
6. T/R circuit according to any one of claims 1 to 4 characterized in that said phase-shifters are two-bit digital controllable phase-shifters and the value of a first bit represents either 0° or 180° and the value of the second bit represents either 0° or 90° so that any of the following four standard polarizations can be synthesized: linear H or V, right or left circular.
7. T/R circuit according to any one of claims 1 to 4 characterized in that said phase-shifters are one-bit digital controllable phase-shifters and the value of said one bit represents either 0° or 90°.
8. T/R circuit according to any one of claims 1 to 7 characterized in that a controllable attenuator is used to vary the gain of at least one of said power amplifiers.
9. T/R circuit according to any one of claims 1 to 8 characterized in that a controllable attenuator is used to vary the gain of at least one of said low-noise amplifiers.
10. T/R circuit according to any one of claims 1 to 9 characterized in that it further comprises at least two quasi-continuously controllable phase-shifters and at least two quasi-continuously controllable attenuators for synthesizing any linear, circular or elliptical polarization.
11. T/R circuit according to claim 10 characterized in that said phase-shifters and said attenuators are controllable quasi-continuously in the analog domain.
12. T/R circuit according to claim 10 characterized in that said phase-shifters and said attenuators are controllable quasi-continuously in the digital domain using a large number of bits for synthesizing any linear, circular or elliptical polarization.
13. Array antenna with variable synthesized polarization at the array elements characterized in that it comprises transmit/receive circuits according to any one of claims 1 to 12.
14. Array antenna according to claim 13 characterized in that it comprises printed circuit (patch) type array elements.
15. Array antenna according to claim 13 characterized in that it comprises array elements in the form of annular slots photo-chemically etched on one side of a dielectric substrate having low losses at microwave frequencies and excited by photo-chemically etched lines on the opposite side of said substrate.
16. Array antenna according to claim 15 characterized in that said slots are excited by lines photo-chemically etched on a suspended substrate.
17. T/R circuit according to any one of claims 1 to 12 characterized in that said circuit is implemented in the MMIC technology.
18. Array antenna according to any one of claims 13 to 16 characterized in that it is an adaptive polarization antenna.

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