EP0563765A2 - Circuit for producing an electrical voltage reference value depending on an electrical control voltage - Google Patents

Circuit for producing an electrical voltage reference value depending on an electrical control voltage Download PDF

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Publication number
EP0563765A2
EP0563765A2 EP93104805A EP93104805A EP0563765A2 EP 0563765 A2 EP0563765 A2 EP 0563765A2 EP 93104805 A EP93104805 A EP 93104805A EP 93104805 A EP93104805 A EP 93104805A EP 0563765 A2 EP0563765 A2 EP 0563765A2
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EP
European Patent Office
Prior art keywords
voltage
comparator
resistor
input
output
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Granted
Application number
EP93104805A
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German (de)
French (fr)
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EP0563765A3 (en
EP0563765B1 (en
Inventor
Fredy Kamber
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STARKSTROM-ELEKTRONIK AG
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STARKSTROM-ELEKTRONIK AG
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    • G—PHYSICS
    • G05—CONTROLLING; REGULATING
    • G05F—SYSTEMS FOR REGULATING ELECTRIC OR MAGNETIC VARIABLES
    • G05F1/00—Automatic systems in which deviations of an electric quantity from one or more predetermined values are detected at the output of the system and fed back to a device within the system to restore the detected quantity to its predetermined value or values, i.e. retroactive systems
    • G05F1/10—Regulating voltage or current
    • G05F1/46—Regulating voltage or current wherein the variable actually regulated by the final control device is dc
    • G05F1/462—Regulating voltage or current wherein the variable actually regulated by the final control device is dc as a function of the requirements of the load, e.g. delay, temperature, specific voltage/current characteristic
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B39/00—Circuit arrangements or apparatus for operating incandescent light sources
    • H05B39/04—Controlling
    • H05B39/041—Controlling the light-intensity of the source
    • H05B39/042—Controlling the light-intensity of the source by measuring the incident light
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/36—Controlling
    • H05B41/38—Controlling the intensity of light
    • H05B41/39—Controlling the intensity of light continuously
    • H05B41/392—Controlling the intensity of light continuously using semiconductor devices, e.g. thyristor
    • H05B41/3921—Controlling the intensity of light continuously using semiconductor devices, e.g. thyristor with possibility of light intensity variations

Definitions

  • the present invention relates to a circuit arrangement for generating an electrical voltage setpoint in a non-linear manner as a function of an electrical control voltage.
  • Circuit arrangements of this type are known and have been proposed and used for numerous applications. For higher demands on precision and reproducibility, however, the known circuit arrangements are complex and correspondingly expensive.
  • the object of the invention is to provide a circuit arrangement of the type mentioned which, in a simple construction, requires only a few common electrical components and nevertheless produces a stable, reproducible voltage setpoint and is particularly suitable for use in an electronic ballast for regulating the brightness of low-pressure gas discharge lamps.
  • the task is to generate a target voltage to be compared with a reference voltage, hereinafter referred to as the target value, as a function of an externally adjustable control voltage.
  • the comparison value of the setpoint and the reference voltage then serves to control an inverter bridge fed by an AC or DC network in such a way that the brightness of the discharge lamp connected to the inverter bridge changes in accordance with the value of the externally set control voltage. Since the brightness perception of the human eye is quasi-logarithmic, it is inappropriate to use a linearly changing control variable as the basis for brightness control of the discharge lamp.
  • FIG. 1 shows a first circuit part of the circuit arrangement according to the invention.
  • This circuit part has a positive supply voltage line labeled + and a zero line labeled 0.
  • This stabilized DC voltage supply of approximately 12 volts is independent, that is to say galvanically isolated from other parts of the device under consideration.
  • the supply voltage is therefore generated, for example, by means of a transformer connected to an alternating current source, to the secondary winding of which a rectifier and stabilizing circuit is connected. Since such a feed circuit is irrelevant here, it is not shown in FIG. 1.
  • the first circuit part of FIG. 1 also has two terminals 1 and 2, to which an external, adjustable burden 3 can be connected via a control line, not shown.
  • a voltage divider is formed together with a resistor 4 connected to the positive feed line.
  • the resistor 4 is dimensioned such that a voltage which varies between 1 and 10 volts, depending on the setting of the burden 4, results in the voltage divider point 5.
  • the voltage divider point 5 is connected via a high-resistance resistor 6 to the positive input of a first comparator 7, this input also being connected to the zero line via a capacitor 8.
  • the resistor 6 and the capacitor 8 form a protective circuit of the comparator input against unintentional high voltages, for example if the AC mains voltage is accidentally connected to the terminal 2. This ensures that the control cable connected to terminals 1 and 2 has a constant voltage.
  • a second comparator 9 also has its positive input connected to a voltage divider formed by resistors 10 and 11.
  • the negative input of the comparator 9 is connected to the positive feed line via a resistor 12 and to the zero line via a capacitor 13.
  • the output of the comparator 9 is connected on the one hand via a resistor 14 to the positive comparator input and via the series connection of a diode 15 and a resistor 16 to the negative comparator input. Finally, the negative input of the comparator 9 is still connected to the negative input of the comparator 7.
  • resistors 10 and 11 are relatively high, for example 100 kQ and 330 kQ, respectively.
  • the resistor 12 also has a value close to 100 kQ.
  • the resistor 14 is relatively low-resistance, for example 6.8 kQ.
  • the resistor 16 is very low, for example 100 Q.
  • the positive input of the comparator 9 When the supply voltage is first applied to the positive supply line and the zero line, the positive input of the comparator 9 immediately assumes the value determined by the resistors 10 and 11. The capacitor 13 is charged via the resistor 12 with the time constant of these two components, so that the voltage at the negative input of the comparator increases exponentially. No current flows through the resistor 14, the diode 15 and the resistor, since the voltage at the positive input of the comparator 9 is higher than that at the negative comparator input and because the comparator output is open.
  • the comparator 9 switches, that is, its output practically assumes the voltage of the zero line.
  • the capacitor 13 discharges rapidly via the resistor 16 and the diode 15, while there is a very low voltage at the positive comparator input, since the resistor 14 is connected in parallel to the resistor 11 in this state.
  • the capacitor 13 can therefore discharge to this low voltage. If the voltage at the negative comparator input drops further, that at the positive comparator input is relatively higher.
  • the comparator 9 thus returns to the blocking state with an open output, which corresponds to the initial state, so that the capacitor 13 is charged again via the resistor 12.
  • the transmission-side diode 18 of the optocoupler 19 is accordingly active as long as each of the exponentially increasing pulses generated by the comparator 9 has a lower voltage than the control voltage generated by the set burden 3.
  • the width of the square-wave pulses emitted by the active diode 18 is a measure of the magnitude of the control voltage. However, as intended, the relationship between the width of the pulses and the magnitude of the control voltage is not linear. Because of the exponential course of the pulses of the comparator 9 controlling the diode 18, the width of the diode pulses for changes in the small control voltages of the burden 3 increases more slowly than for changes in large control voltages.
  • comparators 7 and 9 are used in the exemplary embodiment described. However, it is also possible to provide other semiconductor components and other circuits for this purpose, for example operational amplifiers, etc.
  • a second circuit part, shown in FIG. 2, is provided to receive the square-wave pulses transmitted by the diode 18 of FIG. 1, to additionally deform and smooth them non-linearly, in order to generate a nominal value of low ripple.
  • a receiver-side transistor 21 of the optocoupler 19 is connected via a resistor 22 on the one hand to a positive voltage line + and on the other hand connected to the associated zero line 0.
  • the control electrode of a field-effect transistor 23 is connected to the transistor 21 of the optocoupler 19, the drain electrode of which is also fed by the positive voltage line via a resistor 24.
  • the drain electrode is connected via a further resistor 25 to a terminal 26 at which, as explained below, the desired setpoint occurs. Terminal 26 is connected to the zero line via a capacitor 27.
  • the diode 18 of the optocoupler 19 (FIG. 1) is conductive, a current also flows through the transistor 21 of the optocoupler 19 (FIG. 2). Then the field effect transistor 23 is blocked, so that the capacitor charges via the series connection of the resistors 24 and 25. If the diode 18 and thus also the transistor 21 of the optocoupler 19 are subsequently blocked, the field effect transistor 23 becomes conductive, so that the capacitor 27 can discharge via the field effect transistor 23 and the resistor 25. There the resistor 25 has an approximately four times smaller value than the sum of the resistors 24 and 25, the capacitor 27 discharges faster than its charging.
  • the voltage at the terminal 26 rises significantly faster at a high control voltage than at a low control voltage, which significantly improves the desired quasi-logarithmic characteristic of the dependence of the voltage at the terminal 26 on the control voltage.
  • the capacitor 27 has a relatively large capacitance (for example approximately 3 ⁇ F)
  • the ripple in the voltage at the terminal 26 is very low.
  • the voltage at terminal 26 represents the desired setpoint with a quasi-logarithmic dependence on the adjustable control voltage provided by burden 3 in FIG. 1.
  • the present circuit arrangement generates reproducible voltage setpoints.
  • the brightness of a large number of discharge lamps, the ballasts of which each have an existing circuit arrangement can be controlled without noticeable differences in the brightness set for the individual discharge lamps.

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  • Engineering & Computer Science (AREA)
  • Physics & Mathematics (AREA)
  • Electromagnetism (AREA)
  • General Physics & Mathematics (AREA)
  • Radar, Positioning & Navigation (AREA)
  • Automation & Control Theory (AREA)
  • Circuit Arrangements For Discharge Lamps (AREA)
  • Details Of Television Scanning (AREA)
  • Relay Circuits (AREA)
  • Electrophonic Musical Instruments (AREA)
  • Emergency Protection Circuit Devices (AREA)
  • Oscillators With Electromechanical Resonators (AREA)
  • Control Of Electrical Variables (AREA)

Abstract

An RC section (12, 13) connected between feed lines (+, 0) is connected to the first input of a comparator (9). The second input of the comparator (9) is connected to a fixed voltage divider (10, 11). The output of the comparator (9) is connected to the first input via another resistor (16). The capacitor (13) of the RC section is charged up via the resistor (12) of the RC section until the voltage at the first input reaches the fixed voltage at the second input. The output of the comparator (9) then discharges the capacitor (13) via the further resistor (16) whereupon the capacitor (13) is charged up again. The periodic exponential capacitor voltage is supplied to the first input of a further comparator (7), at the second input of which the adjustable control voltage is present. When the voltages are equal, a voltage change occurs at the output of the comparator (7) which is transferred to a switching transistor (23) via an optocoupler (19). The signal present at its output and smoothed forms a voltage reference value which quasi-logarithmically depends on the settings of the control voltage and can be used, for example, for controlling the brightness of low-voltage gas discharge lamps. <IMAGE>

Description

Die vorliegende Erfindung bezieht sich auf eine Schaltungsanordnung zur Erzeugung eines elektrischen Spannungs-Sollwertes in nichtlinearer Abhängigkeit von einer elektrischen Steuerspannung.The present invention relates to a circuit arrangement for generating an electrical voltage setpoint in a non-linear manner as a function of an electrical control voltage.

Schaltungsanordnungen dieser Art sind bekannt und für zahlreiche Anwendungen vorgeschlagen und in Gebrauch. Für höhere Ansprüche an Präzision und Reproduzierbarkeit sind die bekannten Schaltungsanordnungen jedoch aufwendig und entsprechend kostspielig.Circuit arrangements of this type are known and have been proposed and used for numerous applications. For higher demands on precision and reproducibility, however, the known circuit arrangements are complex and correspondingly expensive.

Aufgabe der Erfindung ist, eine Schaltungsanordnung der eingangs genannten Art zu schaffen, welche in einfachem Aufbau nur wenige gängige elektrische Bauteile benötigt und trotzdem einen stabilen, reproduzierbaren Spannungs-Sollwert erzeugt und sich insbesondere zur Verwendung in einem elektronischen Vorschaltgerät für die Helligkeitsregulierung von Niederdruckgasentladungslampen eignet.The object of the invention is to provide a circuit arrangement of the type mentioned which, in a simple construction, requires only a few common electrical components and nevertheless produces a stable, reproducible voltage setpoint and is particularly suitable for use in an electronic ballast for regulating the brightness of low-pressure gas discharge lamps.

Zur Lösung dieser Aufgabe weist die Erfindung die im Patentanspruch 1 angeführten Merkmale auf.To achieve this object, the invention has the features stated in claim 1.

Einzelheiten des Erfindungsgegenstandes werden anhand der Zeichnung beispielsweise erläutert. Es zeigen:

  • Fig. 1 ein Schaltungsschema eines ersten Schaltungsteils der Schaltungsanordnung, und
  • Fig. 2 ein Schaltungsschema eines zweiten Schaltungsteils, der mit dem ersten Schaltungsteil in Wirkverbindung steht.
Details of the subject matter of the invention are explained for example with reference to the drawing. Show it:
  • Fig. 1 is a circuit diagram of a first circuit part of the circuit arrangement, and
  • Fig. 2 is a circuit diagram of a second circuit part which is operatively connected to the first circuit part.

Die erfindungsgemässe Schaltungsanordnung wird nachstehend in der Anwendung in einem elektronischen Vorschaltgerät für eine Niederdruckgasentladungslampe erläutert. In einer bestimmten Ausführungsform eines solchen Geräts stellt sich die Aufgabe, eine mit einer Referenzspannung zu vergleichende Sollspannung, nachfolgend Sollwert genannt, in Abhängigkeit von einer extern einstellbaren Steuerspannung zu erzeugen. Der Vergleichswert des Sollwerts und der Referenzspannung dient dann dazu, eine von einem Wechselstrom- oder Gleichstromnetz gespeiste Wechselrichterbrücke so zu steuern, dass die Helligkeit der an die Wechselrichterbrücke angeschlossene Entladungslampe entsprechend dem Wert der extern eingestellten Steuerspannung ändert. Da das Helligkeitsempfinden des menschlichen Auges quasi-logarithmisch ist, ist es unzweckmässig, einer Helligkeitssteuerung der Entladungslampe eine linear ändernde Steuergrösse zugrunde zu legen. Dies ergibt die nachteilige Charakteristik, dass bei kleiner Helligkeit eine sehr kleine Aenderung der Steuergrösse bereits eine starke Helligkeitsänderung hervorruft, während bei grosser Helligkeit eine grosse Aenderung der Steuergrösse erforderlich ist, um eine auch nur geringe Helligkeitsänderung zu bewirken. Durch die nachfolgend erläuterte Schaltungsanordnung kann diesem Mangel abgeholfen werden.The circuit arrangement according to the invention is explained below in the application in an electronic ballast for a low-pressure gas discharge lamp. In a specific embodiment of such a device, the task is to generate a target voltage to be compared with a reference voltage, hereinafter referred to as the target value, as a function of an externally adjustable control voltage. The comparison value of the setpoint and the reference voltage then serves to control an inverter bridge fed by an AC or DC network in such a way that the brightness of the discharge lamp connected to the inverter bridge changes in accordance with the value of the externally set control voltage. Since the brightness perception of the human eye is quasi-logarithmic, it is inappropriate to use a linearly changing control variable as the basis for brightness control of the discharge lamp. This results in the disadvantageous characteristic that with a low brightness, a very small change in the control variable already causes a strong change in brightness, while with a high brightness a large change in the control variable is required in order to bring about even a slight change in brightness. This defect can be remedied by the circuit arrangement explained below.

In Fig. 1 ist ein erster Schaltungsteil der erfindungsgemässen Schaltungsanordnung dargestellt. Dieser Schaltungsteil weist eine mit + bezeichnete positive Speisespannungsleitung und eine mit 0 bezeichnete Null-Leitung auf. Diese stabilisierte Gleichspannungsspeisung von etwa 12 Volt ist unabhängig, das heisst galvanisch getrennt von anderen Teilen des in Betracht gezogenen Geräts. Die Speisespannung wird deshalb zum Beispiel mittels eines an eine Wechselstromquelle angeschlossenen Transformators erzeugt, mit dessen Sekundärwicklung eine Gleichrichter- und Stabilisierschaltung verbunden ist. Da eine solche Speiseschaltung hier ohne Belang ist, ist sie in Fig. 1 nicht dargestellt.1 shows a first circuit part of the circuit arrangement according to the invention. This circuit part has a positive supply voltage line labeled + and a zero line labeled 0. This stabilized DC voltage supply of approximately 12 volts is independent, that is to say galvanically isolated from other parts of the device under consideration. The supply voltage is therefore generated, for example, by means of a transformer connected to an alternating current source, to the secondary winding of which a rectifier and stabilizing circuit is connected. Since such a feed circuit is irrelevant here, it is not shown in FIG. 1.

Der erste Schaltungsteil der Fig. 1 weist zudem zwei Klemmen 1 und 2 auf, welchen eine externe, einstellbare Bürde 3 über eine nicht dargestellte Steuerleitung anschliessbar ist. Bei angeschlossener Bürde 3 ist zusammen mit einem an die positive Speiseleitung angeschlossenen Widerstand 4 ein Spannungsteiler gebildet. Der Widerstand 4 ist so dimensioniert, dass sich im Spannungsteilerpunkt 5 eine je nach Einstellung der Bürde 4 zwischen 1 und 10 Volt veränderliche Spannung ergibt. Der Spannungsteilerpunkt 5 ist über einen hochohmigen Widerstand 6 mit dem positiven Eingang eines ersten Komparators 7 verbunden, wobei dieser Eingang zudem über einen Kondensator 8 mit der Null-Leitung verbunden ist. Der Widerstand 6 und der Kondensator 8 bilden eine Schutzschaltung des Komparatoreingangs gegenüber unbeabsichtigten hohen Spannungen, beispielsweise wenn an die Klemme 2 versehentlich die Netzwechselspannung angeschlossen wird. Somit ist die Netzspannungsfestigkeit der an die Klemmen 1 und 2 angeschlossenen Steuerleitung gewährleistet.The first circuit part of FIG. 1 also has two terminals 1 and 2, to which an external, adjustable burden 3 can be connected via a control line, not shown. When the load 3 is connected, a voltage divider is formed together with a resistor 4 connected to the positive feed line. The resistor 4 is dimensioned such that a voltage which varies between 1 and 10 volts, depending on the setting of the burden 4, results in the voltage divider point 5. The voltage divider point 5 is connected via a high-resistance resistor 6 to the positive input of a first comparator 7, this input also being connected to the zero line via a capacitor 8. The resistor 6 and the capacitor 8 form a protective circuit of the comparator input against unintentional high voltages, for example if the AC mains voltage is accidentally connected to the terminal 2. This ensures that the control cable connected to terminals 1 and 2 has a constant voltage.

Ein zweiter Komparator 9 hat seinen positiven Eingang ebenfalls an einen durch Widerstände 10 und 11 gebildeten Spannungsteiler angeschlossen. Der negative Eingang des Komparators 9 ist über einen Widerstand 12 mit der positiven Speiseleitung und über einen Kondensator 13 mit der Null-Leitung verbunden. Der Ausgang des Komparators 9 ist einerseits über einen Widerstand 14 mit dem positiven Komparatoreingang und über die Reihenschaltung einer Diode 15 und eines Widerstands 16 mit dem negativen Komparatoreingang verbunden. Schliesslich ist der negative Eingang des Komparators 9 noch mit dem negativen Eingang des Komparators 7 verbunden.A second comparator 9 also has its positive input connected to a voltage divider formed by resistors 10 and 11. The negative input of the comparator 9 is connected to the positive feed line via a resistor 12 and to the zero line via a capacitor 13. The output of the comparator 9 is connected on the one hand via a resistor 14 to the positive comparator input and via the series connection of a diode 15 and a resistor 16 to the negative comparator input. Finally, the negative input of the comparator 9 is still connected to the negative input of the comparator 7.

Die Werte der Widerstände 10 und 11 sind relativ hoch, beispielsweise 100 kQ bzw. 330 kQ. Der Widerstand 12 hat ebenfalls einen Wert nahe 100 kQ. Der Widerstand 14 ist relativ niederohmig, zum Beispiel 6,8 kQ. Der Widerstand 16 ist sehr niederohmig, beispielsweise 100 Q.The values of resistors 10 and 11 are relatively high, for example 100 kQ and 330 kQ, respectively. The resistor 12 also has a value close to 100 kQ. The resistor 14 is relatively low-resistance, for example 6.8 kQ. The resistor 16 is very low, for example 100 Q.

Beim ersten Anlegen der Speisespannung an die positive Speiseleitung und die Null-Leitung nimmt der positive Eingang des Komparators 9 sofort den durch die Widerstände 10 und 11 bestimmten Wert an. Ueber den Widerstand 12 wird der Kondensator 13 mit der Zeitkonstante dieser beiden Bauteile geladen, so dass die Spannung am negativen Eingang des Komparators exponentiell ansteigt. Ueber den Widerstand 14, die Diode 15 und den Widerstand fliesst kein Strom, da die Spannung am positiven Eingang des Komparators 9 höher als diejenige am negativen Komparatoreingang und da der Komparatorausgang offen ist.When the supply voltage is first applied to the positive supply line and the zero line, the positive input of the comparator 9 immediately assumes the value determined by the resistors 10 and 11. The capacitor 13 is charged via the resistor 12 with the time constant of these two components, so that the voltage at the negative input of the comparator increases exponentially. No current flows through the resistor 14, the diode 15 and the resistor, since the voltage at the positive input of the comparator 9 is higher than that at the negative comparator input and because the comparator output is open.

Sobald die Spannung am negativen Komparatoreingang diejenige am positiven Komparatoreingang erreicht, schaltet der Komparator 9, das heisst, sein Ausgang nimmt praktisch die Spannung der Null-Leitung an. Dadurch entlädt sich der Kondensator 13 rasch über den Widerstand 16 und die Diode 15, während am positiven Komparatoreingang eine sehr niedrige Spannung liegt, da der Widerstand 14 in diesem Zustand parallel zum Widerstand 11 geschaltet ist. Der Kondensator 13 kann sich also bis auf diese niedrige Spannung entladen. Sinkt die Spannung am negativen Komparatoreingang weiter, so ist diejenige am positiven Komparatoreingang relativ höher. Der Komparator 9 gelangt somit erneut in den Sperrzustand mit offenem Ausgang, welcher dem anfänglichen Zustand entspricht, so dass der Kondensator 13 erneut über den Widerstand 12 geladen wird.As soon as the voltage at the negative comparator input reaches that at the positive comparator input, the comparator 9 switches, that is, its output practically assumes the voltage of the zero line. As a result, the capacitor 13 discharges rapidly via the resistor 16 and the diode 15, while there is a very low voltage at the positive comparator input, since the resistor 14 is connected in parallel to the resistor 11 in this state. The capacitor 13 can therefore discharge to this low voltage. If the voltage at the negative comparator input drops further, that at the positive comparator input is relatively higher. The comparator 9 thus returns to the blocking state with an open output, which corresponds to the initial state, so that the capacitor 13 is charged again via the resistor 12.

Am negativen Eingang des Komparators 7 liegt demnach eine Folge von exponentiell ansteigenden Impulsen mit steiler Flanke, die jedoch nicht ganz auf den Wert null absinken, da sich der Kondensator 13 wegen der Parallelschaltung der Widerstaände 11 und 14 nicht vollständig entladen kann. Solange die Spannung am negativen Eingang des Komparators 7 kleiner ist als die durch die Bürde 3 eingestellte Spannung am positiven Eingang, ist der Ausgang des Komparators offen. An diesen Ausgang ist die Reihenschaltung eines Widerstands 17 und der Diode 18 eines Optokopplers 19 angeschlossen. Bei offenem Ausgang des Komparators 7 fliesst demnach ein Strom durch die Diode 18. Sobald die am negativen Eingang des Komparators exponentiell ansteigende Spannung den Wert der Spannung am positiven Eingang erreicht, schaltet der Komparator 7, das heisst, sein Ausgang nimmt praktisch die Nullspannung an. Dadurch wird die Diode 18 kurzgeschlossen, und der durch den Widerstand 17 fliessende Strom gelangt direkt zur Null-Leitung.At the negative input of the comparator 7 there is therefore a sequence of exponentially rising pulses with a steep flank, which, however, do not drop completely to zero, since the capacitor 13 cannot discharge completely due to the parallel connection of the resistors 11 and 14. As long as the voltage at the negative input of the comparator 7 is less than the voltage at the positive input set by the load 3, the output of the comparator is open. The series connection of a resistor 17 and the diode 18 of an optocoupler 19 is connected to this output. With the output of the comparator 7 open, a current therefore flows through the diode 18. As soon as the voltage at the negative input of the comparator increases exponentially to the value of the voltage at the positive input, the comparator 7 switches, that is to say its output practically assumes the zero voltage. As a result, the diode 18 is short-circuited, and the current flowing through the resistor 17 goes directly to the zero line.

Die sendeseitige Diode 18 des Optokopplers 19 ist demnach aktiv, solange jeder der mittels des Komparators 9 erzeugten, exponentiell ansteigenden Impulse eine kleinere Spannung hat als die durch die eingestellte Bürde 3 erzeugte Steuerspannung. Die Breite der von der aktiven Diode 18 emittierten Rechteckimpulse ist ein Mass für die Grösse der Steuerspannung. Der Zusammenhang zwischen der Breite der Impulse und der Grösse der Steuerspannung ist jedoch, wie beabsichtigt, nicht linear. Wegen des exponentiellen Verlaufs der die Diode 18 steuernden Impulse des Komparators 9 nimmt die Breite der Diodenimpulse für Aenderungen kleiner Steuerspannungen der Bürde 3 langsamer zu als für Aenderungen grosser Steuerspannungen. Mit anderen Worten ist für eine bestimmte Breitenänderung der Diodenimpulse bei kleinen Steuerspannungen eine grössere Steuerspannungsänderung erforderlich als bei grossen Steuerspannungen. Dies ist bereits eine quasi-logarithmische Charakteristik eines Sollwertes, der im vorliegenden Beispiel für die Helligkeit der gesteuerten Entladungslampe massgebend ist.The transmission-side diode 18 of the optocoupler 19 is accordingly active as long as each of the exponentially increasing pulses generated by the comparator 9 has a lower voltage than the control voltage generated by the set burden 3. The width of the square-wave pulses emitted by the active diode 18 is a measure of the magnitude of the control voltage. However, as intended, the relationship between the width of the pulses and the magnitude of the control voltage is not linear. Because of the exponential course of the pulses of the comparator 9 controlling the diode 18, the width of the diode pulses for changes in the small control voltages of the burden 3 increases more slowly than for changes in large control voltages. In other words, a larger change in the control voltage is required for a certain change in the width of the diode pulses for small control voltages than for large control voltages. This is already a quasi-logarithmic characteristic of a target value, which in the present example is decisive for the brightness of the controlled discharge lamp.

Zur Erzeugung der exponentiell ansteigenden Impulse und zu ihrem Vergleich mit der jeweils eingestellten Steuerspannung sind beim beschriebenen Ausführungsbeispiel die Komparatoren 7 und 9 verwendet. Es ist jedoch auch möglich, hierzu andere Halbleiter-Bauelemente und andere Schaltungen vorzusehen, beispielsweise Operationsverstärker usw.To generate the exponentially increasing pulses and to compare them with the respectively set control voltage, the comparators 7 and 9 are used in the exemplary embodiment described. However, it is also possible to provide other semiconductor components and other circuits for this purpose, for example operational amplifiers, etc.

Ein zweiter, in Fig. 2 dargestellter Schaltungsteil ist dazu vorgesehen, die von der Diode 18 der Fig. 1 gesendeten Rechteckimpulse zu empfangen, zusätzlich nichtlinear zu verformen und zu glätten, um einen Sollwert geringer Welligkeit zu erzeugen. Ein empfangsseitiger Transistor 21 des Optokopplers 19 ist über einen Widerstand 22 einerseits an eine positive Spannungsleitung + angeschlossen und andererseits mit der zugehörigen Null-Leitung 0 verbunden. An den Transistor 21 des Optokopplers 19 ist die Steuerelektrode eines FeldeffektTransistors 23 angeschlossen, dessen Drain-Elektrode über einen Widerstand 24 ebenfalls von der positiven Spannungsleitung gespeist ist. Die Drain-Elektrode steht über einen weiteren Widerstand 25 mit einer Klemme 26 in Verbindung, an welcher, wie nachfolgend erläutert, der gesuchte Sollwert auftritt. Die Klemme 26 ist über einen Kondensator 27 mit der Null-Leitung verbundcn.A second circuit part, shown in FIG. 2, is provided to receive the square-wave pulses transmitted by the diode 18 of FIG. 1, to additionally deform and smooth them non-linearly, in order to generate a nominal value of low ripple. A receiver-side transistor 21 of the optocoupler 19 is connected via a resistor 22 on the one hand to a positive voltage line + and on the other hand connected to the associated zero line 0. The control electrode of a field-effect transistor 23 is connected to the transistor 21 of the optocoupler 19, the drain electrode of which is also fed by the positive voltage line via a resistor 24. The drain electrode is connected via a further resistor 25 to a terminal 26 at which, as explained below, the desired setpoint occurs. Terminal 26 is connected to the zero line via a capacitor 27.

Wenn die Diode 18 des Optokopplers 19 (Fig. 1) stromleitend ist, fliesst auch ein Strom durch den Transistor 21 des Optokopplers 19 (Fig. 2). Dann ist der Feldeffekt-Transistor 23 gesperrt, so dass sich der Kondensator über die Reihenschaltung der Widerstände 24 und 25 auflädt. Wenn anschliessend die Diode 18 und damit auch der Transistor 21 des Optokopplers 19 gesperrt sind, wird der Feldeffekt-Transistor 23 leitend, so dass sich der Kondensator 27 über den Feldeffekt-Transistor 23 und den Widerstand 25 entladen kann. Da der Widerstand 25 einen etwa viermal kleineren Wert als die Summe der Widerstände 24 und 25 hat, erfolgt die Entladung des Kondensators 27 schneller als dessen Aufladung. Dadurch steigt die Spannung an der Klemme 26 bei hoher Steuerspannung erheblich schneller als bei niedriger Steuerspannung, was die angestrebte quasi-logarithmische Charakteristik der Abhängigkeit der Spannung an der Klemme 26 von der Steuerspannung wesentlich verbessert. Da jedoch der Kondensator 27 eine verhältnismässig grosse Kapazität hat (beispielsweise etwa 3 uF), ist die Welligkeit der Spannung an der Klemme 26 sehr gering. Somit stellt die Spannung an der Klemme 26 den erstrebten Sollwert mit quasi-logarithmischer Abhängigkeit von der durch die Bürde 3 der Fig. 1 gelieferten, einstellbaren Steuerspannung dar.If the diode 18 of the optocoupler 19 (FIG. 1) is conductive, a current also flows through the transistor 21 of the optocoupler 19 (FIG. 2). Then the field effect transistor 23 is blocked, so that the capacitor charges via the series connection of the resistors 24 and 25. If the diode 18 and thus also the transistor 21 of the optocoupler 19 are subsequently blocked, the field effect transistor 23 becomes conductive, so that the capacitor 27 can discharge via the field effect transistor 23 and the resistor 25. There the resistor 25 has an approximately four times smaller value than the sum of the resistors 24 and 25, the capacitor 27 discharges faster than its charging. As a result, the voltage at the terminal 26 rises significantly faster at a high control voltage than at a low control voltage, which significantly improves the desired quasi-logarithmic characteristic of the dependence of the voltage at the terminal 26 on the control voltage. However, since the capacitor 27 has a relatively large capacitance (for example approximately 3 μF), the ripple in the voltage at the terminal 26 is very low. Thus, the voltage at terminal 26 represents the desired setpoint with a quasi-logarithmic dependence on the adjustable control voltage provided by burden 3 in FIG. 1.

Trotz ihrem einfachen Aufbau erzeugt die vorliegende Schaltungsanordnug reproduzierbare Spannungs-Sollwerte. Beispielsweise kann bei der beschriebenen Anwendung der Schaltungsanordnung in einem elektronischen Vorschaltgerät für eine Niederdruckgasentladungslampe mit einer einzigen Steuerspannung die Helligkeit einer grossen Zahl von Entladungslampen, deren Vorschaltgeräte je eine vorliegende Schaltungsanordnung enthalten, ohne merkbare Unterschiede der eingestellten Helligkeit der einzelnen Entladungslampen gesteuert werden.In spite of its simple structure, the present circuit arrangement generates reproducible voltage setpoints. For example, in the described application of the circuit arrangement in an electronic ballast for a low-pressure gas discharge lamp with a single control voltage, the brightness of a large number of discharge lamps, the ballasts of which each have an existing circuit arrangement, can be controlled without noticeable differences in the brightness set for the individual discharge lamps.

Claims (10)

1. Schaltungsanordnung zur Erzeugung eines elektrischen Spannungs-Sollwertes in nichtlinearer Abhängigkeit von einer elektrischen, einstellbaren Steuerspannung, gekennzeichnet durch einen Signalgenerator (9 bis 15) zur Bildung einer periodischen Signalspannung, welche während jeder Periode in gleicher Weise nichtlinear und monoton verläuft, und durch eine Vergleichsschaltung (4 bis 8), welche Eingänge für die Signalspannung des Signalgenerators und die einstellbare Steuerspannung sowie einen Ausgang aufweist, welcher dazu bestimmt ist, ein Vergleichssignal abzugeben, wenn die Signalspannung des Signalgenerators den Wert der einstellbaren Steuerspannung erreicht, wobei das Vergleichssignal bis zum Beginn der nächsten Periode der Signalspannung des Signalgenerators andauert.1. Circuit arrangement for generating an electrical voltage setpoint in a non-linear dependence on an electrical, adjustable control voltage, characterized by a signal generator (9 to 15) to form a periodic signal voltage, which is non-linear and monotonic in the same way during each period, and by a Comparison circuit (4 to 8) which has inputs for the signal voltage of the signal generator and the adjustable control voltage and an output which is intended to emit a comparison signal when the signal voltage of the signal generator reaches the value of the adjustable control voltage, the comparison signal up to the start the next period of the signal voltage of the signal generator continues. 2. Schaltungsanordnung nach Anspruch 1, dadurch gekennzeichnet, dass die Signalspannung des Signalgenerators (9 bis 15) einen exponentiellen Verlauf hat.2. Circuit arrangement according to claim 1, characterized in that the signal voltage of the signal generator (9 to 15) has an exponential curve. 3. Schaltungsanordnung nach Anspruch 2, dadurch gekennzeichnet, dass zur Bildung des exponentiellen Verlaufs der Signalspannung die Reihenschaltung eines Widerstandes (12) und eines Kondensators (13) zwischen die Speiseleitungen (+, 0) einer stabilisierten Gleichstromversorgung geschaltet ist.3. Circuit arrangement according to claim 2, characterized in that the series circuit of a resistor (12) and a capacitor (13) is connected between the feed lines (+, 0) of a stabilized DC power supply to form the exponential profile of the signal voltage. 4. Schaltungsanordnung nach Anspruch 3, dadurch gekennzeichnet, dass an den ersten Eingang eines Komparators (9) der Verbindungspunkt des Widerstands (12) und des Kondensators (13) angeschlossen ist, dass ferner an den zweiten Eingang des Komparators (9) der Verbindungspunkt zweier, zwischen den Speiseleitungen (+, 0) einen Spannungsteiler bildenden Widerstände (10, 11) angeschlossen ist, und dass der erste Eingang des Komparators (9) über einen Widerstand (16) mit dem Ausgang des Komparators (9) verbunden ist.4. Circuit arrangement according to claim 3, characterized in that the connection point of the resistor (12) and the capacitor (13) is connected to the first input of a comparator (9), that furthermore the connection point of two to the second input of the comparator (9) , between the feed lines (+, 0) forming a voltage divider resistors (10, 11), and that the first input of the comparator (9) is connected via a resistor (16) to the output of the comparator (9). 5. Schaltungsanordnung nach Anspruch 5, dadurch gekennzeichnet, dass der Ausgang des Komparators (9) zudem über einen Widerstand (14) mit dem zweiten Eingang des Komparators (9) verbunden ist.5. Circuit arrangement according to claim 5, characterized in that the output of the comparator (9) is also connected via a resistor (14) to the second input of the comparator (9). 6. Schaltungsanordnung nach Anspruch 4 oder 5, dadurch gekennzeichnet, dass der Verbindungspunkt des Widerstands (12) und des Kondensators (13) an den ersten Eingang eines weiteren Komparators (7) angeschlossen ist, und dass die Steuerspannung dem zweiten Eingang des weiteren Komparators (7) zugeführt ist.6. Circuit arrangement according to claim 4 or 5, characterized in that the connection point of the resistor (12) and the capacitor (13) is connected to the first input of a further comparator (7), and that the control voltage to the second input of the further comparator ( 7) is supplied. 7. Schaltungsanordnung nach Anspruch 6, dadurch gekennzeichnet, dass an den Ausgang des weiteren Komparators (7) ein zwischen die Speiseleitungen (+, 0) geschalteter Spannungsteiler (17, 18) angeschlossen ist.7. Circuit arrangement according to claim 6, characterized in that a voltage divider (17, 18) connected between the feed lines (+, 0) is connected to the output of the further comparator (7). 8. Schaltungsanordnung nach den Ansprüchen 4 und 6, dadurch gekennzeichnet, dass der Ausgang des weiteren Komparators (7) mit der Steuerelektrode eines Schalttransistors (23), an dessen Ausgangselektrode eine Lade- und Entladekreis (24, 25) für einen Glättungskondensator (27) angeschlossen ist, in Verbindung steht.8. Circuit arrangement according to claims 4 and 6, characterized in that the output of the further comparator (7) with the control electrode of a switching transistor (23), at the output electrode of a charging and discharging circuit (24, 25) for a smoothing capacitor (27) is connected. 9. Schaltungsanordnung nach Anspruch 8, dadurch gekennzeichnet, dass die Ausgangselektrode des Schalttransistors (23) über einen ersten Widerstand (24) mit der einen Speiseleitung (+) und über einen zweiten Widerstand (25) mit einer Anschlussklemme (26) für den Spannungs-Sollwert verbunden ist, wobei die Anschlussklemme (26) über den Glättungskondensator (27) mit der anderen Speiseleitung (0) verbunden ist, derart, dass bei gesperrtem Schalttransistor (23) der Glättungskondensator (27) über die Reihenschaltung der beiden Widerstände (24, 25) aufgeladen und bei leitendem Schalttransistor (23) über den zweiten Widerstand (25) und den Schalttransistor (23) entladen wird.9. Circuit arrangement according to claim 8, characterized in that the output electrode of the switching transistor (23) via a first resistor (24) with the one feed line (+) and via a second resistor (25) with a connecting terminal (26) for the voltage Setpoint value is connected, the connecting terminal (26) via the smoothing capacitor (27) to the other feed line (0) is connected in such a way that when the switching transistor (23) is blocked, the smoothing capacitor (27) is charged via the series connection of the two resistors (24, 25) and discharged via the second resistor (25) and the switching transistor (23) when the switching transistor (23) is conductive becomes. 10. Schaltungsanordnung nach Anspruch 8 oder 9, dadurch gekennzeichnet, dass der Ausgang des weiteren Komparators (7) mit der Steuerelektrode des Transistors (23) über einen Optokoppler (19) in Verbindung steht, wobei eine senderseitige Diode des Optokopplers (19) den einen Teil (18) des am Ausgang des Komparators (7) angeordneten Spannungsteilers (17, 18) bildet, und wobei ein empfängerseitiger Transistor (21) des Optokopplers (19) mit der Steuerelektrode des Schalttransistors (23) und einem Speisewiderstand (22) verbunden ist.10. Circuit arrangement according to claim 8 or 9, characterized in that the output of the further comparator (7) is connected to the control electrode of the transistor (23) via an optocoupler (19), with a transmitter-side diode of the optocoupler (19) the one Forms part (18) of the voltage divider (17, 18) arranged at the output of the comparator (7), and a receiver-side transistor (21) of the optocoupler (19) is connected to the control electrode of the switching transistor (23) and a supply resistor (22) .
EP93104805A 1992-04-02 1993-03-24 Circuit for producing an electrical voltage reference value depending on an electrical control voltage Expired - Lifetime EP0563765B1 (en)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
CH1073/92A CH683462A5 (en) 1992-04-02 1992-04-02 A circuit arrangement for generating an electric voltage command value in dependence on an electrical control voltage.
CH1073/92 1992-04-02

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EP0563765A2 true EP0563765A2 (en) 1993-10-06
EP0563765A3 EP0563765A3 (en) 1993-12-08
EP0563765B1 EP0563765B1 (en) 1995-08-23

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EP93104805A Expired - Lifetime EP0563765B1 (en) 1992-04-02 1993-03-24 Circuit for producing an electrical voltage reference value depending on an electrical control voltage

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EP (1) EP0563765B1 (en)
AT (1) ATE126907T1 (en)
CH (1) CH683462A5 (en)
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Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US3458801A (en) * 1967-06-01 1969-07-29 Itek Corp High voltage operational amplifier for use as an electronically controllable power supply regulator
EP0350518A1 (en) * 1988-07-12 1990-01-17 Eppendorf-Netheler-Hinz Gmbh Photometer
EP0432845A2 (en) * 1989-12-11 1991-06-19 Koninklijke Philips Electronics N.V. Method for controlling fluorescent lamp dimmers and circuit for providing such control

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US3458801A (en) * 1967-06-01 1969-07-29 Itek Corp High voltage operational amplifier for use as an electronically controllable power supply regulator
EP0350518A1 (en) * 1988-07-12 1990-01-17 Eppendorf-Netheler-Hinz Gmbh Photometer
EP0432845A2 (en) * 1989-12-11 1991-06-19 Koninklijke Philips Electronics N.V. Method for controlling fluorescent lamp dimmers and circuit for providing such control

Non-Patent Citations (2)

* Cited by examiner, † Cited by third party
Title
IBM TECHNICAL DISCLOSURE BULLETIN. Bd. 25, Nr. 11A, April 1983, NEW YORK US Seiten 5796 - 5797 D. BALLATORE 'CONSTANT VOLTAGE-RESISTIVE FEEDER FOR TELEPHONE LINE INTERFACE CIRCUITS.' *
RESEARCH DISCLOSURE 20. April 1991, HAVANT GB Seite 295 , XP000229641 'EXPONENTIAL DIMMING CHARACTERISTIC FOR LAMPS' *

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EP0563765A3 (en) 1993-12-08
CH683462A5 (en) 1994-03-15
EP0563765B1 (en) 1995-08-23
ATE126907T1 (en) 1995-09-15
DK0563765T3 (en) 1995-12-27
DE59300495D1 (en) 1995-09-28

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