Classifications

• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03K—PULSE TECHNIQUE
  • H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
  • H03K17/06—Modifications for ensuring a fully conducting state
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03K—PULSE TECHNIQUE
  • H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
  • H03K17/04—Modifications for accelerating switching
  • H03K17/042—Modifications for accelerating switching by feedback from the output circuit to the control circuit
  • H03K17/0424—Modifications for accelerating switching by feedback from the output circuit to the control circuit by the use of a transformer
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03K—PULSE TECHNIQUE
  • H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
  • H03K17/51—Electronic switching or gating, i.e. not by contact-making and –breaking characterised by the components used
  • H03K17/56—Electronic switching or gating, i.e. not by contact-making and –breaking characterised by the components used by the use, as active elements, of semiconductor devices
  • H03K17/60—Electronic switching or gating, i.e. not by contact-making and –breaking characterised by the components used by the use, as active elements, of semiconductor devices the devices being bipolar transistors
  • H03K17/601—Electronic switching or gating, i.e. not by contact-making and –breaking characterised by the components used by the use, as active elements, of semiconductor devices the devices being bipolar transistors using transformer coupling
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03K—PULSE TECHNIQUE
  • H03K2217/00—Indexing scheme related to electronic switching or gating, i.e. not by contact-making or -breaking covered by H03K17/00
  • H03K2217/0036—Means reducing energy consumption

Definitions

• This invention relates to electrical switches and switching systems.
• the invention particularly relates to switches and switching systems comprising bi-polar transistors.
• ⁇ sat. is the ratio of collector current to base current to achieve a given collector-emitter voltage when the transistor is saturated, for transistors of high current handling capacity ⁇ Sat. is generally in the order of 5 or 6. It is possible to generate base current by means of a s ⁇ parate low voltage power supply and an appropriate network arranged to maintain a constant ⁇ . between the instantaneous collector current and the base current.
• the present invention is applicable to Darlington connections having either two or more than two transistors in cascade but particularly concerns the penultimate and ultimate transistors in such a connection.
• a Darlington connected composite transistor the gain under saturated conditions of the composite transistor is the product of the gain of each individual transistor in the chain.
• a gai 'of 25 is readily available at a high current level
• the system may be made substantially more efficient and easier to cool. It is accordingly one purpose of the present invention to preserve the simplicity and high gain of a switching system employing the Darlington connection yet substantially to reduce loss arising from the magnitude of the base-emitter voltage of the output transistor when that transistor is saturated.
• an electrical switch comprises a first and a second bi-polar transistor in Darlington connection, and includes a transformer which responds to current flow through the collector of the second transistor to provide a regenerative feedback voltage to the collector of the first transistor.
• the aforementioned voltage overcomes the base-emitter voltage of the second transistor and thus substantially reduces or eliminates the aforementioned component of loss and permits the switch to operate more efficiently.
• an electrical switch comprises a composite Darlington transistor, of which the emitter of a first bi-polar transistor is connected to the base of a second bi-polar transistor and the collector of the first transistor is connected to the collector of the second transistor, and a transformer of which the primary winding is disposed in series with the collector/ emitter circuit of the second transistor and of which- the secondary is disposed to produce in the collector/ emitter circuit of the first transistor a voltage which
• OMPI reduces the base-emitter voltage of the second- transistor.
• the magnetic flux in the transformer will require resetting.
• One advantageous further development of the basic circuit disclosed herein is an electrical switch arrangement comprising two switches each with a transformer as previously described; the transformers share a common magnetic circuit and the arrangement is such that the flow of respective load currents through the second or output transistors produce magnetic fluxes in opposite senses in the said common magnetic circuit.
• switches are arranged to provide current flow in opposite directions through a common load.
• inverter in which switches as aforesaid are arranged to provide current flow in opposite directions through a common load.
• there may therefore be a winding, in series with the load, constituting a common primary for the transformers associated with the switches.
• the switches may also be useful in a circuit in which the switches ' are disposed in parallel to provide unidirectional
• the primary windings of the transformers for the twp switches may be disposed electrically in parallel and be wound in opposite senses on a common core.
• an electrical switch arrangement comprises at least one Darlington pair of transistors constituted by a driver transistor and an output transistor and a transformer of which the primary winding is disposed to carry load current of the output transistor and of which a secondary winding provides regenerative feed back to the collector of the driver transistor of the Darlington pair, there being provided in series with a load and the said output transistor a further winding which shares a common magnetic core or circuit with the aforesaid transformer and which is arranged
• Such an arrangement may include more than one Darlington pair each with its transformer sharing the same magnetic core; it is preferable that the Darlington switches be controlled so that only one is conductive at any particular time.
• the magnetic flux in the common core can be reset by the free-wheeling load current and such an arrangement provides the maximum on time for which the transformer overdrive action in the Darlington pair is maintained; the core is or can be driven from one saturation limit to the other and optimum use may be made of the magnetic characteristics of the material of the core.
• FIG 2 illustrates a basic form of the invention
• Figure 3 illustrates an inverter including two switches according to the invention
• Figure 4 illustrates another circuit according to the invention
• Figure 5 illustrates a further development of the basic circuit of the invention.
• Figure 1 of the accompanying drawings illustrates a basic form of a composite transistor comprising two bi-polar transistors in Darlington connection.
• the first or penultimate transistor 1 usually termed driver transistor, is connected so that its emitter current corresponds to the base current of the second, or output, transistor 2.
• the collectors of the two transistors are connected together to a common point 3.
• the connection shown in Figure 1 is not the only form of Darlington connection for which the present invention is suitable.
• the second or output -transistor 2 may be constituted by a plurality of transistors arranged in parallel , fed from a single input transistor corresponding•to the transistor 1 in Figure 1.
• the transistor 2 would normally be constituted by the last transistor in the Darlington connection.
• One advantage of the Darlington connection as shown in Figure 1 is the multiplication of the gains of the individual stages.
• an important disadvantage is the increase in the collector/emitter voltage of the transistor 2.
• the two transistors are saturated, it is clear that the potential difference between the point 3 and the emitter of the transistor 2 cannot be less than the sum of the collector/emitter voltage of the transistor 1 and the base-emitter voltage of the transistor 2.
• the component represented by the base-emitter voltage of the transistor 2 may represent a significant fraction of the supply voltage and the power loss due to this component may represent a very significant part, perhaps 60% to 70%, of the total power loss within the transistor 2.
• Figure 2 illustrates a Darlington connection as shown in Figure 1 but including a transformer 4 which provides regenerative feedback.
• the primary winding 5 of this transformer is in series with the collector/emitter circuit of the output transistor 2 whereas the secondary winding 6 of the transformer 4 is in series with the collector circuit of the input transistor 1.
• the winding 6 is connected between the collectors of the transistors 1 and 2.
• the transformer has a core 7 with a square B-H characteristic and preferably low hysteresis loss.
• the collector current of transistor 2 flows through the primary winding 5 of the transformer 4 and induces a secondary current, of which the magnitude is governed by the ratio of the turns of windings 5 and 6, in a direction which aids the conduction of the transistor 2.
• This current is driven by a voltage appearing across the secondary 6 in a sense which, for the conductivities shown, raises the collector- of the transistor 1 and thereby raises the base potential of the transistor 2 even though its collector potential is depressed relative to the emitter thereof.
• the required drive voltage to overcome the base- emitter voltage of the second transistor is generated by the transformer.
• the transformer 4 constitutes a current transformer which determines the saturated gain ⁇ , of the transistor 2; the gain is given by the ratio N between the number of turns of the primary winding to the number of turns of the secondary winding.
• N the ratio N between the number of turns of the primary winding to the number of turns of the secondary winding.
• the means for resetting might comprise a third winding which is energised alternately with the conduction periods of the transistors 1 and 2, but in many circumstances the
• FIG. 3 illustrates the invention employed in a half bridge inverter.
• Other forms of inverter, whether single or polyphase, can be constituted in a similar manner.
• the second switch of which the components are denoted by the postscript "a" is arranged with its output transistor 2a in series with the output transistor 2 of the first switch, the switches being connected between a positive rail 9 and a negative rail 10.
• the primary winding 5 is common to the two switches and is connected to a point between the emitter of the transistor 2a and the junction point 3 of the first switch and is also connected in series with a load 11 of which theother side is connected to earth 12.
• the two switches are connected to a point between the emitter of the transistor 2a and the junction point 3 of the first switch and is also connected in series with a load 11 of which theother side is connected to earth 12.
• Darlington switches conduct currents which are in opposite directions through the load 11. Input drive signals may be applied to the bases 8 and 8a of the respective input transistors 1 and la.. Each Darlington switch operates in the manner already described with
• FIG. 4 illustrates another circuit in which two Darlington switches are employed. In this circuit the switches constituted by the transistors 1,2 and la,2a respectively, are in parallel and the primary windings 5 and 5a. of the respective transformers are ' constituted by single turns on a common core 7. Each primary winding 5,5a is in series with the common load 11. A free-wheeling diode 13 is connected around the load 11.
• the circuit shown in Figure 4 could form one arm of a multi-phase inverter but could be used on its own in order to provide uni-directional current flow through the load 11.
• the Darlington switches are turned on alternately, the periods of conduction of the two switches being preferably, though not essentially, equal.
• Each Darlington switch operates , in the manner described with reference to Figure 2 and the conduction of each Darlington switch automatically resets the flux of the transformer for the other Darlington switch.
• Figure 5 illustrates another circuit which combines the transformer overdrive for a Darlington pair of transistors with the use of load current to reset the overdrive flux. It will be understood that when bi-polar transistors are used to control high currents the lowest voltage drop in the on state of the transistor is achieved when the transistor is driven into its saturated state by the injection of a large base current sufficient to depress the collector/emitter voltage to its lowest level. If a collector current of, for example, 500 amps is to be controlled then a base current of about 100 .amps is required. This is difficult and expensive to provide and would cause a high power loss external to the transistor, making the overall system inefficient.
• the base drive required to control 500 amps may be of the order of 5 to 8 amps but the voltage of the Darlington switch in the conductive state rises to the sum of the
• the power loss in the on state is two to three times that of a single transistor.
• the power loss in a Darlington switch can be reduced to the order of that in a single transistor, while the current gain is preserved, using the overdrive described previously but in some circumstances it may become difficult or expensive to reset the flux in the transformer core if the transistor switch is required to operate at low frequencies, for example at less than 1 kilohertz, and with a high mark to space ratio. In particular, such difficulty arises unless the switch is operated at a high frequency with a maximum of 50% mark to space ratio or the transformer core is very large.
• circuit shown in Figure 5 by way of example is suitable particularly for comparatively low frequency operation though is not limited thereto.
• the transistors 1 and 2 are arranged in Darlington connection.
• the primary winding 5 of a transformer is in series with the collector of the transistor 2 and the secondary of this transformer is connected between junction 3, to which the primary 5 is connected, and the collector
• a second Darlington switch is constituted by the driver 'transistor 1a and the output transistor 2a which are associated with the transformer having a primary ' 5a and a secondary 6a similar to the primary 5 and secondary 6. It is desirable that the driver transistors 1 and ' 1a be operated such that the two Darlington switches are never conductive at the same time but apart from this constraint the Darlington switches may be turned on and off independently by appropriate signals on the respective input lines 8 and 8a..
• Figure 5 illustrates the two Darlington switches connected in a half bridge configuration driving a single load 11.
• the same configuration may be adopted in various inverter and converter circuits, both single-phase and multi-phase.
• the same scheme may be applied to a single Darlington switch which may be used,-for example, as a DC chopper.
• the operation of the circuit will be described with reference to only one Darlington switch, as constituted by the transistors 1 and 2.
• All the transformer windings shown in Figure 5 are on a common core. These windings include a further winding 20 which is connected between the common junction 3 and the load 11. The polarity of this winding is such that it produces flux in the common magnetic core opposite to that produced by normal load current through the primary windings 5 and 5a.
• the transistors ' 1a and ' 2a are normally non-conductive and the transistors 1 and 2 are being turned on and off under the control of signals applied to the line 8 so as to draw current out of the load over a comparatively long period, for example over one half cycle.
• the load current can free-wheel through free-wheeling diode 21 connected from the winding 20 to the positive rail 9.
• a further free-wheeling diode 22 is connected between the winding 20 and the negative rail 10 in this particular embodiment.
• the off period is short compared to the time constant of the load the load current will remain almost constant.
• the uni-directional current flow through the winding 20 will saturate the core of the transformer in one direction when none of the other windings 5, 6, 5a and 6a are carrying current.
• the load current is diverted from the diode 21 to flow through the windings 5 and 6 in appropriate proportions.
• the winding 20
• OMPI has one turn, the winding 5 has two turns and- the winding 6 has six turns.
• the flow of load current through windings 5 and 6 is in a ratio of approximately 6:1 and in a direction that produces magnetic flux 5 opposing that produced by the winding 20.
• the magnetic flux produced by the winding 5 will exceed the magnetic flux produced by the winding 20 by a factor of approximately 2 (if the effect of the winding 6 is ignored) so that the transformer core
• This arrangement provides the maximum on time for which the transformer overdrive action, described earlier herein, is maintained and since the core is being driven from one saturation limit to the other 0 optimum use may be made of the material of the core.
• the transistor 1 When the transistor 1 is turned off, its collector swings positive until either a diode 25, which is optional, fitted between the collector of transistor 1 and the positive rail 9, conducts so as to clip the 5 . voltage across winding 6 or, if the diode 25 is not provided, the collector of the transistor 1 swings positive until the transformer action between the winding 5 and the winding 6a causes the conduction of a diode 23,. which is connected between the
• winding 6 will equal the supply voltage and the voltage at the collector of transistor 2 is, by transformer action, above the voltage of the positive supply rail by the product of the supply voltage and the turns ratio of the windings 5 and 6.

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• Engineering & Computer Science (AREA)
• Power Engineering (AREA)
• Inverter Devices (AREA)
• Electronic Switches (AREA)

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• 1983
  • 1983-12-20 WO PCT/GB1983/000339 patent/WO1984002624A1/en unknown
  • 1983-12-20 EP EP19840900285 patent/EP0128940A1/de not_active Withdrawn

Non-Patent Citations (1)

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