Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/01—Equalisers
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04B—TRANSMISSION
  • H04B15/00—Suppression or limitation of noise or interference

Definitions

• This invention relates to apparatus for processing a signal received from a transmission channel, which signal represents a plurality of data signal values, the apparatus comprising a first decision circuit for providing signals representing tentative decisions as to the data signal values in response to the received signals, a canceller for forming cancellation signals in response to the tentative decisions.
• Data signals are subject to alteration by various types of interference.
• phase and/or amplitude distortion in a transmission channel can result in intersymbol interference.
• the latter is conventionally dealt with in state-of-the-art receivers via adaptive feedforward equalization which uses weighted sums of, for example, passband samples of the received data signal to generate a stream of equalizer outputs which are substantially interference- free.
• the passband equalizer outputs are demodulated to form a stream of baseband outputs in response to which decisions are formed as to what was the transmitted data.
• An error signal formed in response to the difference between each baseband output and a corresponding decision is used to adaptively update the equalizer weighting coefficients in such a way as to minimize, for example, the average of the square of the error signal.
• Adaptive equalizers--or structures that operate like adaptive equalizers--are also used to deal with interference which is not channel-distortion-induced.
• the interference is in the form of so-called directional noise, or jamming signals.
• directional noise or jamming signals.
• These signals which impinge on the antenna from directions other than the direction of the desired signal, are effectively removed from the received, interference-corrupted signal by the adaptive equalization, or (more properly in this case) adaptive filtering, process. See, for example, B. Widrow et al, "Adaptive Antenna Systems," Proc. of IEEE, Vol. 55, No. 12, December 1967, pp. 2143-59.
• the weighting coefficients used in the canceller are adaptively updated in response to an error signal derived from the difference between each final decision and the corresponding cancelled sample.
• the noise-enhancement problem is solved in accordance with the present invention by apparatus for processing signals in which the apparatus further includes a phase compensator circuitry for generating first samples of the received signal from which interference due to phase distortion has been substantially removed, each of the cancellation signals being substantially equal to the interference in a respective one of the first samples, and circuitry for forming signals representing final decisions as to the data signal values in response to the first samples and the cancellation signals.
• the cancellation signals are combined not with samples of the received signal per se but, rather, with phase-compensated samples thereof.
• the frequency spectrum of the phase ⁇ rcompensated samples is a function of
• the phase-compensated samples represent samples of the received signal from which the component of the interference due to phase distortion in the channel has been substantially removed.
• the present invention has been found to provide error rates over the technique in the aforementioned patent when, for example, the data signals have been transmitted over a noisy channel having both high amplitude and phase distortion.
• the phase-compensated samples of the received data signal may be generated by transversal filtering the received signal samples with a "matched filter” having a transfer function substantially equal to
• the frequency spectrum of the phase- compensated samples is substantially equal to
• the weighting coefficients used to achieve the matched filter characteristic are illustratively adapted, or updated, in response to the same above-mentioned error signal employed to adapt the weighting coefficients used in the canceller.
• phase compensated samples can be generated via an all-phase compensator having a transfer function substantially equal to e -j ⁇ Y ( ⁇ ) .
• the frequency spectrum of the phase ⁇ compensated samples is substantially equal to
• the present invention provides about a 2 dB improvement in output signals to-noise ratio (SNR o ) over a conventional feedforward equalizer at a baud rate of 2400 and about 1 dB improvement in SNR o over combined feedforward and decision feedback equalization.
• SNR o output signals to-noise ratio
• the invention thus provides a one- to two-order-of- magnitude improvement in bit error rate over such prior art.
• FIG. 1 depicts a receiver 10 for double sideband-quadrature carrier (DSB-QC) data signals transmitted from a far-end transmitter (not shown) over a voiceband telephone channel.
• a complex data signal value each of whose two, i.e., real and imaginary, components can take on one of a predetermined set of values, is generated during each of a succession of baud intervals of duration T.
• the two signal values representing the real and imaginary components of each complex data signal value, amplitude modulate respective 1800 Hz in-phase and quadraturer-phase carrier waves, which, in combination, comprise the DSB -QC signal transmitted to receiver 10.
• the complex data signal values typically represent a stream of binary data to be transmitted.
• a two-dimensional scheme is employed in which each complex data signal value, on its own, represents p data bits.
• each complex data signal value represents an independent data "symbol.”
• an n-dimensional scheme, n>2 could be employed.
• each of the complex data signal values transmitted in n/2 successive baud intervals could be used to represent np/2 data bits.
• each data "symbol" is comprised of n/2 successive complex data signal values.
• the received DSB-QC signal s r (t) is applied to input circuit 15.
• the latter illustratively includes? a series combination of an analog bandpass filter, which filters out energy in signal s r (t) outside the transmission band of interest; an automatic gain control circuit; an analog phase splitter; and an a/d converter.
• the latter samples the two outputs of the phase splitter to provide a sampled line signal comprised of complex passband line samples Y k .
• Samples Y k are illustratively formed at twice the baud rate, i.e., at a rate of 2/T complex samples/ sec, the index k thus advancing at that rate.
• phase compensator 40 provides a phase-compensated passband signal comprised of complex passband samples Q m .
• the frequency spectrum of samples Q m is Q( ⁇ ).
• the latter is a function of
• phase compensator 40 is illustratively a "matched filter" having a transfer function substantially equal to
• phase compensator 40 can be a so-called all- phase compensator having a transfer function substantially equal to so that Q( ⁇ ) is substantially equal to
• samples Q m represent samples of the received (bandpass filtered) signal with the intersymbol interference due to phase distortion in the transmission channel substantially removed.
• Samples Q m are provided at the baud rate 1/T, the index "m” thus advancing at that rate.
• Samples Q m are applied to demodulator 50 which generates a phase-compensated, or phase-distortion- removed, baseband signal comprised of complex phases- compensated, or phase-distortion-removed, baseband samples U m .
• the latter representing baseband weighted combinations of samples Y k , are, in turn, applied to the "+" input of a combiner 55.
• Samples U m are generated by the demodulator in accordance with
• Line samples Y k are also applied to tentative decision former 20.
• the latter provides a stream of tentative complex decisions ⁇ m-M as to what the transmitted complex data signal values were. (In the subscript "m-M”, "m” is, again, the baud index and "M” is a constant equal to half the baud interval span of canceller 30, described below. )
• Tentative decision former 20 may comprise any circuitry capable of forming decisions as to the values of the transmitted complex data signals. It may comprise, for example, circuitry which is conventionally used to form "final" decisions such as the adaptive transversal equalizer/demodulator/decision former circuitry shown in U. S. Patent 4,245,345, U. S. Patent 4,247,940 and U. S. Patent 4, 237, 554.
• Tentative decision former 20 could also include decision feedback equalization circuitry such as shown by D. A. George et al in "An Adaptive Decision Feedback Equalizer," IEEE Trans, on Comm. Tech., Vol. COM 19, No. 3, June 1971, pp. 281-293, and by D. D.
• Tentative decision former 20 could also include the nonlinear feedforward and feedback equalization circuitry shown in U. S. Patents 4,213,095 and 4,181,888.
• Tentative decisions ⁇ m-M are applied to canceller 30, described in detail below. For the present it suffices to say that canceller 30 operates in response to the tentative decisions to generate a cancellation signal comprised of complex signals V m each of which is an estimate of the intersymbol interference component of a corresponding one of samples U m .
• phase compensator 40 may include a bulk delay unit for delaying samples Y k by an appropriate amount.
• Samples P m are applied to final decision circuit 60 which quantizes them to form a stream of complex final decisions ⁇ m as to what the transmitted data signal values were. ("Final" decisions ⁇ m could, if desired, serve as tentative decisions for yet further signal processing circuitry.)
• phase compensator 40 is illustratively a matched filter having a transfer function substantially equal to
• phase compensator 40 could be arranged to operate on baud-spaced line samples rather than fractionally-spaced line samples. The latter is advantageous, however, because, unlike the former, it does not require precise control of the sampler timing phase.
• L may be chosen to be greater than the baud interval span of the channel by an amount which ensures that each U m is generated at substantially the same time as the corresponding V m .
• Canceller 30 is also illustratively a transversal filter, which generates signals V m by forming a complex linear combination of the tentative decisions.
• the corresponding tentative decision, ⁇ m is not used in generating the intersymbol interference estimate V m .
• canceller 30 illustratively generates each of the signals V m by forming a complex linear combination of the (2M-1) most recent tentative decisions (excluding ⁇ m ) in accordance with the relation
• ⁇ m + (P m - ⁇ m ).
• the error signals ⁇ m are formed by combiner 65 which subtracts each of the final decisions ⁇ m from the corresponding cancelled sample P m . Updating of the C k (m)'s used in canceller 30 is illustratively carried out within the canceller using a stochastic gradient criterion realized by updating relation
• ⁇ is a selected step si ze whose sign is, determined by the selected pol ar ity (+ ) of error signals ⁇ m ( in the present embod iment , ⁇ i s a posi tive number ) , and ⁇ 1 i s a so-called " leakage" term .
• the func tion of the leakag e term is to prevent the C k (m) ' s from drifting to overly-larg e val ues and it may take any of several fo rms.
• ⁇ is a constant and SGN is the complex signum function whose value is +1+j, depending on the signs of the real and imaginary components of its arguments.
• This form of leakage makes it particularly easy to carry out the updating relation in embodiments in which the C k (m)'s are stored in two's complement form.
• the term is combined with C k (m) to form a partially updated coefficient.
• the leakage portion of the updating relation is then realized by combining the sign bit for each component of the partially updated coefficient with a particular bit position in that component.
• the bit position is determined by the desired value of ⁇ which, in turn, can be determined experimentally. If the coefficient drifting phenomenon does not manifest itself in a particular embodiment, ⁇ 1 can be zero.
• ⁇ is a second selected step size which could be equal to ⁇ and ⁇ 2 is another tap leakage term.
• ⁇ 2 is another tap leakage term.
• the form of ⁇ 2 , as well as the manner in which this updating relation is carried out can be analogous to what is described above in connection with updating of the canceller coefficients.
• the step sizes ⁇ and ⁇ used in the above updating relations must satisfy the constraint that their magnitudes are less than the quantity
• ⁇ x 2 > is the average input signal power.
• Receiver 10 can be started up in any desired way.
• receiver 10 could be modified by, for example, providing an M-baud delay unit whose input is taken from the output of tentative decision former 20 and whose output is applied to the "-" (subtractive) input of combiner 65 in place of the output of final decision circuit 60.
• Each filter might comprise, for example, a programmed general-purpose processor (microprocessor), or specials-purpose LSI circuitry operating under the control of micro-coded instructions.
• microprocessor general-purpose processor
• Such equalizer sections could be used not only to implement canceller 30 and/or phase compensator 40, but the equalizer within tentative decision former 20. Indeed, the functions of any portion or all of the entire receiver can be realized by, for example, one or more programmed general purpose processors, microcoded circuits, discrete components or combinations of these.
• each component could in fact be comprised of a) circuitry shared in whole or in part with the other components b) combined with stored program instructions which cause that circuitry to be operated so as to provide the function of the component in question.
• the weighting coefficients used in the phase compensator and canceller can be updated using updating relations other than those disclosed herein. It is tacticly assumed hereinabove that V m , Q m , U m , P m , ⁇ m , ⁇ m and E m are generated within the same baud interval of duration T. In practical implementations, however, it may not be possible or desirable to do this.
• the updating relations for the canceller and phase compensator weighting coefficients would have to be modified to take account of the fact that one or both of error signals ⁇ m and E m may not be available in time to update the C k (m)'s and W k (m) 's.
• the following updating relations may be used:
• d 1 (d 2 ) is at least equal to the number of baud intervals of delay between the generation of V m (Q m ) and error signals ( ⁇ m ) E m .
• d 1 and d 2 can take on any integer values including zero, the latter case corresponding, of course, to the updating relations first presented hereinabove.
• d 1 being an even number
• d 2 being an even number
• phase compensator 40 could be a so-called all-phase compensator having a transfer, function substantially equal to e -j ⁇ Y( ⁇ ).
• an all phase compensator could be realized by, for example, a fast Fourier transform circuit which would generate a) a complex frequency-domain line spectrum of the samples Y k and b) generate as the phase compensator output baud-spaced time-domain samples derived from the amplitude component of that line spectrum.
• phase compensator 40 could feed phase compensator 40 with equalized samples from within tentative decision former 20 rather than with line samples. This, however, might make the system less stable.
• final decisions ⁇ m- (M+1) ... ⁇ m could be fed back for use in canceller 30 in place of tentative decisions ⁇ m- (M+1 ) ... ⁇ m . Since the final decisions should contain fewer errors than the tentative decisions, this may result in a somewhat more accurate V m and, thus, a somewhat lower final decision error rate.
• demodulator 50 could precede phase compensator 40, in which case, of course, the phase compensator would form weighted combinations of the demodulator's output samples rather than weighted combinations of samples Y k .
• the updating relations for the W k (m) 's would include baseband error signals ⁇ m rather than passband error signals E m .
• the frequency spectrum of the phase compensator input samples could, as a matter of notation, still be denominated Y( ⁇ ) as long as Y( ⁇ ) is understood to be a baseband spectrum.
• the transfer function of phase compensator 40 could still be represented as
• the invention is applicable not only to passband systems, but also to baseband systems and not only to systems using n-dimensional (n>2) modulation, but to systems using one-dimensional modulation, e.g., amplitude modulation.
• the invention is applicable not only to transmission over voiceband telephone channels, but other types of channels, such as microwave radio channels and is effective in dealing not only with intersymbol interference but other forms of interference such as discussed hereinabove in connection with adaptive antenna systems.
• the invention is thus applicable to such systems as the recording readback arrangement shown by O. R. Luhrs et al in IBM Tech. Disci. Bull., Vol. 20, No. 2, July 1977, pp. 523 et seq.

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• Engineering & Computer Science (AREA)
• Computer Networks & Wireless Communication (AREA)
• Signal Processing (AREA)
• Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)

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• 1981
  • 1981-11-18 US US06/322,517 patent/US4412341A/en not_active Expired - Fee Related
• 1982
  • 1982-10-13 CA CA000413311A patent/CA1174745A/en not_active Expired
  • 1982-10-20 WO PCT/US1982/001414 patent/WO1983001879A1/en active IP Right Grant
  • 1982-10-20 DE DE8282903403T patent/DE3271186D1/de not_active Expired
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