DE69029267T2 - RESONANCE CAVES FOR MAGNETIC NUCLEAR RESONANCE - Google Patents

Description

RESONANCE CAVES FOR NMR

The present invention relates to resonant cavities for NMR and particularly to resonant arrays for receiver and transmitter probes for use at high frequencies. Resonator arrays [Hayes, C., Edelstein, W., Schenk, I., Muller, O. and Eash. M., J. Mag Res. 63, 622-628, (1985)] are becoming increasingly popular for the receiver and transmitter coil probes in nuclear magnetic resonance imaging and spectroscopy. This is all the more so as ever higher magnetic fields and therefore frequencies are applied. The difficulty in tuning standard multi-turn saddle coils and short solenoid coils makes alternative delay lines and resonator arrays more attractive. We have considered several structures that employ the resonator array principle, for example the petal resonator [Mansfield, P., J. Phys. D., 21, 1643-4 (1988)]. In considering these devices, it is necessary to develop the general theory of lumped parameter circuits as applied to these systems. We have considered the theory using a general matrix approach (Fisher, E. M., 1955 Electronic Engineering 27, 198-204, Sander K. F. and Reed G. A. L. 1978 Transmission and Propagation of Electromagnetic Waves, C. U. P. Cambridge). Recently, a perturbation approach for the analysis of non-symmetrical "birdcage" resonators has been published [Tropp, J., J. Mag. Res. 82, 52-62 (1989)].

US-A-4,602,234 discloses a coil assembly for NMR comprising a non-resonant structure including support rings and a plurality of rods, the structure being made into a self-resonant standing wave structure by adding a split guard ring coupled to the inner ring via discrete capacitors.

US-A-4,686,473 discloses a resonant structure supporting a standing wave using two end rings, to support pole conductors, with the necessary resonant characteristics provided by discrete components in the form of added coils and capacitors.

The present invention provides a resonance arrangement for NMR for use at high frequencies as in claim 1.

Now, embodiments of the present invention will be explained by way of example with reference to the accompanying drawings.

Figure 1 shows sections of a transmission line with lumped parameters as (a) π section and (b) T section;

Figure 2 is a graph showing the allowable frequency characteristics for a simple low-pass filter section;

Figure 3 shows a tunable NMR cavity comprising an array of slotted loop resonators, with the effective cavity length l according to the present invention;

Figure 4a shows an end plate of the tunable NMR cavity of Figure 2;

Figure 4b shows an alternative design of an endplate with larger axial access;

Figure 4c shows a driver circuit with phase quadrature arrangement and symmetrical arrangement;

Figure 4d shows a flow guide sleeve or end ring guide tube for use with the alternative end plate design of Figure 4b;

Figure 5 shows a plot of inductance versus diameter for a single slotted loop resonator;

Figure 6a shows an equivalent circuit of the resonant cavity structure, with points A and B connected;

Figure 6b shows a plot of the frequency characteristics for a section of the cavity resonator of Figure 6a;

Figure 7 shows a plot of a linear regression of f² versus 1/l, yielding asymptotic values of ω0;

Figure 8 shows a plot of frequency f versus length for the phase quadrature and symmetric drive modes of the cavity resonator;

Figure 9 shows a section of an end plate for a lower frequency resonator design that allows larger central access;

Figure 10a shows in plan view a schematic arrangement of slotted loop resonators with 2 layers to increase the loop inductance;

Figure 10b shows a section in side view;

Figure 11 shows a schematic arrangement for a cavity resonator with 2 turns;

Figure 12 shows a schematic view of a portion of an end plate arrangement for a 2-turn cavity resonator;

Figure 13 shows an alternative arrangement in which the loops are rotated 90º out of the plane of the plate;

Figure 14 shows a schematic diagram of an end plate of a high-pass resonant cavity coil;

Figure 15a shows a modification of the cavity resonator of Figure 14;

Figure 15b shows the equivalent circuit for the resonator of Figure 15a;

Figure 16 shows a split or half-resonator arrangement with reflective conductive shield according to the present invention;

Figure 17a shows a diagram of an end view of a wire arrangement for a saddle coil, where additional wires W, W' have been added for symmetry reasons but without carrying current, i.e. at standing wave nodes;

Figure 17b shows an equivalent split or half-saddle arrangement with reflective shielding;

Figure 18a shows a sketch of a split or half-saddle arrangement with shielding;

Figure 18b shows a strip coil with shielding;

Figure 18c shows an equivalent circuit of a split saddle or strip coil with driver.

THEORY

In the following we will represent a section of our resonant structure by a lumped parameter circuit in the form of a π section or a T section as shown in Figure 1. We assume that there is no interaction between the sections (see below).

The transmission matrix A for the section, considered as part of a longer transmission line, satisfies the propagation equation

Ψn = AΨn+1 (2

where En, En+1 etc. from equation 1 are the output and input voltages and currents respectively and Ψn etc. stand for the column matrices.

For N identical sections, equation 2 implies that

Ψ₀0; = ANΨN. (3)

If the wave propagation along the transmission line is maintained, we can also write

Ψn = µΨn+1 (4)

where the matrix µ represents the joint losses and phase changes per section. The combination of equations 2 and 4 gives the characteristic equation

AΨ = µΨ (5)

The eigenvalues of equation 5 represent the sustainable frequency and phase properties of the circuit. They can be obtained by diagonalizing A. The diagonal values are obtained from the determinant

det (A - µ) = 0 (6)

Since we are dealing with a passive network, det A = 1. The eigenvalues of equation 5 are generally complex and can be written because µ₁µ₂ = 1 as

µ₁₋₀ = 1/2A&sub0; + [(A₀/2)² - 1]1/2 = ey (7a)

µ₂ = 1/2A&sub0; - [(A₀/2)² - 1]1/2 = e-y (7b)

where

A₀ = Tr A (8)

where Tr is the trace or diagonal sum. Later we will see that y is the propagation constant of the transmission line. This is generally complex and is given by

y = tanh⊃min;¹ [(A�0;/2)² - 1]1/2/(A�0;/2) = α + iβ (9)

If α is zero, which corresponds to a lossless line, we see from equations (7-9) that due to the invariance of the trace with respect to the basis functions Ψn

Tr A = 2 cos β (10)

Let S be the matrix of the collinear transformation that diagonalizes A. Then we can write

S⊃min;¹AS = µ (11)

which can be inverted to

A = SµS⊃min;¹ (12)

Circuit details

Before going further, it is necessary to consider the transfer matrix for the particular circuit section. For the π section in Figure 1a,

while for the T section in Figure 1b

Assuming A₁₁ = A₂₂, i.e. for a symmetrical section, it is easy to show that the characteristic impedance of the section Z�0 is given by

Z²&sub0; = A₁₂/A₂₁ . (15)

Using the invariance of Tr A together with equations 7a, 7b and 8, it follows that the transfer matrices, equations 13 and 14, can be generally written as

which can also be generated from matrix 5 as follows

where the matrices S, S⊃min;¹ are given by

Using S we see that

and by inverting

For a lossless transmission line, with N sections corresponding to a wavelength P (P integer), Equation 20 shows that the phase relationship along the line satisfies the following relationship

Nβ = 2πMP (21)

where β is the phase shift per section and M (integer) is the resonant mode.

Using this formulation, a one-turn cyclic transmission line was designed, which forms the basis of the petal resonator [Mansfield, P., J. Phys. D., 21, 1643-4, (1988)]. Other related structures are what we have called the "chain mail" coil and the chain or "necklace" resonator, useful for studying limited parts of the anatomy, for example the neck.

In these designs, the current distribution in the successive elements of the transmission line around a cylindrical surface follows a cosine or sine variation as a function of the cylindrical azimuth angle θ. For straight wires lying on the surface of a cylinder parallel to the cylinder axis, a cosine current distribution produces a uniform magnetic field transverse to the cylinder axis. This will be the case if the straight wires form the impedance elements Z1 of Figure la and the voltage on the line follows a cosine variation around the driving point. In this case, the current Iwn flowing through the n-th wire is given as

Iwn = En/Z₁ . (22)

This current follows En, while In, equations 1, 3 and 16, varies as sin Nβ for a lossless line.

In NMR imaging applications, a coil structure that generates a uniform transverse magnetic field is useful as both a transmitter coil and a receiver coil. We also note that multi-turn resonator structures with P > 1 are possible.

Input impedance

When designing cavity resonators, it is useful to have an expression for the input impedance of the device. This can be obtained by first considering the expression for an N-element transmission line in Equation 20 together with Equations 1 and 3. Let the input and output voltage and current be V₁, I₁ and V₂, I₂, respectively. Then we get

V1; = V2 cosh Ny + I₂Z�0; sinh ny (23a)

and

I1; = (V₂/Z�0) sinh Ny + I₂ cosh Ny. (23b)

Therefore, the input impedance is Z₁ = V₁/I₁. Let the line be terminated by Z₂ = V₂/I₂. Substituting into the above, we obtain an expression for Z₁ of a discrete line that is similar to the known result for a continuously distributed line, i.e.

Z1=Z0 [Z2; cosh Ny + Z&sub0; sinh Ny]/Z&sub2; sinh Ny + Z&sub0; cosh Ny]. (24)

For an open line with wavelength P, Z₂ = ∞. In this case,

Z1; = Z0/tanh Ny (25a)

and for a small argument Ny

Z�1 Z�0/Ny. (25b)

It is worth pointing out that the impedance of an open Pλ line is not changed if its output is connected to its input. This means that all resonator designs herein can be either physically cyclic, i.e. actually connected at the beginning and end, or cut off at a point of high impedance. The cyclic boundary conditions are identical in both cases. We also note that the expression Z1 is developed for a π or T section as shown in Figure 1. The actual circuits designed are symmetrical about the ground point as discussed below with reference to the high frequency probe.

Q of the cavity resonator

The quality factor of the cavity is obtained from equation 25 by replacing y = α + iβ and noting, that for a Pλ line due to Nβ = 2πMP in resonance a small shift δω results in the following

tanh N Nα + 2π δω/ω (26)

Using this approximation in equation 25a, we note that it produces a Lorentz variation of Z1 versus δω with a given linewidth at half height if

2Nα = 4π δω/ω) = 2π/Q. (27)

The input resistance is in resonance

R = Z�0/Nα . (28)

By combining equations 27 and 28 we obtain the Q factor

Q = πR/Z�0 . (29)

Simple resonator designs

Low pass

If Z�1 = 2/jωC and Z�2 =( jωL + r), the transfer matrix for this π section is given by

From equations 10 and 30 we see that for r = 0

Tr A = 2 - (ω/ω0)2 = 2 cos β (31)

where

ω²�0 = 1/LC. (32)

Applying this transcendental equation yields the allowed frequency response hatched in Figure 2 for this low-pass filter section.

Combining equations 21 and 31, we see that the condition for a sustained standing wave in an open line or in a resonant cyclic structure involving a fixed length line looped back on itself so that the input and output terminals are connected is given by

(ω/ω�0)² = 4 sin²(πMP/N) (33)

for 1 ≤ MP ≤ N/2. Solutions of equation 33 show that the mode frequencies increase from the lowest mode M = 1 to the limiting mode M = N/2P.

As expected, we also find that

α; = r(C/L)1/2 = r/2Z&sub0; and β = ω/ω0 . (34)

In this case we get the ratio

R = 2Z²₀/No. (35)

From equations 27 and 35 we also obtain the value for Q

Q = 2πZ�0/Nr (36)

i.e. Q is determined by the resistance characteristics of a single section.

High-pass line

In this arrangement, Z�1 = 2(j L + r) and Z�2 = 1/jωC. With these parameters we find for r = 0

(ω₀/ω²) = 4 sin²(πMP/N)

As in the case of the low-pass section, at high frequencies Z0 = (L/c)1/2. If losses are included, the following also applies:

α; = 4[r/2Z�0;] sin²(πMP/N) and β = -ω0/ω . (38a)

For small β this leads to a Q-factor of

Q = NZ₀/2πr. (38b)

In the "birdcage" resonator design (Hayes et al., 1985), considered as a series of π sections, Z₂ is inductive, while Z₁ includes an inductor and a capacitor in series. This has a design disadvantage in that circuits with many elements require many bulky high-voltage capacitors for individual tuning.

High frequency probe

Miniaturization of RF probes for high frequency operation can be problematic because dimensions are often limited by the use of components with lumped parameters.

This work introduces a new tunable RF cavity design that is applicable at RF frequencies around 500 MHz or above, but could be scaled and adapted for lower frequencies.

The design of the RF cavity was inspired by the microwave magnetron cavity resonator. It is similar to the "birdcage" resonator (Hayes et al., 1985), but has the advantage that it can be precisely constructed from machined solid copper and rods. Theoretical The basis of our approach is what has been presented above.

The resonator consists of two end plates 10, each of which has a symmetrical array of slotted loop resonators 20 connected to a number of rod inductors 30, Figure 3. In Figure 4a, a top view of an end plate is shown. In a particular embodiment, each slotted loop resonator 20 has a diameter of 10 mm with an inductance of 11 nH and a gap 40 corresponding to a capacitance of 12 pF, giving a resonant frequency of 438 MHz. Leadless chip capacitors can also be used to increase the capacitance of the gap. An alternative end ring arrangement 10' is shown in Figure 4b. This provides greater axial access.

In the embodiment of Figure 4b, the end ring comprises a ring 10' having loop resonators 20' with slots or gaps 40'. The ring has an inner surface 41 and an outer surface 42 with the circular openings 20' formed therebetween. At locations central to the resonators 20', rods 30' are connected to an end surface 43. A second series of slots 45 are formed in the second end surface 44 toward the first end surface to accommodate a flux guide or sleeve as in Figure 4(d).

To prevent inductive coupling of the slotted loop resonators, a flux guide sleeve or end ring guide tube 50 may be fitted over each end ring 10. One such sleeve arrangement is shown in Figure 4d. It comprises two short coaxial conductive cylinders 11, 12, the inner cylinder 12 being held centrally within the outer cylinder by a series of conductive metallic spacers or fins 13. The fins 13 are arranged to engage the slots 45 between the slotted loop resonators of Figure 4b. The slots 45 must be insulated so that neither the fins 13 nor the flux guide rings 11, 12 affect the resonator. 10, Figure 4b. A suitable insulating material could be an insulating tape or a layer of varnish. With this or a similar arrangement, the magnetic flux from one slotted loop resonator 20' is prevented from coupling with other slotted loop resonators. Provided the annular space between the end rings 11, 12 of the guide tube 50 is large enough, a flux return path is provided for each slotted loop resonator 20, thereby keeping its self-inductance close to the unshielded value.

The characteristics of a single slotted loop resonator were determined empirically by measuring the inductance of a single loop formed by drilling a hole in a copper block. In Figure 5, the inductance is plotted against the hole diameter. When assembled, the complete coil resonates in a frequency range around 500 MHz. Tuning of this design can be done by hand by sliding an end plate along the rod inductances. The equivalent circuit for the resonator coil is shown in Figure 6a, with A and B connected.

The allowed frequency response for a section of this circuit (r1 = r2 = 0) is shown in Figure 6b as a hatched line. The cutoff bandwidth frequency is set by the parallel resonant elements, see below.

From the circuit parameters of Figure 6a and the above analysis with r₁ = r₂ = 0, it follows that the resonant angular frequency of the cavity ω (when driven correctly) is given by

= ω² = ω²₂{1 + L₂/4 L�1; sin²(πM/N)} (39)

For the main mode M = 1 and for N = 6 sections this simplifies to

ω² = ω²₂{1 + L₂/L₁} (40a)

with ω²₂ = 1/L₂C = 4π²f²₂ and with an inductance of the rods of L₁ = kl, where k is a constant and l is the length of the rods. With these substitutions, equation (40a) becomes

f² = f²&sub2;{1 + (L&sub2;/kl)} (4db)

where f is the cavity frequency. In general there will be a mutual inductance between the rods, but this is small and is therefore ignored in this application.

From equations 14 and 15 we find that for ω/ω2 > 1, Z0 (L1/C) 1/2. If we assume that ohmic losses in the cavity arise mainly in the rods, a reasonable assumption since the end plates are made of solid copper, r2 = 0 and in this case for ω/ω2 > 1

α; = [r₁/2Z�0](ω₂/ω)² . (41a)

However, if r₁ = 0 and the losses occur in the slotted loop resonators, we find that

α; = [r₂/2Z�0] (L₁/L₂){1 + (L₁/L₂)}(ω₂/ω))² . (41b)

In both cases, β = ω1/ω.

Results

Measurements of the resonance frequency for the main mode (M = 1) were made for different lengths of the cavity and it was found that the resonant behavior of the coil when properly driven was in agreement with theory, thus supporting our initial assumption that mutual induction effects between sections can be ignored in this case. The coil is constructed as above described in symmetric mode driven from one end. With N slotted loop resonators and N rod inductors, N/2 resonance modes are observed.

Driver circuit

The cavity can most easily be driven from an end plate, Figure 4a, via AB or A'B' through the split capacitor arrangement, Figure 4c. The center point is grounded and the coaxial driver connected to A or alternatively C (alternative drive and connection points are indicated by dashes). To ensure proper matching, a single capacitor of 1/2 Cd should be connected across the corresponding points at the other end of the cavity. The cavity frequency squared f is plotted against 1/1 in Figure 7 according to linear regression, equation 4db. The intercept at the origin gives an experimental value for the fundamental frequency f2 = 424 MHz.

The data are also plotted as f versus 1 in Figure 8. Using the measured parameters allows the theoretical curve to be drawn (solid line). It is also worth noting that when the input driver is connected to point A, Figure 4a, this forms an antinode or high voltage point. Apart from a phase shift of the RF carrier, the currents in the rods follow the voltage of the end plate, see Equation 22. The drive point therefore corresponds to an antinode of the current in the rods. Point B has the opposite RF phase. The additional capacitance Cd/2 is distributed around all the slotted loop resonators to produce an additional slotted loop capacitance of Cd/2N. This will therefore affect the fundamental frequency f2 in Equation 4db and Figure 7. This behavior is confirmed experimentally. This can therefore be used as an adjustment to fine tune the cavity.

Alternatively, the cavity length can be reduced to restore the desired operating frequency.

At 500 MHz, the coil Q was 160 and the cavity length was 4.0 cm. From equation 4db, we deduce that L1 = 7.48 nH, which produces a characteristic impedance of Z0 = 22.3 Ω. The input resistance from equation 29 is R = 1135 Ω. The cavity was matched to 50 Ω using a variable matching capacitor Cm of 6.8 pF from Figure 4c with Cd = 3 pF. Taking Cd into account, the theoretical closed loop resonant frequency is 429 MHz, in agreement with the measured value from Figure 7.

We stress that the input impedance Z₁ is correct for end-plate drive arrangements between points A and C, but will be 4 times larger for symmetrical drive across A and B. For the slotted loop cavity resonator, the drive circuit, Figure 4c, affects f₂ in the expected way via the additional distributed capacitance Cd/2N introduced into each slotted loop resonator. Up to this extent, the drive capacitance will predictably change Z₁ through its effects on Z�0, Equation 15.

Alternative cavity constructions

In an alternative low frequency cavity design, a portion of the end plate of which is shown in Figure 9, the capacitance of the slotted loops is increased by adding a sectioned guard ring 21. Alternatively, actual small capacitors 22 can be inserted between the slots of the loops.

In another arrangement, the loop inductance can be increased by slightly offset stacking machined end plates 10, 10' as shown in the schematic of Figure 10. In this arrangement, successive inductance layers must be effectively coupled in series as indicated. In order to avoid offsets To ensure that successive loops do not block the hole, the second 10' and subsequent layers of loops must be appropriately extended.

All the arrangements discussed so far are single turn cavities in which a standing wave around the transmission line structure obeys the phase relationship Nβ = 2πMP, where β is the phase shift per section, N is the number of sections, and M and P are integers. For a λ line, the operating frequency is P = 1. The main mode M = 1 means that there is exactly 2π radians of phase shift. However, if P = 2 at the same frequency, then this would imply a two turn structure as sketched in Figure 11. Such an arrangement would produce twice the RF field per unit current at the coil center and would thus provide a means of effectively increasing the resonator impedance by mutually coupling the inductive elements. A sketch of a portion of an end plate assembly for a two-turn cavity resonator is shown in Figure 12.

All inductive loops to date have been either flat and lying in the plane of the end plate or within an end ring arrangement. However, in an alternative arrangement of loops these can be rotated 90º out of the plane of the plate, Figure 13. In this arrangement the magnetic flux of the loops forms a toroidal shape which is effectively contained within a torus for a large number of elements. The loops must be arranged with sufficient spacing so that there is effectively no mutual inductance. Alternatively, flux protection plates can be introduced to magnetically separate the loops.

In a further modification of the cavity resonator, the π equivalent circuit per section of Figure 1a comprises a capacitor in series with an inductor for Z₂ and an inductor for Z₁. This gives a high-pass transmission line section, in which case the end plates of the cavity could be considered as connected by rod inductors connected divided ring. An end plate is outlined in Figure 14. Each section 60 is made from a solid block and the spaces 70 form a ring of capacitors connected in series. The sections can be suitably spaced with a dielectric material.

The above arrangement can be further modified as in Figure 1a. Here the inductance of the block can be increased by forming a slotted loop. The equivalent circuit for this arrangement is shown in Figure 1b. If C is small due to a wide slot (i.e. C = 0), the equivalent circuit reduces to that of Figure 14.

RF shielding

In some situations it may be desirable to surround the cavity resonator with an RF shielding can, thereby making the resonator characteristics independent of surrounding metal structures. The effects of the can will be to reduce the rod inductance and also introduce a stray capacitance Q which is shunted to each rod. The shunt capacitance can be easily built into the theory. The net effect is an increase in the operating frequency f for a given cavity length l. The magnitude of the effect depends on the proximity of the shield. For a shield/cavity diameter ratio of 1.25 the frequency change is about 15%. This can be compensated for either by increasing the resonator length or by increasing Cd, which reduces f2 as shown in the driver circuit description.

Split resonator designs

The resonator coil designs described so far are all cage-like arrangements that completely surround the object around the cylinder axis. However, there are a number of situations where it is more convenient to have a split coil system that provides easy access to the subject. Such an arrangement is desirable in the case of very small subjects and also in clinical imaging for easy access of limbs, torso, head, etc.

Such a new split resonator coil arrangement is sketched in Figure 16. We take here as an example only a half cavity arrangement 100 which is simply placed close to but not necessarily in contact with a large grounded conductive metal plate 102. Two standing λ/4 waves are generated around the driving point, provided that all four corners of the half cage are grounded. Because of the special symmetry of the resonator wires, a magnetic field parallel to the conductor surface is doubled and made uniform by the induced image currents in the plate. This arrangement will behave magnetically and electrically as if it were a single cylindrical resonator as described earlier, since the boundary conditions for an open λ/2 line are the same as for an open or cyclic λ line. In practice, however, it is possible to raise the coil in the manner of a cake or cheese plate cover, allowing easy access to the object. A half-cage coil design without the conductive plate was described by Ballon, D., Graham, M. C., Devitt, B. L., Koutcher, J. A., Proc. Soc. Mag. Res. in Med. 8 th Annual meeting, Amsterdam 2, 951 (1989), but is less useful due to poor RF homogeneity and a lower signal response.

The arrangement sketched in Figure 16 is semi-circular in cross-section. In general, however, it is possible to generate uniform transverse magnetic fields with a semi-elliptical structure in which the axes of the ellipse are 2a and 2b. Such an arrangement could be extremely practical as a head coil or as a leg or knee coil. The corners A and B can be connected via an internal return wire 104 which provides a continuation of the current path around the end plate. The corners P and Q should be connected accordingly by a wire 106.

If the number of wires in a circular cage is reduced to six, the coil structure effectively becomes a saddle coil arrangement, Figure 17a, in which wires w, w' do not carry current, i.e. are located at locations corresponding to wave nodes. If a reflective shield is introduced as in Figure 17b, wires 1 and 4 form a ground, while wires 2 and 3 can be connected since their currents are in phase. The arrangement is sketched in Figure 18a. To force wires 1 and 4 into the nodal plane, the circuit must be driven as indicated. If the device does not actually touch the shield plate, current return paths in the form of internal return wires 104', 106' can be provided between points A and B and points P and Q as sketched. In a further modification, the pair of wires is replaced by a single conductor strip 110, Figure 18b, and the rods 2 and 3 are combined together to form a flat conductive plate 112. In either case, this arrangement provides a removable coil system useful for flat objects in either microscopy or whole body imaging.

The presence of the conductive plate serves to make the arrangement symmetrical and thereby increase the field while making it more uniform. The equivalent circuit and driving arrangement are shown in Figure 18c.

The alternative end plate design of Figure 4b can be halved along its diameter to produce a split or half resonator design, replacing the two half end plates shown in Figure 16. To overcome the mutual inductance between the loops, the end ring guide tube of Figure 4d can also be halved to fit the end rings of the half resonator.

In the embodiment of Figures 16 and 18, the conductive plate 102 is preferably not a continuous plate, but may comprise a plurality of stripes as indicated by dashed lines 102' in Figure 16. This serves to satisfy the RF current boundary conditions but to block other induced currents at lower frequencies that would otherwise be caused by the switched gradients used in NMR. The stripes may be formed from a continuous plate and slots at appropriate intervals. Alternative arrangements include cutting the plate into appropriately shaped flat loops that follow the contours of the induced RF current in an otherwise continuous conductive plate.

In a further modification, the passive conductive plate is replaced by an actively driven flat wire array that is powered to simulate the induced shielding currents in a flat passive conductive plate.

Using the matrix approach, we have designed a cavity resonator NMR coil operating at 500 MHz. The developed analysis ignores mutual inductive effects between the sections of the transmission line. The experimental results obtained with the resonator coil confirm the theoretical expectations for the fundamental mode. The general frequency properties of the higher resonant modes are also sufficiently well described by the theory, although there are differences between the observed relative frequencies and those predicted by the theory. These could well be due to the ignored mutual inductive effects. However, in NMR applications, we are generally only interested in the fundamental mode, since the higher modes in these resonant structures generate spatially inhomogeneous RF fields. A symmetric driver arrangement gives the best performance of the circuit when applied from one end of the cavity resonator.

The idea of a split resonator design is introduced, in which a resonator array is placed close to, but not in contact with, a flat conductive plate. Since the cross-section of the split coil can be semicircular or semi-elliptical and is not attached to the plate, the entire structure is removable, allowing easy access for imaging of limbs, head or whole body.

A variant of the split coil design is also described, which corresponds to a half-saddle arrangement. This can be further modified to produce a strip coil in proximity to an insulated conductive plate.

Claims (12)

1. A resonant assembly for NMR for use at high frequencies comprising two identical end structures and a plurality of continuous electrically conductive rods electrically connected to the identical end structures, the rods thereby maintaining the end structures at a predetermined distance from each other, characterized in that the rods (30) and the end structures (10) form a plurality of π or T electrical circuit sections and are shaped and dimensioned so as to contribute substantially as a whole to the electrical characteristics of the assembly and enable it to support a standing wave. 2. A resonance arrangement for NMR as in claim 1, wherein each end structure comprises a machined plate (10). 3. A resonant assembly as in claim 2, wherein each end plate (10) comprises a substantially circular disc having circular apertures (20) evenly spaced on the surface of the disc, wherein each aperture has one end of an elongate channel (40) extending into a common central region of the plate connected to it, the other end of each elongate channel being connected to form a central aperture, and wherein each of the plurality of continuous rods is connected to the plate (10) at a position midway between each elongate channel (40) to form the assembly. 4. A resonance arrangement as in claim 2, wherein the rods are connected to an end plate by means of a plurality of through holes in the end plate, each rod being slidably mounted in its respective hole in the end plate to allow adjustment of the distance between the end plates. 5. A resonance arrangement as in claim 2, wherein each end plate comprises a main ring (10') having inner (41) and outer (41) cylindrical surfaces defined by the thickness of the main ring and first (43) and second (44) end surfaces defined by the width of the main ring, the main ring being provided with a plurality of circular openings (20') evenly spaced around its surface and providing circular holes through the main ring from the inner to the outer surface, the main ring also being provided with a plurality of first slots (45') connecting each circular opening to the first end surface and a plurality of second slots (45) formed in the main ring from the second end surface towards the first surface so as to partially separate the circular openings, and a respective rod (30') of the plurality of rods having the main ring is connected at a first end surface (43) at a respective position between the first slots (40'), the rods being arranged at an equal distance around the ring. 6. A resonant arrangement as in claim 5, including a flux guide sleeve (50), the flux guide sleeve (50) comprising an inner ring (12) and an outer ring (11), the inner ring (12) having a smaller diameter than the main ring and the outer ring having a larger diameter than the main ring (10'), wherein the inner (11) and outer (12) rings are connected to one another by a series of fins (13), wherein the fins (13) and the inner and outer rings are electrically conductive and wherein the fins are arranged at spaced intervals around the rings so that they, together with the second Slots (45) allow the flux guide sleeve to be slid over each ring. 7. A resonant arrangement, wherein the arrangement comprises a first and a second arrangement as in claim 3, wherein the first arrangement is larger than the second arrangement and the second arrangement is located within the first arrangement. 8. A resonant assembly as in claim 1, wherein each end plate comprises a ring, the ring being constructed from a plurality of electrically conductive segments (60), each segment being separated from its next adjacent segments by an electrical insulator (70) to thereby electrically isolate each segment, and in which a respective rod (30) is connected to each segment to thereby evenly space the rods around the ring. 9. A resonant assembly as in claim 1, wherein each end plate comprises a half ring, the rods thereby interconnecting each half ring to form a "cheese plate assembly" (100) and including an electrically conductive plate (102) on which the "cheese plate assembly" is placed but does not contact the conductive plate. 10. A resonant arrangement as in claim 9, wherein each half ring is provided with a plurality of circular openings therethrough and a plurality of elongated channels connect each circular opening to an outer surface of the half ring. 11. A resonance arrangement as in claim 1, characterized in that first, second, third and fourth rods are provided and the end structure each comprises a partial ring to which the second and third rods are connected, that the second and third rods each are electrically connected to the first and fourth rods via capacitors at each end of the rods, that an electrically conductive plate is provided close to but conductively separated from the first (AP) and fourth (BQ) rods, and that conductors (104', 106') connect the ends of the first and fourth rods directly to one another. 12. A resonant arrangement as in claim 11, wherein the second and third rods and the partial rings to which they are connected form a first continuous conductive plate (112) and the first and fourth rods and the conductors connecting their ends form a continuous conductive strip (110).