DE3789328T2 - Method and system for determining the position of a moving platform, for example a ship, using the signals from GPS satellites. - Google Patents

Description

The present invention relates generally to a method and apparatus for measuring position on the earth from a moving platform, such as a ship, and more particularly to a method or apparatus using signals from the NAVSTAR Global Positioning System satellites, commonly referred to as GPS satellites. In particular, the present invention, as defined in the appended claims 1 and 24, is applicable to civil GPS receivers, i.e. receivers which do not use knowledge of the potentially unavailable P-code component of the GPS signals to determine position information.

Conventional civilian GPS receivers use simultaneous "pseudorange", in other words, group delay observations of the C/A code components of the L1 band signals received from multiple GPS satellites to determine position information. A major source of position errors in such conventional GPS receivers is multipath. Multipath errors can be reduced by time averaging observations made from a fixed position. However, conventional time averaging cannot be used to increase the accuracy of ship-based receivers because the resulting position information would relate to the ship's average position during the observation period, not its instantaneous position.

Position errors also arise due to group delay effects in the ionosphere in such pseudorange measurements. The amount of delay a signal experiences in the ionosphere varies depending on local conditions and cannot be predicted with sufficient accuracy to be eliminated from position measurements made by GPS receivers. However, the magnitudes of such errors are frequency dependent, and can be determined from simultaneous measurements of signals in different frequency bands. In particular, the GPS system was designed so that simultaneous measurements of signals in the L1 and L2 bands could be used to determine the ionospheric delay. This approach is routinely used in military GPS receivers. Conventional civilian receivers measure the C/A code group delay in the L1 band, but cannot make L2 band C/A code group delay measurements because C/A code modulation is not currently used on signals transmitted in the L2 band. Therefore, simultaneous measurements for both L-band signals cannot be easily made.

In general, conventional civilian receivers have limited accuracy due to their dependence on group delay and because they require knowledge of a code that modulates the signals in a GPS band in order to measure the group delay of signals received in that band. For example, U.S. Patent No. 4,578,678 estimates group delay using knowledge of either the P or C/A code.

Various aspects of the present invention are set forth in the appended claims.

Examples of the present invention provide a method and apparatus for accurately determining position information using signals transmitted by GPS satellites and received on mobile platforms, such as ships, independent of a knowledge of the P-code component of the signals. An accurate measurement of the ship's distance to each satellite is made based on the L1 center frequency carrier phase. A correction for ionospheric effects is determined by simultaneously observing the group delays of the wide bandwidth P-code modulations in both the L1 and L2 bands. These group delays are determined by measuring the phases of carrier waves contained in the broadened spectrum signals received in both bands. These carriers are reconstructed from the L1 and L2 band signals from each satellite without knowledge of the P-code. The unknown systematic errors in the range measurements of the L1 center frequency carrier phase are determined from simultaneous time-averaging pseudorange measurements. The instantaneous position of the ship can then be determined from the ranges thus determined, with both the systematic errors and the ionospheric effects eliminated.

For a better understanding of the present invention, and to show how it may be carried into effect, reference will now be made, by way of example, to the accompanying drawings in which:

Fig. 1 shows a system for determining position information using signals received on a ship from GPS satellites in accordance with a preferred embodiment of the present invention;

Fig. 2 is a block diagram of a terminal for receiving GPS signals, which can be used both on board ships and on land-based devices, corresponding to the system shown in Fig. 1;

Fig. 3 is a more detailed block diagram of the receiver subsystem and associated elements of the terminal shown in Fig. 2;

Fig. 4 is a detailed signal flow diagram of the radio frequency to video frequency downconversion subsystem of the receiver shown in Fig. 3;

Fig. 5 is a detailed signal flow diagram of one of the fo carrier reconstruction and downconversion subsystems of the receiver shown in Fig. 3;

Fig. 6 is a block diagram of one of the satellite tracking channels shown in the terminal shown in Fig. 2;

Fig. 7 is a block diagram of the 308 fo carrier phase and C/A code delay error detector subsystem of the satellite tracking channel of Fig. 6;

Fig. 8 is a series of graphical representations of phasors in the complex plane representing signals in various other figures;

Fig. 9 is a block diagram of one of the counter rotators shown in the error detection system of Fig. 7 and truth tables relating to their operation;

Fig. 10 is a block diagram of the correlator used in the error detection system shown in Fig. 7;

Fig. 11 is a block diagram of the C/A code generator used in the error detection system shown in Fig. 7 is shown;

Fig. 12 is a block diagram of the fo carrier phase detector included in the satellite tracking channel shown in Fig. 6;

Fig. 13 is a block diagram of the register subsystem of the satellite tracking channel shown in Fig. 6;

Fig. 14 is a detailed signal flow diagram of the overall system shown in Fig. 2; and

Fig. 15 shows alternative embodiments of observable combining means for combining the implicit carrier phase observable φ for 308 fo, the C/A code group delay observable τ, the implicit fo carrier phase observable ψL1 for the L1 band, and the implicit fo carrier phase observable ψL2 for the L2 band to unambiguously determine a position that is relatively free of ionospheric disturbances.

1. The GPS system

The following description of the GPS system and the structure of its signals is provided for clarity and to facilitate explanation of the invention. More detailed discussions are available in the literature.

The signals of primary interest for determining position are transmitted by each GPS satellite in the same two frequency bands, one known as the L1 band and the other as the L2 band.

Within each of these GPS bands, the transmitted signal is a broadband, noise-like, pseudostatic signal, which does not contain any discrete spectral components. The signals are therefore referred to as suppressed carrier signals.

The term "carrier" is used here in the same sense as in the radio field, so that a carrier is a periodic wave of essentially constant amplitude, frequency and phase. Information can be transmitted or "carried" by varying the amplitude, frequency and/or phase of such a signal. A carrier can be called a "subcarrier" if its frequency is less than the bandwidth of the signal. A signal can have several carriers. For example, a television broadcast signal is said to contain a video carrier and an audio carrier.

Although no carriers are present in the transmitted GPS signals, it can be said that various carriers are implicitly contained therein, so that such carriers can be recovered or reconstructed from the GPS signal.

Within each GPS satellite, a frequency standard, such as a cesium atom beam device, provides a fundamental frequency of 5.115 megahertz, designated fo, from which all other critical satellite frequencies are derived by integer multiplication or division. The L1 band center frequency carrier frequency of GPS signals is 308 times fo, or 1575.42 megahertz, and the L2 band center frequency carrier frequency is 240 times fo, or 1227.60 megahertz. The fo fundamental frequency is a carrier frequency that can be reconstructed from the GPS signals.

GPS signals are biphasic or quadriphasic modulated. In particular, quadrature components of an L-band center frequency carrier can be mixed in the satellite with aperiodic, pseudo-statistical, binary weighted waves m(t) and n(t) containing several implicit periodic carrier waves. The polarity or phase reversals of m(t) and n(t) are constrained to occur only at times that are integer multiples of fixed time intervals tm and tn, known as the slice widths of m(t) and n(t), respectively. If a polarity reversal occurred at every multiple of tm, then m(t) would be a periodic square wave with a frequency of 1/(2 tm) Since the polarity reversals actually only occur pseudo-randomly, on average only half the time, the 1/(2 tm) frequency carrier wave is suppressed, as is the carrier of the band center frequency.

The wave m(t) is the product of several different waves whose time intervals between polarity reversals are integer multiples of tm. Therefore, additional carriers are also implicitly included in the GPS signals, whose frequencies correspond to submultiples of 1/(2 tm). One of the waveforms used to generate m(t) for the L1 band signals is the C/A code, which is a satellite-specific pseudostatic binary sequence of 1023 pieces repeated periodically with a period of 1 millisecond, or at a frequency of 1 kilohertz. For the C/A code component of the L1 band signals, 1/(2 tm) is equal to fo/10 or 0.5115 megahertz. Another factor in m(t) is the binary navigation data, which has a piece width of 20 milliseconds, and therefore a carrier frequency of 25 hertz. This data includes the current time shown by the satellite clock, a description of the current Position of the satellite in the orbit, and a description of any corrections to be made to the time given by the satellite's clock.

The conventional method for determining the position of a receiver from signals received from GPS satellites uses observations of the times indicated by the satellite clocks as reflected in the transmitted binary navigation data. Typically, four different satellites are observed simultaneously. As explained below with reference to Fig. 8E, at a time when a clock in the receiver indicates a time tR, the modulation of the signals received from the i-th satellite includes in its navigation data a satellite clock indication ti, and satellite position coordinates xi, yi, and zi. For ease of description, it is assumed that any indicated clock corrections have already been applied.

If the receiver's clock were synchronized with that of the satellite, the difference between tR and ti would be exactly equal to the signal propagation time between the satellite and the receiver. The distance or "range" from the satellite to the receiver could then be determined by multiplying this time difference by the speed of light. Since the clocks are not synchronized, the range determined by this method has a systematic error of unknown magnitude, equal to the deviation from synchronization tR multiplied by the speed of light. Such range measurements are known as "pseudorange" measurements.

ΔtR can be determined simultaneously with the three position coordinates xR, yR and zR of the receiver by that four simultaneous observations of the pseudorange are made from four suitably positioned satellites, and four coupled algebraic equations are solved concerning these four unknowns according to known procedures.

Such conventional measurements of the apparent propagation time delay, known as group delay, are limited by multipath and ionospheric effects. The term "multipath" refers to the simultaneous reception of signals that have reached the receiving antenna via different paths from the transmitting antenna. Harmful multipath effects most often arise due to reflection of signals from objects near the receiver. Even weak multipath effects can severely limit the position accuracy available from pseudorange receivers.

Multipath effects often occur in the form of statistically varying noise and can be reduced by simply time averaging the observations as long as the position of the receiver is fixed. Furthermore, it is known that multipath effects generally affect the phase delay of a signal significantly less than the group delay. Therefore, the variation in the range resulting from the center frequency carrier phase is significantly less than that of the range obtained from the code modulation of a GPS signal.

The speed of propagation of a radio signal through the ionosphere differs from the speed of the same signal through other media, such as the vacuum of space or the Earth's troposphere. Ionospheric propagation effects limit the accuracy of conventional, civil Position measurements made using pseudorange measurements because the ionosphere can delay the propagation of the signal modulation by an unknown amount. This amount can vary with time and location in unknown ways, but is frequency dependent in a precisely known way. The group delay imposed by the ionosphere on a given propagation path from the satellite to the receiver at a given time is known to be inversely proportional to the square of the center frequency of the band in which the signal is received.

2. The preferred embodiment Fig.1

In Fig. 1, a system according to a preferred embodiment of the present invention is shown for determining the position of a vessel 10 from observation of the plurality of GPS satellites represented by GPS 12, GPS 13 and GPS 14 that are currently visible to the vessel 10.

According to the current design of the GPS system, a minimum of 18 GPS satellites will be available by about 1988. The way they are arranged will ensure that at least four satellites will be visible at all times from any point on the earth. The present invention can best determine position information when at least four satellites are visible during an observation period. Satisfactory measurements can be made when only three satellites are visible at the same time if an unknown does not have to be determined, for example the height of the phase center of the receiving antenna above sea level.

During operation of the system according to the invention, measurements are made on the ship 10 of at least four observables of GPS signals 15 transmitted by each of the GPS satellites, for example GPS12, GPS13 and GPS14, and received on the ship via the antenna 22. These four observables are:

a) the observable Φ for the implicit 308-fo carrier phase of the L1 center frequency carrier;

b) the C/A code group delay observable τ of the C/A code component of the L1 band signals (also known as the pseudo-orange region);

c) the L1 band implicit fo carrier phase observable ΨL1 of the fundamental frequency fo subcarrier implicitly contained in the signals received in the L1 band; and

d) the observable ΨL2 for the implicit fo carrier phase of the L2 band of the fundamental frequency fo subcarrier contained in the signals received in the L2 band.

These observables may be processed either in data processing facilities on board the ship 11 or in a data processing center 20, either in real time or retrospectively, to determine position information. In addition, shore stations 16, 17 and 18 may also be used to receive GPS signals 15 via antennas 22 and determine position information, as well as other specific data relating to the GPS satellites. This additional information and/or data may include ephemeris information or orbital position information for each satellite, as well as required Satellite clock error or synchronization information that needs to be determined.

At present, satellite ephemeris information and satellite clock synchronization information are included in the navigation messages contained in the signals transmitted by the GPS satellites and can be decoded and used by civilian users of the GPS system. It is possible that such information may not always be satisfactorily available, either due to limitations in their accuracy or due to deliberate encryption of the information for military reasons. In any event, for high-accuracy civilian operations it is desirable to be able to determine orbit information and clock error or synchronization information without having to rely on the transmitted orbit information. This can be achieved by several ground stations working in cooperation with the receivers on board ships.

Shore stations 16, 17 and 18 may include data processing subsystems, or such subsystems may be provided on the ship 10 or in a separate data processing center 20, as shown in Fig. 1. The shore stations 16, 17 and 18 may communicate with each other, with the ship 10 and with the data processing center 20 via a data link 21. The data processing center 20 may further include radio communication devices for transmitting data to and from the ship 10 as part of the data link 21, although such radio communication may not be required to determine position information for some purposes. Raw data may be collected on the ship 10 and later processed by any suitable means, not shown. facility to the data processing center 20 for subsequent processing. However, if the position information is determined for navigation purposes in real time for use by the ship 10 in maintaining its own course, the above-mentioned radio communication link may well be necessary for the proper operation of the system.

The shore station 18 may also be used to allow relative position measurements, that is, measurements of the position of the vessel 10 with respect to the shore station 18. Such measurements are also called "line-of-position measurements," where the relative position vector extending from the shore station 18 to the vessel 10 is referred to as the line-of-position vector. It is known that such relative measurements can be significantly more accurate than single point position measurements made by the vessel 10 alone, since certain measurement errors occur together at both ends of the line of position and can cancel out when the observations at the vessel 10 and the shore station 18 are combined.

Conventionally, one end of a position line is chosen to be a known location, such as a survey point. In the systems described here, it is most convenient to locate the shore station forming one end of the position line at the known location. The position of the vessel can then be calculated as the end point of the position line vector emanating from the known location. Of course, a relatively short distance between the vessel 10 and the shore station 18 is desirable. This may not always be the case in practice. The only real limitation on the length of the position line, i.e. the maximum permissible distance between the vessel 10 and the nearest shore station, is that the same GPS satellite is used for both ends of the The base line must be visible if simultaneous observations at both ends are to be directly evaluated to cancel out common errors.

The terminal sections of the shore stations 16, 17 and 18 may be constructed and/or operated identically to the terminals on the ship 11. However, it may be preferable for the shore stations 16, 17 and 18 to measure additional observables not required on the ship 10, including observations of the phase of the center frequency carrier of the L2 band signals. Simultaneous observation of the phases of the center frequency carriers of both bands may result in a more accurate determination of the ionospheric delay, and hence of the satellite orbits and the position of the ship 10. The terminal 23 of the ship 10, which as mentioned above may also be used in the shore stations 16, 17 and 18, will be explained in more detail with reference to Fig. 2.

Fig.2

In Fig. 2, the terminal 23 is shown in block diagram form. As previously mentioned, identical terminals 23 may be used both in the shore stations 16, 17 and 18 and on board the ship 10, although it may be preferable to provide additional equipment within the terminals 23 on the ship 10 or in the shore stations 16, 17 and 18, for example uncoded L2 center frequency carrier phase tracking channels for determining orbital data.

The terminal 23 is preferably located in a protected equipment room in the ship 10, together with other complex electronic ship equipment. However, it must be connected to the Antenna 22 mounted above the deck of vessel 10, preferably on a high mast (not shown), above and as far away as possible from potentially interfering superstructures, such as metal deck plates and superstructures, which could exacerbate errors caused by multipath effects.

The antenna 22 receives GPS signals 15 from the several GPS satellites visible at the time, of which only the GPS satellite 12 is shown in this Figure 2. L-band GPS signals 15 are fed from the antenna 22 to the receiver 24 within the terminal 23, which also receives the 4-fo constant frequency reference signal 26 generated by a frequency standard 28. The frequency standard 28 may be a specially designed cesium atom beam standard that provides a constant frequency output signal exactly at 20.46 megahertz, four times the fundamental frequency fo of the satellite. On the other hand, it may be simpler and cheaper to use a more readily available 5 megahertz cesium atomic beam standard and a frequency synthesizer, not shown, to convert the 5 megahertz standard signal to the desired constant frequency reference signal of 4 fo at 20.46 megahertz.

Receiver 24 is used to down-convert GPS signals 15; receiver 24 therefore serves to provide output signals 30 which are relatively low in frequency compared to the L-band microwave radio signals received by antenna 22 from GPS 12. Signals 30 are fed to multiple satellite tracking channels 32. A single satellite tracking channel 32 is required for each satellite to be observed in a given observation period.

The satellite to be observed by each satellite tracking channel 32 is determined by the satellite assignment 41 from the real-time computer 40. The terminal 23 may have six satellite tracking channels to provide redundancy, although only four satellite channels are required in normal operation. It is good operational practice to track more than the minimum number of satellites at any one time so that if one of the satellites becomes occulted, or sinks below the horizon, or is otherwise unavailable or inoperable during part of the observation period, the data collected will still be sufficient.

Furthermore, the tracking channel 32 continuously receives a real-time digital indication 34, t, generated by the real-time clock 36. The real-time clock 36 is controlled by the frequency standard 28, which provides a clock synchronization signal 29. The time t is obtained from the clock 36 in a conventional manner, by an initial real-time setting or determination, and subsequent counting of oscillation cycles of the signals from the frequency standard 28.

The operation of all tracking channels 32, one for each of the GPS satellites, is the same. The operation of the tracking channel 32 shown in Fig. 2, which operates on GPS signals 15 from the GPS satellite 12, will now be explained. From this, the operation of the tracking channels 32 for other GPS satellites can be understood. In addition to the low frequency signals 30, the 4-fo reference signal 26, and the real-time digital display 34, the tracking channel 32 receives estimates 38, which include estimates or predictions of the received signal frequency and group delay for the GPS satellite 12, from the real-time computer 40. The tracking channel 32 sends measurements 42 relating to the Signals from this satellite are returned to the real-time computer 40.

The real-time computer 40 generates estimates 38 based on the input of external information 44 received from external sources and also on the information contained in the measurements 42. Such external information 44 may include data regarding the orbital position of the satellite 12 and also data concerning the corrections that must be made to the timing information contained in the signals 15. Orbit and timing correction information 46 shown in Fig. 2 may represent information transmitted over the data link 21 from land-based locations, for example from the data processing center 20 as shown in Fig. 1. Information 46 could also be obtained by the terminal 23 from the transmitted navigation message contained in the GPS signals 15 from the satellite 12, by decoding the so-called navigation message transmitted by the GPS satellite 12, but could also originate from any other device.

The real-time computer 40 may also receive information from observations made by one or more of the shore stations 16, 17 and/or 18 regarding the signals 15 from the satellite 12. This information preferably comprises data representing the phases of carrier signals implicit in the GPS signals 15 received by that shore station. In addition, such data may include pseudorange measurements related to the satellite 12 and determined by observing and measuring the group delay of the C/A code component of the L1 band signals in the signals 15 received from the satellite 12. This information is in 2 as phase and delay observations from the shore stations 48. Furthermore, the real-time computer 40 may also receive information relating to the current position and/or speed of the vessel 10 from the vessel position and speed information 50. Such information may be provided by any of the conventionally available commercial devices commonly used to determine such information, for example, conventional navigation and docking equipment.

The real-time computer 40 uses the measurements 42 and the information input 44 from the sources 46, 48 and 50 to calculate an improved determination of the current or past position 52 of the vessel 10. The position determination 52 may be provided to the position information display or use device 53. The real-time computer 40 may also provide raw measurement data 54 from the satellite tracking channel 32 to the data storage 56. Raw data 54 from the data storage 56 may be processed at a later time, on board the vessel 10 or onshore at the data processing center 20, to provide improved position information based on measurements or other data not conveniently available on the vessel 10 during observation of the satellite 12, for example, improved satellite orbit or clock information. Accordingly, phase and delay observations from some other shore stations not shown may be available in later processing that were not available during the observation period. Even improved data concerning the position and/or speed of the vessel 10 may be used in post-processing to improve the accuracy of the position information.

Conventional GPS receivers do not utilize external information inputs, such as those discussed above with respect to external information 44. Conventional GPS receivers obtain all such satellite orbit and satellite clock information directly from the navigation message transmitted by each satellite. It is particularly important that terminal 23 can utilize external inputs, as discussed above, to maintain operation and accuracy during times when the Department of Defense elects to degrade certain portions of the transmitted information from the GPS system to inhibit the high accuracy normally available to military users. Although the accuracy available to users of a system in accordance with the present invention is not sufficient for military operations, civilian users can continue surveying and navigation operations without interruption.

It has been announced that such accuracy blocking may include intentionally broadcasting inaccurate information regarding GPS satellite orbits and/or satellite clocks. Broadcasting inaccurate information regarding satellite clocks is commonly referred to as dithering.

The generally expected way of jittering satellite clock information would be to intentionally change the clock rate in the GPS satellite 12, or to allow the normal variations in the clock rate to affect accuracy, and then restrict access to the clock correction information. As explained above, each GPS satellite, in transmitting navigation data, transmits data that includes both the time provided by the satellite's clock as well as the detailed information concerning the corrections required to use this transmitted indicated time accurately. A simple approach to dealing with the jitter would therefore be to block public access to such clock correction information by encrypting the clock correction data contained in the satellite's transmitted clock correction message.

Other sources of external information 44 may include inertial navigation systems, compass, water speed indicator, Doppler sonar, or information regarding the orientation of the vessel 10, i.e. its orientation in space, roll, pitch and yaw. As the vessel 10 rolls and pitches, its antenna, which is mounted on top of a mast, moves around in space. The position determined by the computer 40 is therefore the position of this moving point at the top of the mast. This additional information may be required to determine the center of gravity of the vessel 10.

Fig.3

In Fig. 3, the receiver 24 of the terminal 23 is shown in more detail, together with the antenna 22 and the separate filter and preamplifier package 62, which is preferably located very close to the antenna 22. The remote filter and preamplifier package 62 serves to connect the antenna 22 to the receiver 24.

In particular, the transmission line 60 carries the L1 and L2 band radio frequency signals received by the antenna 22 to the remote filter and preamplifier package 62. The preamplifier 62 is preferably spatially located close to the antenna 22 to minimize transmission line losses in the line 60 between the antenna 22 and the low noise amplifier 68 within the preamplifier 62. It is useful to place the preamplifier 62 on the same mast or structure on which the antenna 22 is located.

In preamplifier 62, the L-band signals in line 60 are coupled to the inputs of bandpass filters 64 and 66. Filter 64 is a high quality, low loss bandpass filter tuned to the approximate center of the L1 band at 1575.42 megahertz. The bandwidth of filter 64 may be on the order of 30 megahertz. Filter 66 is tuned to the center of the L2 band at 1227.60 megahertz with a corresponding bandwidth. Filters 64 and 66 serve to prevent strong signals originating outside the GPS frequency bands from reaching the sensitive low noise amplifier 68 and overloading or possibly damaging it. Bandpass filters 64 and 66 are commercially available from various vendors and should be selected to provide low attenuation in the L1 and L2 frequency bands and high attenuation outside these bands.

The outputs of the filters 64 and 66 are combined at the input of the low noise amplifier 68. The amplified L-band signals present at the output of the low noise amplifier 68 are fed to the receiver 24 via the transmission line 70. The receiver 24 is conveniently located within the vessel 10 in a protected area, typically in the same environment used to protect other electronic equipment. The transmission line 70 can therefore be relatively long. The low noise amplifier 68 serves to amplify the L-band signals input to the transmission line 70 to a level sufficient to compensate for the attenuation of the line 70. The low noise amplifier 68 may be a commercially available gallium arsenide field effect transistor amplifier having a low noise figure of about 10 dB.

The receiver 24 receives the output of the low noise amplifier 68 via the transmission line 70 as one input and the 4-fo constant frequency reference signal 26 from the frequency standard 28 shown in Fig. 2 as a second input. Within the receiver 24, the transmission line 70 is connected to the input of a diplex filter 72. The diplex filter 72 is a frequency selective signal splitter which splits the L-band radio frequency signals received via the transmission line 70 into L1-band radio frequency signals 74 and L2-band radio frequency signals 78.

L1 band radio frequency signals 74 are applied to the radio frequency/video frequency down converter 76 where they are mixed with quadrature components of a reference signal at the nominal center frequency of the L1 band, namely 1575.42 megahertz, which is exactly 308 times fo. These quadrature components are derived from the 4 fo reference signal 26. The operation of the down converter 76 is described in more detail below with reference to Fig. 4. Generally speaking, the down converter 76 serves to convert the L1 band radio frequency signals 74 from a center frequency of 308 fo or 1575.42 megahertz to the baseband, i.e. a center frequency of 0 hertz. As will be explained in more detail below with reference to Fig. 4, the video frequency downconverter 76 produces an in-phase video signal 82 VI and a quadrature video signal 84 VQ which together form the complex video signal V.

Quadrature video signals 82 and 84 are included in the low frequency signals 30 and are used in each satellite tracking channel 32 in the terminal 23 and also by the L1 band fo carrier reconstruction and downconversion device 86 in the receiver 24.

The L1 band fo carrier reconstruction and downconversion device 86 is described in more detail below in connection with Figure 5. The reconstruction device 86 generally serves to reconstruct carriers of the satellite fundamental frequency fo that are implicitly contained in the L1 band radio frequency signals 74. Satellite fundamental frequency fo carriers from each GPS satellite are implicitly contained in the L1 band radio frequency signals 74 from the diplex filter 72, and in the quadrature video signals 82 and 84, since they were implicitly contained in the GPS signals 15, and none of the operations performed on the signals 15 up to this stage have altered this implicit relationship.

It is important to note that the converter 76 and the reconstruction device 86 use reference frequencies which are derived by fixed multiplication of the constant frequency 4-fo reference signal 26 from the frequency standard 28 of Figure 2, so that all of these signals are in phase with each other.

The L1 band fo carrier reconstruction and downconversion device 86 produces an in-phase reconstructed L1 band fo carrier composite signal 88, SI(L1), and a quadrature phase reconstructed L1 band fo carrier composite signal 90, SQ(L1), which together form the L1 band reconstructed fo carrier complex composite signal S(L1). The reconstructed carrier composite vector 89 of the L1 band reconstructed fo carrier 8912 shown below in Fig. 8D is the instantaneous value of the complex composite signal S(L1) of the L1 band reconstructed fo carrier resulting from the superposition of the phasors representing the L1 band reconstructed fo carriers from all GPS satellites in view. As shown in Fig. 8D, the GPS reconstructed L1 band fo carrier phasor 8912 represents the particular L1 band reconstructed fo carrier from the GPS satellite 12.

The in-phase and quadrature reconstructed L1-band fo-carrier composite signals 88 and 90 are very low frequency signals that are filtered within the reconstruction device 86 to a bandwidth of approximately 1 kilohertz.

L2 band radio frequency signals 78 are applied to down converter 80 where they are mixed with quadrature components of a reference signal at the nominal center frequency of L2 band, 1227.60 megahertz, which is exactly 240 times fo. These quadrature components are also derived from the 4 fo reference signal 26. The operation of down converter 80 is essentially identical to the operation of down converter 76 and can be understood in more detail by reference to the description of down converter 76 in Fig. 4. In general, down converter 80 serves to convert L2 band radio frequency signals 78 from a center frequency of 240 fo, or 1227.60 megahertz, to baseband. As will be described in detail with respect to Figure 4, the L2 band video frequency downconverter 80 produces an in-phase L2 band video signal 92 and a quadrature phase L2 band video signal 94.

The in-phase and quadrature L2-band video signals 92 and 94 are derived from the L2-band fo carrier reconstruction and downconverter 96. Reconstructor 96 operates identically to reconstructor 86 and can be understood from its description with reference to Fig. 5. Generally speaking, reconstructor 96 serves to reconstruct the satellite fundamental frequency fo carriers implicitly contained in L2 band radio frequency signals 78. Satellite fundamental frequency fo carriers from each GPS satellite are implicitly contained in L2 band radio frequency signals 78 from diplexer 72, and in in-phase L2 band video signal 92 and in quadrature phase L2 band video signal 94 because they were implicitly contained in GPS signals 15 and none of the operations performed on GPS signals 15 up to this stage have altered this implicit relationship.

The L2 band fo carrier reconstruction and downconversion device 96 produces the in-phase reconstructed L2 band fo carrier composite signal 98, SI(L2), and the quadrature phase reconstructed L2 band fo carrier composite signal 100, SQ(L2), which together form the L2 band reconstructed fo carrier complex composite signal S(L2). The in-phase and quadrature reconstructed L2 band fo carrier composite signals 98 and 100 are very low frequency signals that are filtered within the reconstruction device 96 to a bandwidth of approximately 1 kilohertz.

Fig.4

In Fig. 4, the downconverter 76 of the receiver 24 of Fig. 3 is shown in more detail. As previously explained, the converter 76 serves to downshift L1 band signals 74 in the same manner as the converter 80 serves to downshift L2 band signals 78.

L1 band radio frequency input signals 74 from the diplex filter 72 are applied to a pair of identical double phase mixers 110 and 112. A local oscillator input signal 140 is applied to the mixer 110 from the frequency multiplier 116 which multiplies the 4-fo reference frequency signal 26 of the standard 28 by a factor of 77. This operation produces the local oscillator input signal 114 at a frequency of 1575.42 megahertz. The corresponding frequency multiplier in the L2 band video frequency downconverter 80 uses a multiplication factor of 60 to achieve a frequency of 1227.60 megahertz.

The mixer output signal 122 contains signal components with frequencies equal to the sum and difference of the frequencies of the input signals 114 and 74.

The signal components of the mixer output signal 122 at the sum of the frequencies of the input signals to the mixer 110 are approximately 2 times the L1 center frequency. These signals are generally referred to as sum frequencies, and are rejected by the video low pass filter 124.

The signal components of the mixer output signal 122 that are equal to the frequency difference between the frequencies of the input signals to the mixer 110 are commonly referred to as the beat frequencies or difference frequencies. The beat frequency components of the mixer output signal 122 have frequencies equal to the difference in frequency of the L1 band signals 74 and the L1 center frequency carrier reference signal 114.

GPS signals 15 do not contain any useful amounts of signal power beyond approximately 9 megahertz from the center frequency of the band. Therefore, L1 band signals 74 be considered to have a bandwidth less than about 18 megahertz, centered approximately at the L1 center frequency carrier. The beat frequency component of the mixer output signal 122 is therefore the less than 18 megahertz wide component of the L1 band signal 74 which has been down-converted in frequency to baseband, that is, to a low frequency range from near zero to less than about 9 megahertz. This beat frequency component of the mixer output signal 122 is not rejected by the video low pass filter 124, and is included in the in-phase video signal 82.

The video low-pass filter 124 should be approximately matched to the spectral power density of the baseband of the n(t) or P-code related component of the GPS signals. This spectral density N(f) is approximately given by

N(f) - sinc² (f/2 fo)

where f is the video frequency, fo is the GPS fundamental frequency, and sinc² (f/2 fo) is the square of the "sinc" function, defined by

sinc (x) = sin (π*x)/(π*x)

where x, equal to f/2 fo, is the argument of the function, and π is equal to 3.1416 . . . N(f) has a one-sided bandwidth at half power of about 4.5 megahertz, and has a zero at f equal to 2 fo. The video low-pass filter 124 may be of conventional lumped ideal element construction, preferably with 3 to 5 poles. It is essential that the video low-pass filter 124 have a linear phase, and that the filter 130 be exactly phase matched to the video low-pass filter 124. to optimize the performance of the L1 band fo carrier reconstruction and downconversion device 86 which receives the video signals 82 and 84. Furthermore, it is desirable that the video low pass filters 124 and 130 have a high attenuation at the video frequency f equal to 2 fo so that the video signals 82 and 84 do not contain spurious components which interfere with the detection of the reconstructed carrier and the phase measurement in the L1 band fo carrier phase detector 206.

In addition to rejecting the double frequency mixing products in the mixer output signal 122, the video low pass filter 124 generally serves to limit the bandwidth of the in-phase video signal 82 to the bandwidth of the wanted signal. This limitation limits the amount of noise and potential interference passed to the subsequent stages of the receiver 24 and the satellite tracking channel 32.

The local oscillator input signal 114 is applied to the 90° phase delay circuit 118 to produce a delayed local oscillator input signal 120 which therefore lags the local oscillator input signal 114 by exactly one quarter cycle. The delayed local oscillator input signal 120 is applied as an input signal to the mixer 112 in the same manner as the undelayed local oscillator input signal 114 is applied to the mixer 110. The mixer output signal 128 from the mixer 112 is therefore substantially identical to the mixer output signal 122 from the mixer 110 except that the beat frequency components are shifted in phase by one quarter cycle.

Fig.5

In Fig. 5, the reconstruction device 86 of the receiver 24 of Fig. 3 is shown in more detail. As mentioned above with respect to Fig. 3, the reconstruction devices 86 and 96 are identical in construction and operation, except that the input signals supplied to them are different. The construction and operation of the reconstruction device 96 can therefore be understood from the discussion of the reconstruction device 86 below.

The downconverted in-phase video signal 82 and the downconverted quadrature video signal 84 are produced by the L1 band radio frequency to video frequency downconverter 76 shown in Fig. 3. As previously explained, the video signals 82 and 84 are identical except for a relative phase shift of exactly 90°. The direction of this phase shift, i.e. whether there is a leading phase shift or a lagging phase shift of VQ with respect to VI, depends on whether the L1 band input frequency was above or below the frequency of the local reference signal 114 shown in Fig. 4. The signal 114, as already mentioned with reference to Fig. 4, is nominally equal to a frequency of 308 fo, the center frequency carrier of the transmitted L1 band signals.

It is therefore possible to separate the upper sideband components of the L1 band signals 74 (the components of the signal 74 above the center carrier frequency of 1575.42 megahertz) from their lower sideband components (the components of the signal 74 which have a lower frequency than the center carrier frequency) by delaying the phase of the video signal 84 by an additional 90° with respect to the video signal 82. and subsequently adding and subtracting the results to form upper and lower sideband signals. The wideband video quadrature hybrid circuit 140, to which in-phase and quadrature video signals 82 and 84 are applied as inputs, performs this additional phase shifting and the addition and subtraction operations to form the upper sideband signal 142 and the lower sideband signal 144.

The wideband video quadrature hybrid circuit 140 may be conveniently constructed as described in detail in an article in the Proceedings of the IEEE, Vol. 59 (1971), pages 1617-1618, by Alan E. E. Rogers.

The upper sideband signal 142 at the output of the wideband video quadrature hybrid circuit 140 is equal to the arithmetic sum of the in-phase and quadrature video input signals 82 and 84, both input signals having been delayed in phase by amounts that are frequency dependent, but with the phase delay of the quadrature video input signal 84 being greater than that of the in-phase video input signal 82 by a constant 90 degrees regardless of frequency. The lower sideband signal 144 at the output of the wideband video quadrature hybrid circuit 140 is equal to the arithmetic difference of these two differentially phase shifted quadrature video input signals 82 and 84.

In operation of the reconstruction device 86, the upper sideband signal 142 at the output of the wideband video quadrature hybrid circuit 140 is filtered by the filter 146. The lower sideband signal 144 is filtered by the filter 148, which is identical to the filter 146.

The output signals of filters 146 and 148 are applied as input signals to double balanced mixer 150. The output signal of mixer 150 includes mixed product signals at both the sum and difference frequencies of the upper and lower sideband input signals 142 and 144. The output signal of mixer 150 is filtered by narrow band pass filter 152. The center frequency of narrow band pass filter 152 is 2 fo, or 10.23 megahertz. Therefore, narrow band pass filter 152 selects the sum frequency component of the output signal from 150.

The video input signals 82 and 84, as well as the upper and lower sideband signals 142 and 144, comprise a composite signal of all broadened spectrum signals received from all visible GPS satellites in the L1 frequency band. The components representing individual satellites in these composite signals are pseudo-random broadened spectrum signals whose spectra are noise-like and do not contain any discrete frequency or continuous wave components. However, after the upper sideband signal 142 is mixed with the lower sideband signal 144 in the mixer 150, reconstructed continuous wave carriers having discrete frequencies and phases are present in the second composite signal 154 at the output of the bandpass filter 152.

The filters 146 and 148 suppress signals that may interfere with the fo carrier reconstruction and phase measurement operations performed in subsequent sections of the L1 band fo carrier reconstruction apparatus and downconverter 86, as well as in the L1 band fo carrier phase detector 206. In particular, the filters 146 and 148 are designed with narrow band rejection characteristics. or "notch characteristics" to reject narrow bandwidth spurious signals that may be present in the upper and lower sideband signals 142. If not rejected by the filters 146 and 148, such spurious signals could be combined in the mixer 150 to produce spurious signals having frequencies that could not be rejected by the narrow band pass filter 152. It may be particularly useful to provide each of the filters 146 and 148 with a notch centered at a frequency fo, since a continuous wave component of the upper sideband signal 142 having a frequency of fo could mix with a corresponding component of the lower sideband signal 144 to produce a spurious signal having a frequency equal to 2 fo in the second composite signal 154.

It may also be desirable to provide high-pass filters 146 and 148 to reject upper and lower sideband signal components 142 and 144 having frequencies below about 1 megahertz. Such rejected components would contain the majority of the m(t) or C/A code related power of the GPS signals. In general, it is desirable for filters 146 and 148 to reject noise and other signals while passing the desired n(t) or P code related components of the GPS signals.

Regardless of the specific band-stop characteristics selected for the filters 146 and 148, it is essential that these filters be designed to produce the smallest possible phase shifts through the largest portion of the video frequency band of interest, which ranging from about 1 megahertz to 9 megahertz.

The second composite signal 154 is a composite signal of reconstructed carriers, each of which represents the fundamental frequency fo subcarrier from a GPS satellite. Each such reconstructed carrier in the second composite signal 154 has a frequency near 2 fo, the second harmonic of the fundamental frequency subcarrier, or about 10.23 megahertz. However, the frequencies of the reconstructed carriers from different satellites are generally not exactly the same. The frequency of the reconstructed carrier from any GPS satellite will appear in the second composite signal 154 to be different from 2 fo, primarily as a result of the Doppler shift due to the relative motion of that satellite and the receiver 24.

The maximum Doppler shift observed during ascent or descent of a satellite is on the order of plus or minus 30 hertz, respectively. Therefore, the narrow band pass filter 152 should have a bandwidth of at least 60 hertz in order to pass the reconstructed carrier signals from all satellites. It is undesirable for the bandwidth of the filter 152 to be too wide, since noise and interference far from the desired reconstructed subcarrier frequencies should be rejected. However, it is also undesirable for the bandwidth of the narrow band pass filter 152 to be too narrow, since a narrow bandwidth in a band pass filter is usually accompanied by a phase shift that varies sharply with frequency within the pass band. It is therefore preferable to use a bandwidth of several kilohertz in the narrow band pass filter 152 to maintain the phase linearity of the receiver 24.

The second composite signal 154 at the output of the narrow band pass filter 152 is applied to a pair of identical double balanced mixers 156 and 158. The local oscillator reference input signal 160 applied to the double balanced mixer 156 has a constant frequency of 2 fo which is derived in the frequency divider 162 from the 4 fo reference frequency signal 26 provided by the frequency standard 28 as shown in Fig. 2.

The output signal 164 of the double balanced mixer 156 includes mixing products at both the sum and difference frequencies of its input signals, the reconstructed carriers in the second composite signal 154, and the 2 fo local oscillator reference input signal 160. The sum frequencies are close to 4 fo. The desired difference frequencies are typically less than 30 hertz. The output signal 164 is applied as an input to a low pass filter 166 which has a bandwidth of about 1 kilohertz. The low pass filter 166 serves to suppress the sum frequencies and pass the difference frequencies to achieve composite signals 88 with in-phase reconstructed carrier, which are referred to for simplicity as SI(L1).

The local oscillator reference input signal 160 is also applied to the 90° phase delay circuit 172, and a delayed local oscillator reference input signal 160 is also applied to the 90° phase delay circuit 172 to obtain a delayed local oscillator which lags the local oscillator signal 160 by 90°. The delayed local oscillator signal 170 is applied as an input to the double balanced mixer 158. The output signal 174 of the double balanced mixer includes Mixing products at the sum and difference frequencies of its input signals, the reconstructed carriers in the second composite signal 174, and the delayed local oscillator signal 170. The sum frequencies are close to 4 fo. The difference frequencies are generally less than 30 hertz. The output signal 174 is applied as an input to the low pass filter 176, which has a bandwidth of about 1 kilohertz. The low pass filter 176 serves to reject the sum frequencies and pass the difference frequencies to obtain the composite quadrature phase reconstructed L1 band fo carrier signal 90, which is denoted for simplicity as SQ(L1).

As explained above with reference to Figure 3, the fo carrier reconstruction and down conversion circuit 96 operates in the same manner as the fo carrier reconstruction and down conversion circuit 86 just described above. The reconstructor 96 serves to generate in-phase reconstructed carrier signals 98, designated SI(L2), and composite quadrature phase reconstruction carrier signals 100, designated SQ(L2).

Therefore, referring again to Fig. 3, fo carriers, which are implicitly contained in the signals received in both the L1 and L2 bands, are reconstructed and frequency down-converted from near 10.23 megahertz to near zero. The purpose of the frequency down-conversion is to allow the subsequent signal processing circuits to operate at low speed.

Fig.6

In Fig. 6, the satellite tracking channel 32 shown in Fig. 2 is shown in more detail. As mentioned with reference to Fig. 2, one such tracking channel 32 is required per satellite tracked. Therefore, the receiver 24 typically includes six such tracking channels 32. All of these tracking channels 32 are identical and operate in the same manner. Therefore, the operation of only one of these tracking channels 32 will be described in detail with reference to Fig. 6. For ease of discussion, it is assumed that the tracking channel 32 shown in Fig. 6 has been assigned to the GPS satellite 12 by the satellite assignment 41 of the real-time computer 40.

All tracking channels 32 receive the same set of low frequency signals 30 from receiver 24 as shown in Fig. 2. Low frequency signals 30 include in-phase and quadrature video signals 82 and 84 from L1 radio frequency to video frequency downconverter 76, and also L1 band in-phase and quadrature reconstructed carrier composite signals 88 and 90 from reconstructor 86, and L2 band in-phase and quadrature reconstructed carrier composite signals 98 and 100 from reconstructor 96, all of which are shown in Fig. 3.

All tracking channels 32 also receive a 4 fo reference frequency signal 26 from the frequency standard 28 in Fig. 2 and a continuously updated real-time digital display 34 from the real-time clock 36 as shown in Fig. 2. In addition, each tracking channel 32 receives estimates 38, calculated by the real-time computer 40, of the frequency and C/A code delay of the GPS signals 15 received from the Antenna 22 can receive signals from the GPS satellite to be tracked by this channel.

The estimates 38 include a continuously updated estimate of the frequency of the 308 fo L1 band center frequency carrier received by the GPS satellite 12. The received frequency of any component of the GPS signals 15 differs from the frequency of the corresponding component of the signals transmitted by the GPS satellite 12 due to the Doppler shift resulting from the relative motion of the antenna 22 and the GPS satellite 12.

In addition, the frequency of the L1 band center frequency carrier implicitly included in the signals transmitted by the GPS satellite 12 may differ from exactly 1575.42 megahertz due to any variations, either accidental or intentional, in the satellite fundamental frequency fo used in the GPS satellite 12.

In addition to the frequency estimate of the 308 fo L1 band center frequency carrier, the estimates 38 include a group delay initial value 37 as shown in Fig. 13.

In-phase and quadrature video signals 82 and 84, the 4 fo reference frequency signal 26 and the real-time digital display 34 are applied as inputs to the 308 fo carrier phase and C/A code delay error detector 190, which also receives the 308 fo carrier phase estimate φest 192 and the C/A code group delay estimate τest 194 as inputs from the phase and group delay register subsystem 196. The 308 fo carrier phase estimate φest 192 is a fractional cycle portion of the 308 fo carrier phase estimate φest 856, as shown below. with reference to Fig. 13. The C/A code group delay estimate τest 194 is a fractional period portion of the C/A code group delay estimate τest 876, as also described below with reference to Fig. 13.

The 308 fo carrier phase and C/A code delay error detector 190 serves to generate a phase error signal 198 and a group delay error signal 200, as well as an above-threshold indication 410, a concurrent signal energy indication 348, and a sign bit 201, as will be explained below with reference to Figures 7 and 10.

The 308-fo implicit carrier phase observable φ is implicitly included in the GPS signals 15 received from the GPS satellite 12 and is therefore implicitly included in the complex video signal V, i.e., in-phase and quadrature video signals 82 and 84. Accordingly, the actual group delay occurring between the transmission by the GPS satellite 12 and the reception by the antenna 22 of the C/A component implicitly included in the L1 band portion of the GPS signals 15, namely, the C/A code group delay observable τ, is implicitly included in the complex video signal V.

The phase error signal 198 is an estimate of the difference between the 308 fo implicit carrier phase observable φ and the 308 fo carrier phase estimate φest 192. The group delay error signal 200 represents an estimate of the difference between the C/A code group delay observable τ and the C/A code group delay estimate τest 194.

The 308-fo implicit carrier phase observable φ and the C/A code group delay observable τ are tracked by the phase and group delay register subsystem 196 as a result of interaction with the 308-fo carrier phase and C/A code delay error detector 190. The phase and group delay register subsystem 196 and the 308-fo carrier phase and C/A code delay error detector 190 cooperate to form feedback loops to minimize the phase error signal 198 and the group delay error signal 200. The cooperation of the 308fo carrier phase and C/A code delay error detector 190 and the phase and group delay register subsystem 196 therefore serves to form interrelated and interconnected phase and delay feedback loops which track the phase and group delay of signals implicitly included in the GPS signals 15, respectively.

As described in more detail below with reference to Fig. 7, the carrier phase and C/A code delay register subsystem 196 also generates a fo carrier phase estimate 204 which represents an estimate of the phase of the satellite fundamental frequency fo carrier implicitly contained in the 1575.42 megahertz L1 band signals received from the GPS satellite 12. The fo carrier phase estimate 204 is provided to both the fo carrier phase detector 206 for the L1 band and the fo carrier phase detector 208 for the L2 band.

The L1-fo carrier phase detector 206 receives the composite signal 88 of the in-phase reconstructed L1-band fo carrier and the composite signal 90 of the quadrature-phase reconstructed L1-band fo carrier as input signals from the reconstruction device 86. The L2-fo carrier phase detector 208 receives the composite signals 98 and 100 for the in-phase and quadrature reconstructed carrier, respectively, from the reconstructor 96. The L1 band fo carrier phase estimate 204 is provided to the L1 fo carrier phase detector 206, which performs an L1 fo residual phase measurement 210 equal to the difference between the fo carrier phase estimate 204 and the observable ψL1 of the fo carrier phase implicitly contained in the L1 band portion of the GPS signals 15 received by the antenna 22. The fo carrier phase estimate 204 is also fed to the L2 fo carrier phase detector 208 which makes an L2 fo residual phase measurement 212 equal to the difference between the fo carrier phase estimate 204 and the observable ψL2 for the fo carrier phase implicitly contained in the L2 band portion of the received GPS signals 15. The L1 fo residual phase measurement 210 and the L2 fo residual phase measurement 212 are included in the measurements 42 fed to the real-time computer 40.

Fig.7

In Fig. 7, the 308 fo carrier phase and C/A code delay error detector 190 in the satellite tracking channel 32 of Fig. 6 is shown in more detail. As mentioned above with respect to Fig. 6, the 308 fo carrier phase and C/A code delay error detector 190 cooperates with the phase and group delay register subsystem 196 within the satellite tracking channel 32 to form phase and delay feedback loops. The 308 fo carrier phase and C/A code delay error detector 190 receives the in-phase video signal 82 and the Quadrature video signal 84 as inputs from the L1 downconverter 76 and, using the reference frequency input signal 26 and the real-time digital display 34, serves to generate the phase error signal 198 and the group delay error signal 200 by comparing the current observables ? implicitly contained in the 308 fo carrier phase. and the current C/A code group delay observable τ implicitly contained in the complex video signal V with the 308 fo carrier phase estimate φest 192 and the C/A code group delay estimate τ est 194 from the phase and group delay register subsystem 196. The 308 fo carrier phase and C/A code delay error detector 190 includes a phasor counter rotator 220, a correlator 226, and a C/A code generator 230. The phasor counter rotator 220, the correlator 226, and the C/A code generator 230 are shown in more detail below with reference to FIGS. 9, 10, and 11, respectively.

The in-phase video signal 280 and the quadrature video signal 84 from the L1 downconverter 76 shown in Figure 3 are applied as inputs to the phasor counter rotator 220 which also receives the 308 fo carrier phase estimate φest 192. The phasor counter rotator 220 subtracts the 308 fo carrier phase estimate φest 192 from the phase angle of the complex video signal V. As a result of this subtraction, the rate of change of the center frequency carrier phase implicitly contained in the complex video signal V is reduced by an amount equal to the rate of change of the 308 fo carrier phase estimate φest 192.

The phase counter rotator 220, as described in more detail with reference to Fig. 9 below, produces an in-phase developed video signal 222, UI, and a Quadrature expanded video signal 224 UQ which together form the expanded complex video signal U. The term "expanded" in the expression "expanded complex video signal U" refers to reducing the speed of rotation of the pointer in the complex plane which can be used to represent the expanded complex video signal U. The expanded complex video signal U is "expanded" in the sense that it is substantially equal to the complex video signal V, except that the speed of rotation in the complex plane of the pointer of the expanded complex video signal U is substantially less than the speed of the pointer in the complex plane which can be used to represent the complex video signal V. In other words, the constantly changing phase of the developed complex video signal U changes at a significantly slower rate than the constantly changing phase of the complex video signal V. The representation of the complex video signal V and the developed complex video signal U in the complex plane and the operation of the pointer counter rotator 220 can be understood by referring to Fig. 8.

Fig.8

Referring to Fig. 8, the operation of the pointer counter rotator 220 can be understood in the following manner. As mentioned above, the in-phase video signal 82, VI, and the quadrature video signal 84, VQ, can be considered as the real and imaginary parts of the complex video signal V, respectively; that is:

V = VI + i VQ

where i is the square root of -1. Accordingly, the in-phase developed video signal 222, UI, and the quadrature developed video signal 224, UQ, the real and imaginary parts of the developed complex video signal U; thus:

U = UI + i UQ.

The pointer counter rotator 220 generates the developed complex video signal U according to the following equation:

U = V exp (-i Φest),

where "exp ( )" is the following complex exponential function

exp (i Φest) = cos (Φest) + i sin (Φest).

Therefore, the phase angle of the developed complex video signal U is equal to the phase angle of the complex video signal V minus the 308 fo carrier phase estimate est 192.

Fig. 8A is a representation of the complex video signal V in the complex plane. The in-phase or real component of the complex video signal V is plotted on the horizontal axis 85, and the quadrature or imaginary component of the complex video signal V is plotted on the vertical axis 87. The in-phase video signal 82 is equal to the projection on the horizontal axis 85, and the quadrature video signal 84 is equal to the projection on the quadrature vertical axis 87, of the video pointer 83. The pointer 83, whose angle is shown as an angle 81, R, represents the complex video signal V.

The upper right quadrant of the complex plane in Fig. 8A is the quadrant in which the signs of both the in-phase video signal 82 and the quadrature video signal 84 are positive. This quadrant is referred to as the first quadrant by common convention. Similarly, the upper left quadrant is the quadrant in which the sign of the in-phase video signal 82 is negative and the sign of the quadrature video signal 84 is positive. This quadrant is referred to as the second quadrant. The lower left quadrant is the quadrant in which the signs of both the in-phase video signal 82 and the quadrature video signal 84 are negative. This quadrant is referred to as the third quadrant. The lower right quadrant is the quadrant in which the sign of the in-phase video signal 82 is positive and the sign of the quadrature video signal 84 is negative. This quadrant is referred to as the fourth quadrant.

Fig. 8B is a complex plane representation of the 308 fo carrier phase estimate phasor 193 whose angle is equal to the 308 fo carrier phase estimate Φest 192. Similarly, Fig. 8C is a complex plane representation of the developed complex video phasor 223 whose angle is equal to the difference between the angle θ1, R, and the 308 fo carrier phase estimate Φest 192.

In Fig. 8D, the complex plane of the reconstructed L1 band fo carrier composite signal S(L1) is shown with the real part SI(L1) and the imaginary part SQ(L1) plotted on the horizontal and vertical coordinate axes, respectively. The phasor 89 is the vector sum or resultant of the phasors 8912, 8913 and 8914, which represent the components of S(L1) represented by the composite phasor 89 of the L1 band reconstructed carrier, due to GPS satellites 12, 13 and 14, respectively.

Fig. 8E is a snapshot viewed from the antenna 22 at a time indicative of the relationship between the satellite clock indication t1 601, the delay error signal 200, the 30-bit C/A code group delay estimate τest 876, and the real-time digital indication 34 from the real-time clock 36.

The satellite clock indication t1 601 is the time indicated by the signal received from the GPS satellite 12 for a particular real-time digital indication 34 from the real-time clock 36. The satellite clock indication t1 601 is implicitly contained in the modulation of the GPS signals 15 received from the GPS satellite 12 and can be read explicitly by decoding the transmitted navigation message contained in this signal.

The 30-bit C/A code group delay estimate τest 876 of Figure 13 is generated by or contained in the delay register 842 and represents an estimate of the C/A code group delay observable τ.

The difference between the real-time digital indication 34 and the sum of the satellite clock indication t1 601 and the 30-bit C/A code group delay estimate τest 876 is the delay error, the sign of which is indicated by the delay error signal 200.

Referring again to Fig. 7, the in-phase video signal 82 and the quadrature video signal 84 from the pointer counter rotator 220 are received in analog form by the L1 downconverter 76 and converted to a one-bit digital form in which the value of each bit indicates the sign of the corresponding video signal. The two bits which constitute the in-phase video signal 82 and the quadrature video signal 84, respectively, therefore together indicate the quadrant in the complex plane of the video pointer 83.

The 308-fo carrier phase estimate Φest 192 is received by the phasor counter rotator 220 from the phase and group delay register subsystem 196 in two-bit digital form, which indicates the quadrant of the 308-fo carrier phase estimate phasor 193. The phasor counter rotator 220 produces the in-phase developed video signal 222 and the quadrature developed video signal 224 in one-bit digital form, and together this represents the quadrant of the developed complex video phasor 223 in the complex plane.

If the 308 fo carrier phase estimate Φest 192 indicates that Φ is close to zero, modulo 1 cycle, then the 308 fo carrier phase estimate phasor 193 lies in the fourth quadrant, and the in-phase developed video signal 222 and the quadrature developed video signal 224 are equal to the in-phase video signal 82 and the quadrature video signal 84, respectively, and therefore the developed complex video phasor 223 lies in the same quadrant as the video phasor 83. If the 308 fo carrier phase estimate Φest 192 indicates that Φ is close to zero, modulo 1 cycle, then the 308 fo carrier phase estimate phasor 193 lies in the fourth quadrant, and the in-phase developed video signal 222 and the quadrature developed video signal 224 are equal to the in-phase video signal 82 and the quadrature video signal 84, respectively, and therefore the developed complex video phasor 223 lies in the same quadrant as the video phasor 83. is close to 90º or plus a quarter cycle, then the developed complex video pointer 223 is one quadrant more clockwise than the video pointer 83. If the 308-fo carrier phase estimate 192 indicates that Φ is close to 180º, then a 180º rotation is performed, and so on.

The in-phase developed video signal 222 and the quadrature developed video signal 224 may therefore be formed within the phasor counter rotator 220 by logic gates (not shown) operating according to a simple truth table which is described in more detail with respect to Fig. 9.

Within the 308 fo carrier phase and C/A code delay error detector 190, the in-phase developed video signal 222 and the quadrature developed video signal 224 are applied as inputs to the correlator 226, as shown in Fig. 7. The correlator 226 also receives an 8 fo reference signal 27 from the frequency doubler 25, which produces the 8 fo reference signal 27 by multiplying the 4 fo reference signal 26 from the frequency standard 28 by a factor of 2, and receives a real-time digital display 34 from the real-time clock 36, as shown in Fig. 2. Furthermore, the correlator 226 receives a 1-bit C/A code local model 228 from the C/A code generator 230.

The correlator 226 correlates the in-phase developed video signal 222 and the quadrature developed video signal 224 with the 1-bit C/A code local model 228. The 1-bit C/A code local model 228 is a sequence of bit values representing the C/A code specific to the GPS satellite 12. This sequence is output by the C/A code generator 230 at times corresponding to the estimated time of arrival of the C/A sequence from the GPS satellite to which the C/A code generator 230 has been assigned by the satellite assignment 41 from the real-time computer 40. The timing of the generation of the 1-bit C/A code local model 228 is determined by the real-time digital display 34 provided by the real-time clock 36, by the 8 fo reference signal 27 from the frequency doubler 25, and by the C/A code group delay estimate τest 194 shown in Fig. 6.

The C/A code group delay estimate τest 194 is a multi-bit digital representation of the estimated delay, modulo the code period of 1 millisecond, where the least significant bit represents 1/40th of a bit or piece of the C/A code sequence. As explained above the C/A code is a periodic sequence of 1023 equal-width pieces transmitted with a period of exactly 1 millisecond. The least significant bit of the C/A code group delay estimate τest 194 therefore corresponds to a path length of about 7.5 meters. The RMS quantization error of the C/A code group delay estimate τest 194 is therefore of the order of 7.5 meters divided by the square root of 12, or about 2 meters. This 2-meter RMS quantization error is negligible compared to the other errors affecting the received C/A code, such as multipath errors.

The 1-bit C/A code local model 228 correlates constructively only with the specific component of the developed complex video signal U that comes from the GPS satellite 12. The C/A code used in any other satellite is uncorrelated, i.e. orthogonal to the code of the GPS satellite 12. As a result of this orthogonality, the correlator 226 serves to select the signals from the GPS satellite 12 and to suppress the signals from other satellites.

As mentioned above with respect to the structure of the GPS signals, the modulation related to the C/A code is a biphasic modulation which switches the phase of a component of the 308 fo L1 center frequency carrier transmitted by the GPS satellite 12 between two values 180° apart, according to whether the current code bit is 1 or 0, respectively. These phase reversals occur pseudo-randomly at integer multiples of the C/A code bit width, tm. Therefore, the developed complex video signal U appears to fluctuate in phase by 180° at this rate.

The correlation within the correlator 226 by the 1-bit C/A code local model 228, when the 1-bit C/A code local model 228 is generated synchronously with the received C/A code and matches the sequence of the C/A code specific to the GPS satellite 12, maintains this variation so that the phase remains relatively constant over a time interval sufficient to allow coherent integration and detection of this signal. However, the phase does not remain constant over a very long period of time since every 20 milliseconds the phase may be reversed by the telemetry modulation transmitting the navigation message.

The correlator 226 correlates the in-phase developed video signal 222 and the quadrature developed video signal 224 with the 1-bit C/A code local model 228 for a time interval that is less than 20 milliseconds. If this time interval does not include telemetry inversion, the phase remains relatively constant so that the correlator 226 can generate a phase error signal 198 representing the difference between the implicitly included 308 fo carrier phase observable φ and the 308 fo carrier phase estimate φest 192. The correlator 226 also sets the sign of the difference between the C/A code group delay observable τ and the 308 fo carrier phase estimate φest 192. and the C/A code group delay estimate τest 194.

As explained in more detail below with reference to Figure 10, the correlator 226 determines the sign of this delay difference by comparing the energies of the correlation products of two correlations. The first such correlation is the correlation of the 1-bit C/A code local model 228 with the in-phase developed video signal 222 and the quadrature developed video signal 224 as explained above. The second such Correlation is the correlation of the in-phase developed video signal 222 and the quadrature developed video signal 224 with a further delayed version of the 1-bit C/A code local model 228 where the further delay is equal to one slice width of the C/A code. The energies resulting from each of these two correlations are compared to determine whether correlation with the signal representing the early code results in more or less energy than correlation with the signal representing the later or delayed code. Delay error signal 200 indicates the result of this comparison and is applied to phase and group delay register subsystem 196 where this indication of code delay is used to advance or delay the delay register shown in Fig. 13 to establish a fixed delay relationship between the current value of the C/A code group delay observable τ and the current value of the C/A code group delay observable τ. and the C/A code group delay estimate τest 194.

Fig.9

Referring now to Fig. 9, the phasor counter rotator 220 is described in more detail. As mentioned above with reference to Fig. 7, the phasor counter rotator 220 operates to subtract the 308 fo carrier phase estimate φest 192 from the angle θ1 R of the complex video phasor 83 to produce the developed video signal 222 and the quadrature developed video signal 224, which together form a 2-bit digital representation of the quadrant in the complex plane of the developed video phasor 223.

The function performed by the pointer counter rotator 220 in the 308 fo carrier phase and C/A code delay error detector 190 is also required elsewhere in the system according to the present invention. Corresponding functions are performed within each satellite tracking channel 32 by the phasor counter rotator 240 in the L1-fo carrier phase detector 206 shown in Fig. 12 and by a corresponding phasor counter rotator, not shown, in the L2-fo carrier phase detector 208. The operation of the phasor counter rotator 220 will be described in detail with reference to Fig. 9 and the operations of the phasor counter rotators in the L1-fo carrier phase detector 206 and in the L2-fo carrier phase detector 208 can be understood therefrom.

As shown in Figure 9A, the in-phase video signal 82 and the quadrature video signal 84 at the input of the pointer counter rotator 220 are applied to a pair of identical 1-bit analog-to-digital converters 802 and 804, respectively, which operate as clippers to produce a 1-bit in-phase video sign indicator 806 and a 1-bit quadrature video sign indicator 808, respectively. The 1-bit in-phase video sign indicator 806 is a logical or binary digital signal in which a true or "1" state represents that the sign of the in-phase video signal 82 was positive, and a false or "0" state represents that its sign was negative. The same relationship applies between the states of the 1-bit quadrature video sign indicator 808 and the quadrature video signal 84.

The 1-bit in-phase video sign indicator 806 and the 1-bit quadrature sign indicator 808 together form a 2-bit digital representation of the quadrant of the video pointer 83, as can be seen from the first three columns of the table in Fig. 9B. In particular, it can be seen from a comparison between Figs. 8A and 9B that when the Video pointer 83 is in the first quadrant, the signs of in-phase video signal 82 and quadrature video signal 84 are positive, so that the digital states of 1-bit in-phase video sign indicator 806 and 1-bit quadrature video sign indicator 808 must be true or 1. Similarly, when video pointer 83 is in the second quadrant, 1-bit in-phase video sign indicator 806 and 1-bit quadrature video sign indicator 808 are 0 and 1, respectively. In the third quadrant, they are both 0, and in the fourth quadrant, they are 1 and 0, respectively.

This 2-bit digital representation of the phase angle of the video pointer 83 is modulo one cycle, so therefore the integer number of cycles is ignored in this representation. For example, an instantaneous phase angle between a cycle 0 and a cycle 1 cannot be distinguished from a phase angle between 2 and 3 cycles.

Referring again to FIG. 9A, the 308 fo carrier phase estimate φest 192 from the phase and group delay register subsystem 196 is applied as an input to the phasor counter rotator 220 at the input to the logic circuit 810. More particularly, as will be seen in greater detail below in FIG. 13, the phase register 840 contains a multi-bit 308 fo carrier phase estimate φest 856, only two bits of which are included in the 308 fo carrier phase estimate φest 192. The first bit to the right of the binary point of the 308 fo carrier phase estimate φest 856 is the half-cycle bit R1. 812, which indicates whether the 308-fo carrier phase estimate pointer 193 lies in the first two or the second two quadrants of the complex plane. The second bit to the right of the binary point is the quarter cycle bit R₂ 814, which indicates whether the number of the quadrant of the 308-fo carrier phase estimate pointer 193 is odd or even.

The 308 fo carrier phase estimate φest 192 can be understood by referring to the two rightmost columns in the table in Fig. 9B. When the 308 fo carrier phase estimate pointer 193 is in the first quadrant, the half-cycle bit R1 812 and the quarter-cycle bit R2 814 are both equal to 0, indicating that the pointer has neither a half-cycle nor a quarter-cycle of phase. When the 308 fo carrier phase estimate pointer 193 is in the second quadrant, the half-cycle bit R1 812 and the quarter-cycle bit R2 814 are both equal to 0, respectively. 814 are equal to 0 and 1, respectively. Similarly, the third quadrant is displayed when the half-cycle bit R1 812 and the quarter-cycle bit R2 814 are equal to 1 and 0, and the fourth quadrant is displayed when they are both equal to 1. The difference between this binary form of quadrant display and the sign-based display provided by the 1-bit in-phase video sign indicator 806 and the 1-bit quadrature video sign indicator 808 can be seen by referring to the table in Fig. 9B.

Referring now again to Fig. 9A, logic circuit 810 processes the complex video signal V represented by the 1-bit in-phase video sign indicator 806 and the 1-bit quadrature video quadrant indicator 808 to subtract therefrom the 308-fo carrier phase estimate φest 192 to produce the developed complex video signal U represented by the in-phase developed video signal 222 and the quadrature developed video signal 224. In this operation, phase angles are described only in terms of quadrants. This description is relatively crude in that it can only be used within plus or minus one-eighth cycle. The in-phase developed video signal 222 and the quadrature developed video signal 224 provide a 2-bit digital representation of the quadrant of the developed complex video pointer 223.

The truth table in Fig. 9C shows the desired functional relationship between the in-phase developed video signal 222 output from the logic circuit 810 and the four input signals, namely, the in-phase video sign indicator 806, the quadrature video sign indicator 808, the half-cycle bit R1 812 and the quarter-cycle bit R2 814. Similarly, the truth table shown in Fig. 9D shows the required functional relationship between the quadrature developed video signal 224 output from the logic circuit 810 and these input signals. With these relationships clearly defined according to these truth tables, it is reasonable to provide the logic circuit 810 in the form of logic gates or a read-only memory, not shown.

Fig.10

Referring now to FIG. 10, the operation of the correlator 226 within the 308-fo carrier phase and C/A code delay error detector 190 shown in FIG. 7 is shown in more detail. The in-phase developed video signal 222 from the phasor counter rotator 220 is applied as one input to mixers 300, 302 and 304. The quadrature developed video signal 224 from the phasor counter rotator 220 in FIG. 7 is applied as an input to mixer 306, mixer 308 and mixer 310. The second inputs to mixers 300, 302, 304, 306, 308 and 310 are provided by the 1-bit C/A code local model 228 and its half-delayed and one-delayed versions, 314 and 318, respectively. The in-phase developed video signal 222, the quadrature developed video signal 224, and the 1-bit C/A code local models 228, 314, and 318 all represent one-bit digital or logic signals. Therefore, the mixers 300, 302, 304, 306, 308, and 310 are exclusive NOR logic gates; equivalently, these mixers can be viewed as binary adders modulo 2 or as parity detectors.

The 1-bit C/A code local model 228 is applied as an input to the half-bit delay circuit 312, which produces a half-bit delayed C/A code local model 314. The half-bit delayed C/A code local model 314 is then applied as an input to the half-bit delay circuit 316, which produces the one-bit delayed C/A code local model 318 by delaying the half-bit delayed C/A code local model 314 by an additional half-bit. The half-piece delay circuit 312 and the half-piece delay circuit 316 operate in an identical manner, and each applies a delay equal to exactly one-half the width, tm, of a C/A code piece or bit.

The piece rate of the C/A code in the GPS system is 1023 pieces per millisecond or 1023 million pieces per second or 1023 megahertz or fo / 5. Each piece is therefore equal to 5 oscillation cycles of the reference frequency fo or equal to 20 cycles of the 4 fo reference frequency 26 from the frequency standard 28 shown in Fig. 2, or equal to 40 cycles of the 8 fo reference signal 27. 8 fo is the "clock" rate at which the 1-bit C/A code local model 228 is generated by the C/A code generator 230, as described below with reference to on Fig. 11. The correlator 226 receives the 8 fo reference signal 27 from the frequency doubler 25 shown in Fig. 7 and the real-time digital display 34 from the real-time clock 36 shown in Fig. 2.

The half-delay circuits 312 and 316 may therefore each be formed by a 20-bit shift register clocked by the 8 fo reference frequency 27 from the frequency doubler 25 shown in Figure 7. The 8 fo reference frequency 27 also serves to clock the 0.01 second integrator 324 as explained below. The half-delayed C/A code local model 314, which is a half-delayed version of the 1-bit C/A code local model 228, is applied as an input to the mixers 302 and 308 at the input of the concurrent signal detector 350 and also as an input to the half-delayed circuit 316. The one-bit delayed C/A code local model 318 produced by the half-bit delay circuit 316, which is a one-bit delayed version of the 1-bit C/A code local model 228, is applied as an input to the mixers 300 and 306 at the input of the delay detector 358. The 1-bit C/A code local model 228 is applied directly, without any delay, to the mixers 304 and 310 at the input of the leading signal detector 352.

The operation of the contemporaneous signal detector 350 will be discussed first. The operation of the early signal detector 352 and the late signal detector 358 is similar to the operation of the contemporaneous signal detector 350 and can then be more easily understood.

Within the simultaneous signal detector 350, the mixer 302 generates an output signal 322 which is mentioned above is the exclusive NOR function of the in-phase developed video signal 222 and the one-half delayed C/A code local model 314. The output signal 322 is applied as an input to the 0.01 second integrator 324 where it is integrated to produce the output value 330.

Since the output signal 322 is a 1-bit digital signal, the 0.01 second integrator 324 is most easily implemented by a clocked counter operated in response to the real-time digital display 34 from the real-time clock 36 shown in Fig. 2 to start, stop and restart integration at every integer multiple of 0.01 seconds as indicated by the real-time clock 36. To begin integration, the integrator 324 must be reset upon the occurrence of the real-time digital display 34 equal to an integer multiple of 0.01 seconds. Then, for each cycle of the 8 fo reference signal 27, the count value representing the integrated value is incremented or not depending on whether the output signal 322 is equal to 1 or 0. At the end of the integration period of 0.01 seconds, the value contained in the 0.01 second integrator 324 is read to obtain the output signal 330, and then the 0.01 second integrator 324 is reset to begin the next integration.

When the 0.01 second integrator 324 is thus formed from a clocked counter, its output signal 330 is offset by half the number of cycles of the 8 fo reference frequency 27 contained in an integration interval of 0.01 seconds, or by 4 fo times 0.01 seconds, or by 204,600 counts. This offset is usefully used in the arrangements of the 2-quadrant Inverse tangent function generator 334 and squarer 338, which accept the output signal 330 from the 0.01 second integrator 324.

Accordingly, mixer 308 produces an output signal 326 which is integrated by integrator 328 over 0.01 seconds, beginning and ending at integer 1/100ths of a second, under the control of real time clock 36, to produce output signal 332. At the end of the 0.01 second integration period, output signal 330 from 0.01 second integrator 324 and output signal 332 from integrator 328 are applied as inputs to 2-quadrant inverse tangent function generator 334. The 2-quadrant inverse tangent function generator 334, which may be in read-only memory (ROM), produces a phase error signal 197 equal to the angle in the range minus one-quarter cycle to plus one-quarter cycle whose tangent is equal to Q/I, where Q represents the output signal 332 and I represents the output signal 330. It should be noted that the phase error signal 197 from the 2-quadrant inverse tangent function generator 334 is not affected by telemetry phase reversals in the GPS signals 15 since such reversals act to change the signs of both I and Q, provided that no reversal occurs during the particular 0.01 second interval for which I and Q were generated. The phase error signal 197 from the 2-quadrant inverse tangent generator 334 therefore indicates the angle modulo 0.5 cycles of the average value during the 0.01 second integration period of the complex number whose real part is the output signal 322 and whose imaginary part is the output signal 326. As explained above, the angle of this complex number remains relatively constant during the integration period, except for the Telemetry phase reversals that occur at integer multiples of 0.02 seconds.

It is essential that the integration interval, namely 0.01 seconds, be an integer multiple of milliseconds, and also that it be exactly half the interval between possible telemetry modulation phase reversals. The integration should therefore be carried out over an integer multiple of milliseconds, since a millisecond is the period of the C/A code function. As explained above, the satellite-specific C/A codes are orthogonal to each other, so that the cross-correlation of the C/A code components of the signals of one satellite with the C/A code of another satellite results in zero. However, this desired lack of cross-correlation cannot be achieved exactly unless the integration extends over an integer multiple of the code periods. In the present case, the integer number is 10.

The fact that the integration period is exactly half the telemetry phase reversal interval guarantees that no phase reversals occur in any even-numbered integration intervals, or in any odd-numbered integration periods, over a long period of time.

The angle of the complex signal component from the GPS satellite 12 at the input to the integrators 324 and 328 remains constant during the integration period, and therefore this signal accumulates coherently within the integrators 324 and 328 over all odd or even intervals that do not contain phase reversals. The phase reversals that occur during half of the other integration intervals, even or odd, respectively, reduce the accumulated average value for these intervals. By comparing the average of the signal energy accumulated during even integration intervals with the average of the signal energy accumulated during odd integration periods, it is possible to determine the phase of the 20 millisecond telemetry reversals with respect to the phase of the 20 millisecond period signal defined by the beginning and end of the even or odd integration periods. Conveniently, these averages are calculated and compared within the computer 40, which is fed with the simultaneous signal energy display 348 for each integration interval, as explained below.

The sign indicator 199 determines the sign of the output signal 330 of the 0.01 second integrator 324 and generates the sign bit 201 which indicates whether the output signal 330 was positive or negative. The sign bit 201 is provided to the computer 40. The sequence of the sign bit 201 values from the even or odd 0.01 second integration intervals which are found not to contain any telemetry phase reversals are then interpreted by the computer 40 as the telemetry message. For this interpretation to be possible, phase locking must have been achieved. If phase locking has been achieved, then substantially all of the simultaneous signal energy appears in the output signal 330 and none appears in the output signal 332.

The output signal 330 from the integrator 324 for 0.01 seconds is applied at the end of each integration period of 0.01 seconds as an input signal to the squarer 338, which produces an output signal 342 that represents the square of the output signal 330. The output signal 332 of the The energy of integrator 328 is applied at the end of each 0.01 second integration period to squarer 340 which produces output signal 344 which is equal to the square of output signal 332. Output signals 342 and 344 are applied as inputs to summer 346 to obtain instantaneous signal energy indication 348 for each 0.01 second integration period. The simultaneous signal energy indication 348 is equal to the sum of the squares of the output signals 330 and 332. Conveniently, the phase error signal 197, the sign bit 201 and the simultaneous signal energy indication 348 are obtained directly from the output signal 330 and the output signal 332 by using the values of the output signal 330 and the output signal 332 to address a ROM in which pre-calculated values of the phase error signal 197, the sign bit 201 and the simultaneous signal energy indication 348 have been stored.

The simultaneous signal energy indication 348 is applied as an input to the real-time computer 40 shown in Fig. 2 at the end of each 0.01 second integration interval. In the real-time computer 40, the average of the even interval energies of the simultaneous signal energy indication 348 is compared to the average of its odd interval energies to determine the phase of the telemetry reversals. The phase error signal 197 from the 2-quadrant inverse tangent function generator 334 is clocked through the gate 413 to obtain the phase error signal 198. The phase error signal 198 is output from the gate 413 if and only if the indication 410 indicates for a value above the threshold also input to the gate 413 that the phase error signal 197 is a valid determination of the phase error. As explained below, the display 410 shows for a value above the threshold indicates that the simultaneous signal energy indication 348 is above a predetermined threshold and, therefore, sufficient signal energy has been accumulated during the 0.01 second integration interval to ensure that a valid phase determination could be made by the 2-quadrant inverse tangent function generator 334.

The phase error signal 198, which represents a valid measurement, is applied as an input to the phase and group delay register subsystem 196 shown in Figure 6 to maintain phase lock.

If the 308 fo carrier phase estimate Φest 192 is locked to and therefore equal to the 308 fo implicit carrier phase Φest, then the phase error signal 198 is approximately equal to zero.

As mentioned, the telemetry can be read by the real-time computer 40 by scanning the sequence of even or odd sign bits 201, looking for the frame synchronization pattern transmitted by the GPS satellite 12 to mark the beginning of each telemetry frame. The parity of this synchronization pattern indicates whether the sign bit 201 must be inverted to obtain the telemetry message bits.

The telemetry bit values should be determined from either the even or odd 0.01 second integration times, whichever is found to be free of phase errors, as explained above. Although the phase error signal 198 is better derived from only the odd or only the even integration times, whichever is found not to contain telemetry phase reversals, it is not strictly necessary to distinguish between even and odd integration intervals for this purpose. One of the two values for the phase error signal 198, for the odd interval or for the even interval, is always valid since there are no reversals within that integration period. The other, even or odd, may also be valid unless the telemetry reversals occur very near the midpoint of the integration interval. In the case where the phase reversals occur near the midpoint of the integration period, the very low output value of the output signal 330 and the output signal 332 would result in an unreliable value 197 at the output of the 2-quadrant inverse tangent function generator 334, which would be better not being passed as the phase error signal 198.

To detect such invalid indications and prevent their acceptance for use as the phase error signal 198, the energy of the signal received in each integration interval, as indicated by the simultaneous signal energy indication 348 from summer 346, is compared to the threshold 406 in comparator 408. Comparator 408 produces a binary-valued over-threshold indication 410 indicating whether or not the simultaneous signal energy indication 348 was greater than the threshold 406. Threshold 406 should be selected high enough so that when the simultaneous signal energy indication 348 exceeds threshold 406, the over-threshold indication 410 indicates that the phase error signal 198 is valid. The over-threshold indication 410 from comparator 408 is applied as an input to gate 413 as mentioned above to indicate the presence of a valid measurement. The above-threshold indication 410 is also used by real-time computer 40 during the initial signal acquisition process described below with additional reference to FIG. 6.

Within the delayed signal detector 358, the mixer 300 produces an output signal 323 which can be viewed as the exclusive NOR function of the in-phase developed video signal 222 and the one-bit delayed C/A code local model 318. The output signal 323 is applied as an input to the 0.01 second integrator 325 where it is integrated to form the output signal 331.

Since the output signal 323 is also a one-bit digital signal, the 0.01 second integrator 325 may be formed by a clocked counter operated in response to the real-time digital display 34 from the real-time clock 36 shown in Figure 2 to start and stop integration at every integer multiple of 0.01 seconds indicated by the real-time clock 36. The integrator 325 must be operated in the same manner as the 0.01 second integrator 324 with respect to the simultaneous signal detector 350, as explained above.

The mixer 306 produces an output signal 327 which is integrated for 0.01 seconds by the integrator 329, beginning and ending at integers of 0.01 times a second, under the control of the real-time clock 36 to produce the output signal 333.

The output signal 331 from the 0.01 second integrator 325 is applied as an input to the squarer 339 at the end of the 0.01 second integration period, which produces an output signal 343 equal to the square of the output signal 331. The output signal 333 from the integrator 329 is applied as an input to the squarer 341 at the end of the 0.01 second integration period, which produces the output signal 345 equal to the square of the output signal 333. The output signal 343 and the output signal 345 are applied as inputs to the summer 347, which produces the delayed signal 356, which is therefore a measure of the energy of the signal received from the GPS satellite 12 during the integration period.

Within the leading signal detector 352, the mixer 304 produces an output signal 369 which can be viewed as the exclusive NOR function of the in-phase developed video signal 222 and the 1-bit C/A code local model 228. The output signal 369 is applied as an input to the 0.01 second integrator 349 where it is integrated to form the output signal 351.

Since the output signal 369 is also a one-bit digital signal, the 0.01 second integrator 349 may be formed by a clocked counter operated in response to the real-time digital indication 34 from the real-time clock 36 shown in Fig. 2 to start and stop integration at every integer multiple of 0.01 seconds as indicated by the real-time clock 36. The integrator 349 must be operated in the same manner as the 0.01 second integrator 324, as mentioned above, with respect to the contemporaneous signal detector 350.

Accordingly, the mixer 310 generates the output signal 357.

which is integrated for 0.01 seconds by the integrator 359, beginning and ending at integers of 0.01 times a second under the control of the real-time clock 36, to produce the output signal 361.

At the end of the 0.01 second integration period, the output signal 351 from the 0.01 second integrator 349 is applied as an input to the squarer 353, which produces an output signal 355 equal to the square of the output signal 351. The output signal 361 from the integrator 359 at the end of the 0.01 integration period is applied to the squarer 363, which produces an output signal 365 equal to the square of the output signal 361. The output signal 355 and the output signal 365 are applied as inputs to the summer 367, which produces an early signal 354, which is therefore a measure of the signal energy received from the GPS satellite 12 during the integration period.

As previously explained, the simultaneous signal energy indication 348 from the simultaneous signal detector 350 is applied to the input to the C/A code group delay error detector circuit 401 to produce an above threshold indication 410. The early signal 354 and the late signal 356 are also applied as inputs to the C/A code group delay error detector circuit 401, and in particular are applied as inputs to the comparator 400 for an early-late comparison. The early/late comparator 400 operates to determine which of its input values is greater, and produce an early greater than late indication 402 when the early signal 354 is greater than the late signal 356, and produce an indication 404 of late greater than early when the contrary is true.

The early greater than late indicator 402 and the late greater than early indicator 404 are applied as inputs to a pair of identical logical AND gates 412 and 416. The other inputs to the AND gates 412 and 416 are provided by the above-threshold indicator 410 so that the respective delay register increment signal 414 and the delay register decrement signal 418 are enabled by the gates 412 and 416 when the concurrent signal energy indicator 348 is above the threshold 406. Gates 412 and 416 generate delay register increment signal 414 and delay register decrement signal 418, each taken together forming group delay error signal 200 shown above with reference to Figure 6.

In this way, it can be determined that the correlator 226 is for generating a value for the phase error signal 197, determining from the contemporaneous signal energy whether the phase error signal thus generated is valid, and if it is valid, passing that value as the phase error signal 198 to the phase and group delay register subsystem 196. Accordingly, the correlator 226 passes the group delay error signal 200 to increment or decrement the value for the 30-bit C/A code group delay estimate τest 876 held in the delay register 842 shown in Figure 13, according to the relative power available from the early or late cross-correlation.

Fig. 11

In Fig. 11, the C/A code generator 230 shown in Fig. 7 is shown in further detail. As described above under Referring to Fig. 7, the C/A code generator 230 generates the 1-bit C/A code local model 228 which is a pseudo-random sequence of 1023-bit values representing the C/A code specific to the GPS satellite 12. This sequence is transmitted by the GPS satellite 12 periodically with a period of 1 millisecond.

The 1-bit C/A code local model 228 from the C/A code generator 230 is applied to the correlator 226 in which the 1-bit C/A code local model 228 is correlated with the developed complex video signal U, which represents the in-phase developed video signal 222 and the quadrature developed video signal 224.

The 1-bit C/A code local model 228 constructively correlates only with the particular component of the developed complex video signal U that originates from the GPS satellite 12. The C/A code used in any other satellite is uncorrelated, i.e. orthogonal, with respect to the code of the GPS satellite 12. As a result of this orthogonality, the correlator 226 serves to select the signals from the GPS satellite 12 and reject the signals from other satellites.

As explained with reference to Fig. 2, the real-time computer 40 generates a satellite assignment 41 which is applied to the satellite tracking channel 32 to indicate the particular GPS satellite, namely GPS satellite 12, to be tracked by the satellite tracking channel 32. As shown in Fig. 6, within the satellite tracking channel 32, the satellite assignment 41 is applied to the 308 fo carrier phase and C/A code delay error detector 190, within which, as shown in Fig. 7, the satellite assignment 41 is applied to the C/A code generator 230. As explained below, the Satellite assignment 41 causes C/A code generator 230 to generate the particular code sequence, namely the 1-bit C/A code local model 228, which is specific to GPS satellite 12.

The C/A code generator 230 also receives the real-time digital display 34 from the real-time clock 36, the 8 fo reference frequency 27 from the doubler 25, and the C/A code group delay estimate est 194 from the delay register 842, as shown in Figure 7. These three inputs together determine the timing of the generation of the 1-bit C/A code local model 228.

As explained below with reference to Figure 13, the delay register 842 contains and produces the C/A code group delay estimate τest 194, an estimate of the current C/A code group delay observable τ of the C/A code component of the GPS signals 15 received from the GPS satellite 12. The C/A code group delay estimate τest 194 is adjusted by the action of the delay error signal 200 produced by the correlator 226 to minimize the difference between the C/A code group delay estimate τest 194 and the current C/A code group delay observable τ. Therefore, the C/A code generator 230 is part of a delay-coupled feedback loop which includes the correlator 226 and the delay register 842.

As shown in Fig. 11, the C/A code generator 230 includes a code memory 450, an adjustable modulo 1023 counter 452, and an adjustable modulo 40 counter 454. The satellite assignment 41 from the real-time computer 40 is applied to the code memory 450. The satellite assignment 41 preferably comprises the complete sequence of 1023 bits, which is to be reproduced by the C/A code generator 230 as a 1-bit C/A code local model 228. The transmission of the sequence specific to that satellite to the code memory 450 from the real-time computer 40 therefore has the desired effect of associating the C/A code generator 230 with the GPS satellite 12. Such a transmission must be performed whenever it is desired to begin tracking a satellite through the satellite tracking channel 32. The satellite assignment 41 is retained in the code memory 450 until a different satellite assignment 41 is received from the real-time computer 40.

It should be noted that the assignment of other portions of the satellite tracking channel 32 to the GPS satellite 12 is accomplished by means that do not require the knowledge and use of the satellite specific code. In particular, the L1 band fo carrier phase detector 206 and the L2 band fo carrier phase detector 208 in the satellite tracking channel 32 are assigned to the GPS satellite 12 by using a satellite specific fo carrier phase estimate 204, as shown in Fig. 6 and Fig. 12 and discussed in this context.

For special purposes, such as for experimental purposes, specific bit sequences may be transmitted to and stored in the code memory 450. Alternatively, the code memory 450 could be designed to store more than one sequence of bits and a particular sequence among these sequences could be selected using the satellite map 41. For example, every possible C/A code sequence for each GPS satellite could be permanently contained in the code memory 450 and the satellite map 41 could determine which code should be read from the code memory 450.

The code memory 450 receives the address 460 from the adjustable modulo 1023 counter 452. The address 460 controls the reading of bits from the code memory 450. The value of the address 460 is the address in the code memory 450 of the particular bit in the code sequence that is read from the code memory 450 as the 1-bit C/A code local model 228. Repeatedly incrementing the value of the address 460 causes the bits to be read sequentially.

As mentioned above, if the code memory 450 has been loaded with the 1023-bit sequence corresponding to the C/A code of the GPS satellite 12, with successive bits of the sequence being stored at successive addresses from 0 to 1022, then the correct 1-bit C/A code local model 228 can be generated by incrementing the address 460 in increments of one at a rate of 1,023,000 increments per second. As shown in Fig. 11, this is accomplished by deriving the address 460 from the adjustable modulo 1023 counter 452, which serves as a memory address counter, and incrementing the adjustable modulo 1023 counter 452 by the fo/5 input signal 458 from the adjustable modulo 40 counter 454.

The fo/5 input signal 458 has a frequency of fo/5, or 1,023,000 per second, so the address 460 increases at this rate. The adjustable modulo 1023 counter 452 counts up until a maximum value of 1022 is reached. After reaching a value of 1022, instead of increasing to 1023, the count value returns to zero, causing a return to the starting point of the 1023-bit C/A code sequence.

The adjustable modulo-40 counter 454 receives the 8 fo reference frequency 27 from the doubler 25 and divides that frequency by 40 to produce the fo/5 input signal 458 which is applied to the adjustable modulo-1023 counter 452.

The relative clock, i.e. the phase of the 1-bit C/A code local model 228 with respect to the real-time digital display 34 from the real-time clock 36, is controlled by the C/A code group delay estimate est 194 from the delay register 842. The millisecond clock 456 contained in the real-time digital display 34 is a pulse that occurs once per millisecond, each time exactly an integer multiple of milliseconds is displayed by the real-time digital display 34. In other words, the millisecond clock 456 marks integer milliseconds.

The millisecond clock 456 is applied to the inputs of the adjustable modulo 40 counter 454 and the adjustable modulo 1023 counter 452, which are arranged such that each occurrence of the millisecond clock 456 which marks an integer millisecond according to the real time clock 36 results in the instantaneous value of the C/A code group delay estimate est 194 being copied into the adjustable modulo 40 counter 454 and the adjustable modulo 1023 counter 452. In other words, the millisecond clock 456 enables the setting of the adjustable modulo 40 counter 454 and the adjustable modulo 1023 counter 452, and the C/A code group delay estimate est 194 provides the values to which these counters are currently set.

As shown in Fig. 11 and also shown and explained with respect to Fig. 13, the C/A code group delay estimate est 194 contains a 10-bit total chunk group delay estimate 874 from the modulo 1023 reversible counter 868 in the delay register 842. The 10-bit total chunk group delay estimate 874 indicates the total number of C/A code chunks, from zero to 1022, or modulo 1023, that should have been generated by the C/A code generator 230 at the occurrence of the millisecond clock 456.

The portion of a piece of C/A code that should have been generated by the C/A code generator 230 at the occurrence of the millisecond clock 456 is indicated by a tenth piece delay estimate 872 from the 4-bit, modulo-10 counter 866 of the modulo-40 counter 860 in the delay register 842 and by the 2-bit fo carrier phase estimate 204 from the 2-binary bit, modulo-4 counter 864 of the modulo-40 counter 860 in the delay register 842. As explained with reference to Figure 13, the fo carrier phase estimate 204 represents the delay in units corresponding to a quarter cycle of phase at a frequency of 2 fo. These units also represent 1/40th of a C/A code piece.

As shown in Figure 11, the 10-bit slice group delay estimate 874 is applied to the "value to be adjusted" input of the adjustable modulo 1023 counter 452 so that the instantaneous value of the 10-bit slice group delay estimate 874 will be copied into the adjustable modulo 1023 counter 452 at each occurrence of the millisecond clock 456.

The 2-bit fo carrier phase estimate 204 and the 4-bit estimate 872, which together comprise the entire content of the Modulo-40 counter 860 are applied together to the "value to be adjusted" input of adjustable modulo-40 counter 454 so that the instantaneous values of the fo carrier phase estimate 204 and the 4-bit estimate 872, which together form a fraction of a C/A code piece, are copied into adjustable modulo-40 counter 454 at each occurrence of a millisecond clock 456.

From one occurrence of the millisecond clock 456 to the next, the adjustable modulo 40 counter 454 and the adjustable modulo 1023 counter 452 increment at exactly uniform rates of 8 fo and fo/5, respectively, corresponding to the 8 fo reference frequency 27. Therefore, the 1-bit C/A code local model 228 is generated at a constant rate, except for discontinuous or "stepped" adjustments of the delay or code phase that occur at integer milliseconds corresponding to the millisecond clock 456 from the real-time digital display 34. It would be more desirable to generate the 1-bit C/A code local model 228 at a smoothly varying rate and phase. However, the discontinuities at the integer milliseconds are minor. The distance between the antenna 22 and the GPS satellite 12 typically changes less than 1 meter per millisecond. Every millisecond, the average delay of the 1-bit C/A code local model 228 generated by the C/A code generator 230 is offset by half the amount of the delay discontinuity occurring at the integer millisecond due to the stepwise nature of the delay change generated. This offset is reasonably taken into account in the processing of the C/A code group delay observable τ of the C/A code component of the GPS signals 15 by the real-time computer 40.

Fig. 12

In Fig. 12, the L1 band fo carrier phase detector 206 shown in Fig. 6 is shown in more detail. As explained with reference to Fig. 6, the L2 band fo carrier phase detector 208 has the same operation as the L1 band fo carrier phase detector 206, with the exception that it operates with signals received in the L1 frequency band. The phasor counter rotator 240 at the input of the L1 band fo carrier phase detector 206 receives the in-phase reconstructed L1 band fo carrier composite signal 88 and the quadrature phase reconstructed L1 band fo carrier composite signal 90 as inputs from the L1 band fo carrier reconstructor and downconverter 86, as shown in FIG. 3.

As explained above with reference to Fig. 9, the operation of the phasor counter rotator 240 is identical to that of the phasor counter rotator 220 shown in Fig. 7, with the except that the phasor counter rotator 240 operates with the L1 band reconstructed carrier composite signal phasor 89, whereas the phasor counter rotator 220 operates with the complex video phasor 83. The L1 band reconstructed carrier composite signal phasor 89 is the phasor in the complex plane that represents the composite signals of all individual satellite-specific L1 band reconstructed fo phasors, for example the L1 band reconstructed fo carrier phasor 8912 from the GPS satellite 12.

The phasor counter rotator 240 receives the fo carrier phase estimate 204 from the phase and group delay register subsystem 196 as its second input. As explained below with reference to Fig. 13, the fo carrier phase estimate 204 is received directly from the delay register 842, and indirectly from phase register 840, both of which are included in in-phase and group delay register subsystem 196. Fo carrier phase estimate 204 is a 2-bit signal representing only the quadrant of the estimated reconstructed fo carrier, with a phase modulo one cycle, at the reconstructed carrier frequency. As previously explained, the phase of the reconstructed fo carrier, i.e., the angle of the L1 band reconstructed fo carrier phasor 8912, is twice the corresponding fo carrier implicitly included in the complex video signal V, which is a composite signal of suppressed carrier and broad spectrum signals.

As explained above with reference to Fig. 1, implicit in both the L1 and L2 band portions of the suppressed carrier and broadened spectrum GPS signals 15 transmitted by the GPS satellite 12 is an implicit carrier wave having a frequency nominally equal to fo or 5.115 megahertz. fo is the fundamental frequency generated by a frequency standard (not shown) within the GPS satellite 12 and which forms the basis for the construction of all signals transmitted by the GPS satellite 12. The actual frequency of this implicit carrier may, of course, deviate from exactly fo depending on the accuracy of the satellite's frequency standard. The deviation of the actual implicit carrier frequency from exactly fo is normally less than about 0.05 hertz, or about 1 part in 108 of fo.

Broad spectrum GPS signals 15 from the GPS satellite 12 are received by the antenna 22 and fed to the receiver 24, as described with reference to Fig. 2 and 3, have been delayed and Doppler shifted as a result of the relative position and velocity of the antenna in the GPS satellite 12 (not shown) with respect to the antenna 22. The phases ψL1 and ψL2 of the fo carriers implicitly contained in the L1 and L2 bands of the received GPS signals 15 represent the group delays of the signals received in the respective bands. The L1 band fo carrier phase detector 206 and the L2 band fo carrier phase detector 208 measure ψL1 and ψL2, respectively. The Doppler frequency shifts of the fo carriers, implicitly contained in both the L1 and L2 bands of the received GPS signals 15, vary between plus and minus 15 hertz as the satellite rises, crosses the sky, and sets. Expressed as a fraction of fo, this Doppler frequency shift is between plus and minus three parts in 106. Therefore, the typical Doppler frequency shift is an order of two orders of magnitude larger than the fo carrier frequency deviation that normally occurs due to an offset of the frequency standard within the GPS satellite 12.

The radio frequency to video frequency downconversion performed within the radio frequency to video frequency downconverter 76 of Fig. 3 does not change the frequency of the fo carrier implicitly contained in the received GPS signals 15, nor does this frequency downconversion change the nature of these broadened spectrum signals. The complex video signal V, the real and imaginary parts of which are formed by the in-phase video signal 82 and the quadrature video signal 84, respectively, as shown in Fig. 3, represents a composite signal of the broadened spectrum signals received from all visible satellites. In the complex video signal V, the broadened spectrum component the GPS signals 15 have an implicit carrier with a frequency close to fo, which is shifted by between plus and minus 15 Hertz mainly due to the Doppler effect, as mentioned above.

The broadened spectrum components of the complex video signal V coming from satellites other than the GPS satellite 12 have implicit carrier frequencies that are also close to fo, but are shifted by different amounts, corresponding to the different satellite movements with respect to the antenna 22, and also corresponding to any differences between the offsets of the frequency standards in the different satellites.

The differences between the frequencies of the fo carriers implicitly contained in the signals received from different GPS satellites are used within the L1 band fo carrier phase detector 206 and within the L2 band fo carrier phase detector 208 to separate the signals received from the GPS satellite 12 from the signals received from different satellites. As mentioned above with reference to Fig. 12, an individual satellite tracking channel 32 is assigned within the terminal 23 to each satellite. Within the satellite tracking channel 32 associated with the GPS satellite 12, as shown in Figure 6, the L1 band fo carrier phase detector 206 and the L2 band fo carrier phase detector 208 receive the satellite-specific fo carrier phase estimate 204, namely, a prediction for the time-dependent phase of the fo carriers implicitly contained in the L1 and L2 band signals of that particular satellite. The temporal rate of change of the fo carrier phase estimate 204 represents the predicted frequency shift of the fo carriers implicitly contained in the signals received from the GPS satellite 12.

The fo carrier phase guard value 204 is applied to both the L1 band fo carrier phase detector 206 and the L2 band fo carrier phase detector 208 because theoretically the same fo carrier frequency shifts should be observed in both bands, barring the presence of any time-varying dispersion effects in the signal propagation paths. Minor dispersion effects are in fact present, due to the ionosphere and due to multipath propagation. However, in the present context these effects are normally insignificant.

As explained below with reference to Figure 13, the fo carrier phase estimate 204, and hence the frequency prediction used within the L1 band fo carrier phase detector 206 to distinguish the GPS satellite 12 signal from the signals of other satellites, is derived within the phase and group delay register subsystem 196 from the continuously updated 30-bit C/A code group delay estimate est 876 contained in the delay register 842. The reason for deriving the fo carrier phase estimate 204 from the 30-bit C/A code group delay estimate est 876 is that the time variation of the L1 fo carrier phase ψL1 is similar to that of the group delay of the C/A code modulation of the L1 signal from the same satellite. Therefore, the C/A code group delay variation multiplied by the implicit carrier frequency fo should approximate the implicit carrier phase variation. The required multiplication of the delay by the implicit carrier frequency fo is conveniently accomplished by tapping the appropriate stage in the chain of appropriate counters or dividers forming the delay register 842, as shown and explained in detail with reference to FIG. 13.

Referring again to Fig. 3, the complex video signal V from the radio frequency to video frequency downconverter 76 is applied to the L1 band fo carrier reconstruction and downconversion device 86, in which two basic operations are performed. As explained with reference to Fig. 5, the first basic operation performed within the L1 band fo carrier reconstruction and downconversion device 86 is to reconstruct from the complex video signal V the complex composite signal S(L1) for the L1 band reconstructed fo carrier, which is a composite signal of continuous wave components called reconstructed carriers, representing the L1 fo implicit carrier waves. A second composite signal of continuous wave components first appears at the output of mixer 150 shown in Fig. 5, in which separate spectral components of the complex video signal V are mixed together.

The phases and frequencies of the reconstructed carriers in this second composite signal relate to the phases and frequencies of the respective implicit carriers of the respective broadened spectrum signal components received from the respective satellites. Each reconstructed carrier has twice the phase and twice the frequency of the respective implicit carrier. As a result of the frequency doubling, the narrow band pass filter 152 shown in Fig. 5 has a center frequency of 2 fo instead of fo. As mentioned with reference to Fig. 5, in order to reduce the speed requirements for subsequent signal processing circuits, the reconstructed carriers in the second composite signal 154 of the narrow band pass filter 152 are then down-converted from the narrow band of frequencies centered at 2 fo to a narrow band centered at zero frequency. This downconversion yields the complex composite signal S(L1) for the L1 band reconstructed fo carrier, the real and imaginary parts of which are the in-phase composite signal 88 for the reconstructed L1 band fo carrier and the quadrature-phase reconstructed L1 band fo carrier composite signal 90, respectively.

The frequency shifts of the implicit fo carriers due to the Doppler effect and any satellite frequency standard offsets with respect to the frequency standard 28 of the terminal 23, as shown in Fig. 2, but doubled, are preserved in the downconversion process performed within the L1 band fo carrier reconstruction and downconversion device 86. The effect of this downconversion is to subtract the constant 2 fo reference frequency and phase of the local oscillator reference input signal 160, shown in Fig. 5, from the respective frequencies and phases of the reconstructed carriers.

As shown in FIGS. 2 and 3, the down-converted composite signal 88 for the in-phase reconstructed L1 band fo carrier and the quadrature phase composite signal 90 for the reconstructed L1 band fo carrier are included in the set of low frequency signals 30 that is input to each of the plurality of identical satellite tracking channels, such as the satellite tracking channel 32 shown in FIG. 6. As previously explained, within the satellite tracking channel 32 associated with the GPS satellite 12, the L1 band fo carrier phase detector 206 selects the particular reconstructed carrier signal from the GPS satellite 12 using the satellite specific frequency prediction generated by the phase and group delay register subsystem 196 and by the temporal rate of change of the fo carrier phase estimate 204 is represented.

Referring now to Fig. 12, the L1 band fo carrier phase detector 206 compares the phase observable ψL1 of the L1 band fo carrier of the GPS satellite 12 with the fo carrier phase estimate 204, and this yields the L1 fo residual phase measurement 210. Similarly, the L2 band fo carrier phase detector 208 compares the phase observable ψL2 of the L2 band fo carrier of the GPS satellite 12 with the fo carrier phase estimate 204, and this yields the L2 fo residual phase measurement 212. Since both the L1 and L2 related fo residual phase measurements are determined with respect to the same fo carrier phase estimate 204, any errors in this estimate are common to both the L1 and L2 related fo residual phase measurements. Therefore, any difference between the L1 and L2 related fo residual phase measurements is insensitive to errors in the fo carrier phase estimate 204. This insensitivity is important because the difference between the L1 and L2 related fo carrier phase measurements for each satellite is used by the real-time computer 40 to determine the amount of ionospheric delay and phase shift of the signals without loss of accuracy due to ionospheric effects.

Since the local oscillator reference input signal 160 used in the downconversion performed in the L1 band fo carrier reconstruction and downconversion device 86 to produce S(L1) which is applied to the L1 band fo carrier phase detector 206 is derived from the same 4 fo reference signal 26 from the same frequency standard 28 in the terminal 23 and which is also used in the L1 band fo carrier reconstruction and downconversion device 96 the L2 band fo carrier is used to generate S(L2) which is applied to the L2 band fo carrier phase detector 208, any phase or frequency changes of the frequency standard 28 will also be equally evident in both the L1 fo residual phase measurement 210 and the L2 fo residual phase measurement 212. Therefore, the difference between the L1 and L2 related fo residual phase measurements is also insensitive to phase and frequency changes of the frequency standard 28. This insensitivity is also important for accurately determining the amount of ionospheric delay and phase shift of the satellite's signals in order to determine position information without loss of accuracy due to ionospheric effects.

It will be readily seen that, for the same reason, the antenna 22 and the terminal 23 should be constructed and arranged so that, in general, any source of error affecting the L1 fo residual phase measurement 210 will tend to affect the L2 fo residual phase measurement 212 in the same way. For the same reasons, the overall position determination system shown in Fig. 1 should also be constructed and arranged so that any source of error affecting any measurement of any satellite signals will tend to affect corresponding measurements of all other satellite signals in the same way.

Referring again to Fig. 12, the in-phase composite signal 88 for the reconstructed L1 band fo carrier and the quadrature phase composite signal 90 for the reconstructed L1 band fo carrier are applied as inputs to the phasor counter rotator 240 at one input of the L1 band fo carrier phase detector 206. In addition, the fo carrier phase estimate 204 is applied to the other input of the phasor counter rotator 240. In a similar manner to that provided by the When used with the phasor counter rotator 220 shown in Fig. 7, the phasor counter rotator 240 serves to subtract a phase, in this case the fo carrier phase estimate 204, from the complex composite signal S(L1) for the L1 band reconstructed fo carrier.

As explained with reference to Fig. 8, the manner of phase subtraction can be understood as counter-rotating the composite phasor 89 for the L1-band reconstructed carrier, which represents the complex composite signal S(L1) for the complex L1-band reconstructed fo carrier. In the complex composite signal S(L1) for the L1 band reconstructed fo carrier, the composite phasor 89 for the L1 band reconstructed carrier is rotated, or "unwrapped" or "unwrapped" clockwise by an angle equal to the fo carrier phase estimate 204. However, as mentioned above with reference to the phasor counter rotator 220, only two bits are used to represent each phasor, and the output of the phasor counter rotator 240 is only a pair of one-bit digital signals, namely the in-phase, developed, reconstructed fo carrier residual phase 244 and the quadrature developed, reconstructed fo carrier residual phase 246, which together indicate only the quadrant of the developed, complex, composite, reconstructed carrier residual phase signal.

This 2-bit quadrant display may be considered too coarse, given the position determination accuracy that is desired. The phase quantization of a quarter cycle corresponds to a distance quantization of a quarter of the wavelength of a radio wave with a frequency equal to the reconstructed carrier frequency 2 fo, or about 10 megahertz. Therefore, the distance quantization is about 7.5 meters, and the RMS quantization error resulting from the quantization level divided by the square root of 12 is approximately 2 meters. It is important to note, however, that this quantization level refers only to the instantaneous value of the phase.

As explained below, measurements of the fo carrier phase observables ψL1 by the L1 band fo carrier phase detector 206 and ψL2 by the L2 band fo carrier phase detector 208 come from relatively long integrations of the evolved reconstructed carrier residual phase signals, preferably over time intervals of about 100 seconds. During such a long time interval, the fo carrier phase estimate 204 typically changes by very many phase cycles, and the change is relatively non-uniform due to the accelerations of the GPS satellite 12 and antenna 22. The fo carrier phase estimate 204 rotates one full cycle, or four quadrants, for every 30 meter change in the distance between the GPS satellite 12 and antenna 22.

The projection of the orbital velocity of the GPS satellite 12 along the line of sight to the antenna 22 is typically a few hundred meters per second, and typically changes at a rate of the order of 0.1 meters per second per second. Any acceleration of the antenna 22 will of course also contribute to this. Therefore, the error associated with quantization is averaged over a relatively large number of quarter-cycle phase steps during the course of each 100 second integration, so that the quantization noise remaining in the phase measurement is likely to be negligible.

It is also for reducing the quantization error in the L1-band fo carrier phase detector 206 and in the L2-band fo carrier phase detector 208, it is essential that the signal-to-noise ratio of the composite signal 88 for the in-phase reconstructed L1 bandfo carrier and the composite signal 90 for the quadrature-phase reconstructed L1 bandfo carrier is extremely low. The large amount of noise in these signals acts to smear or "jitter" the quantization.

As previously explained, the phasor counter rotator 240 subtracts from the complex composite signal S(L1) for the L1 band reconstructed fo carrier an amount of phase equal to the instantaneous estimate of the phase of the component of the selected satellite to enable selection of that reconstructed carrier component from the complex composite signal S(L1) for the L1 band reconstructed fo carrier. As a result of this substantially continuous phase subtraction, the phase of the reconstructed carrier component of the selected satellite is substantially stationary in the evolved complex output signal of the phasor counter rotator 240. Therefore, long term integration of the evolved complex output signal of the phasor counter rotator 240 can be used to select the desired signal and reject noise and signals from other satellites. The signals of other satellites are rejected because their evolved phasors will continue to rotate during the integration time interval. The opposite rotation performed by the phasor counter rotator 240 does not cause their phasors to become stationary because the fo carrier phase estimate 204 was not derived for them and they have different frequency shifts.

To perform this satellite selection process, the residual phases 244 for the in-phase, developed, reconstructed fo carrier and the residual phase 240 for the Quadrature-developed, reconstructed fo carriers from the phasor counter rotator 240 are applied as inputs to the in-phase integrator 248 and the quadrature integrator 250. The in-phase integrator 248 and the quadrature integrator 250 may be constructed in the same manner as the 0.01 second integrators 324 and 328 shown in Fig. 10, except that their integration period is set to 100 seconds instead of 0.01 seconds. Because the bandwidths of the signals processed in the L1 band fo carrier phase detector 206 are so much smaller than those involved in the correlator 226, it is possible to use logic circuits in the phasor counter rotator 240, the in-phase integrator 248, and the quadrature integrator 250 that are clocked at a correspondingly lower rate.

There are two reasons for using an extended integration period in the L1 band fo carrier phase detector 206. First, as already mentioned, the signal-to-noise ratio between the reconstructed fo carrier signals in the complex composite signal S(L1) for the L1 band reconstructed fo carrier and the background noise is considerably lower than the signal-to-noise ratio of the 308 fo carrier signals reconstructed in the mixers 302 and 308 in the detector 350 of Fig. 10 for the simultaneous signal, since the latter signals are strongly enhanced by the correlation with the fitted local model of the satellite-specific C/A code.

The lower signal-to-noise ratio in the complex composite signal S(L1) for the L1-band reconstructed fo carrier arises from the fact that the carrier reconstruction performed by the reconstruction and downconversion device 86 for the L1-band fo carrier is performed without the benefit of knowledge or use of the code associated with that carrier. The The longer integration period used in the in-phase integrator 248 and the quadrature integrator 250 improves the effective signal-to-noise ratio by reducing the equivalent noise bandwidth to 0.01 Hertz. In other words, the integration attenuates the noise contained in the residual phase 244 for the in-phase evolved reconstructed fo carrier and the residual phase 246 for the quadrature evolved reconstructed

fo carriers, with frequencies differing by more than about 0.01 Hertz from the frequency of the selected reconstructed carrier.

An equally important reason for the extended integration period used in the L1 band fo carrier phase detector 206 is to reject the reconstructed fo carrier signals from the other GPS satellites that are not tracked by that particular satellite tracking channel 32. As explained above, the Doppler shifts of the reconstructed fo carrier signals from the various GPS satellites generally differ within a range of plus and minus 30 hertz. The fo carrier phase estimate 204 applied to the pointer counter rotator 240 changes in time at a rate that reflects the Doppler shift of the GPS satellite 12 with which the satellite tracking channel 32 is associated.

The Doppler shift of any particular 2fo reconstructed carrier signal is equal to the frequency 2fo multiplied by the time derivative of the C/A code group delay observable τ. As the 30-bit C/A code group delay estimate τest 876 contained in the delay register 842 changes, the fo carrier phase estimate 204 obtained from the complex composite signal S(L1) also changes. of the L1-band reconstructed fo carrier is subtracted.

In this way, it can be seen that the phasor counter rotator 240 applies a frequency shift to the complex composite signal S(L1) for the L1 band reconstructed fo carrier, of equal magnitude and opposite sign to the Doppler shift characteristic of the GPS satellite 12 at that time, depending only on the accuracy of the fo carrier phase estimate 204.

Therefore, the rotation of the phasor for the L1 band reconstructed fo carrier of GPS satellite 12 has essentially stopped, as indicated by the 1-bit outputs of phasor counter rotator 240, namely the in-phase evolved reconstructed fo carrier residual phase 240 and the quadrature evolved reconstructed fo carrier residual phase 246, whereas the fo carrier phasors of other GPS satellites continue to rotate. Their rotation rates are equal to the differences between their frequency shifts and the frequency of the fo carrier phase estimate 204.

As explained above, the filter bandwidth effectively achieved by integrating for 100 seconds is about 0.01 Hertz. This value is about 1/3000th of the maximum Doppler shift magnitude of about 30 Hertz. Therefore, most of the time only the desired L1 band reconstructed fo carrier component lies within the passband of the 0.01 Hertz bandwidth filter formed by the in-phase integrator 248 and the quadrature integrator 250.

Sometimes it is obvious that a reconstructed fo carrier from another satellite has the same Doppler shift within 0.01 Hertz as the GPS satellite 12. However, these moments are irregular and the Doppler shifts of two different satellites do not remain equal to within 0.01 hertz for a very long period of time. It is particularly important that the Doppler shifts of two different satellites generally do not remain equal to within 0.01 hertz for a period as long as 100 seconds. Therefore, in effect, the L1 band reconstructed fo carrier component from the GPS satellite 12 is selected and the signal from any other satellite is rejected by the 100 second integrations performed by the in-phase integrator 248 and the quadrature integrator 250.

The in-phase integrator output 252 and the quadrature integrator output 254 can be viewed as the real and imaginary parts, respectively, of a complex number whose angle is the desired measurement of the L1-fo carrier residual phase. To determine this phase, the in-phase integrator output 252 and the quadrature integrator output 254 are input to the four-quadrant inverse tangent generator 256 whose output is the L1-fo carrier residual phase measurement 210. The four-quadrant inverse tangent generator 256 can be constructed and operate in virtually the same manner as the two-quadrant inverse tangent function generator 334 shown in Figure 10.

The four quadrant inverse tangent generator 256 produces the L1-fo carrier residual phase measurement 210 equal to the angle in the range minus one-half cycle to plus one-half cycle, whose sine is as follows

Q/(I² + Q²)1/2

whose cosine is equal to

I/(I² + Q²)1/2

and whose tangent is equal to Q/T, where I is the output 252 of the in-phase integrator and Q is the output 254 of the quadrature integrator, as shown in Fig. 12. Note that this inverse tangent function generator produces a result with a range of four quadrants or one cycle, whereas the inverse tangent generation performed by the two quadrant inverse tangent function generator 334 shown in Fig. 10 produces a result with a range of two quadrants or one-half cycle. In the four-quadrant inverse tangent generator 256, the generation of a four-quadrant tangent function is useful and preferable because the reconstructed fo carrier residual component present in the in-phase integrator output 252 and the quadrature integrator output 254 does not exhibit phase reversals due to telemetry modulation. However, since the phase of the fo carrier implicitly contained in the in-phase reconstructed L1 band fo carrier composite signal 88 and the quadrature phase reconstructed L1 band fo carrier composite signal 90 input to the L1 band fo carrier phase detector 206 has been doubled by the carrier reconstruction process performed by the L1 band fo carrier residual phase detector 206, the L1 fo carrier residual phase measurement 210 from the four quadrant inverse tangent generator 256 uniquely represents the phase of the implicit fo carrier modulo 0.5 cycles.

Fig. 13

In Fig. 13, the phase and group delay register subsystem 196 is shown in detail, which was previously explained with reference to Fig. 6. Since the Because the phase and group delay register subsystem 196 is at the heart of the operation of the terminal 23, an appreciation of the manner in which the phase and group delay register subsystem 196 interacts with other portions of the terminal 23 is useful for a clear understanding of the present invention.

As mentioned above with reference to Figure 6, the phase and group delay register subsystem 196 performs a major portion of the operation of the satellite tracking channel 32. Within the satellite tracking channel 32, the phase and delays contained in the GPS signals 15 from the GPS satellite 12 are tracked. Phase and delay measurements 202 contained in the measurements 42 are generated by the phase and group delay register subsystem 196 and applied to the real-time computer 40. The phase and group delay register subsystem 196 receives, included within the estimate 38, a group delay initial value 37 and a frequency estimate 39 from the real-time computer 40.

Furthermore, the phase and group delay register subsystem 196 generates the 308 fo carrier phase estimate φest 192 and the C/A code group delay estimate τest 194, both of which are applied as inputs to the 308 fo carrier phase and C/A code delay error detector 190.

As mentioned above with reference to Figure 9, the 308 fo carrier phase estimate Φest 192 represents an estimate of the phase modulo one cycle. The half cycle bit 812 and the quarter cycle bit 814 together indicate only the quadrant of the 308 fo carrier phase estimate Φest 192.

Accordingly, the C/A code group delay estimate τest 194 is modulo one millisecond, i.e. one period of the C/A code function. Therefore, the C/A code group delay estimate τest 194 provides only the fractional period portion of the group delay estimate to the 308 fo carrier phase and C/A code delay error detector 190.

The 308 fo carrier phase and C/A code delay error detector 190 determines the difference between the implicitly contained 308 fo carrier phase observable φ and the 308 fo carrier phase estimate φest 192, modulo one-half cycle, and generates a phase error signal 198. Similarly, the 308 fo carrier phase and C/A code delay error detector 190 determines the difference between the C/A code group delay observable τ and the C/A code group delay estimate τest 194, modulo one millisecond, and generates the delay error signal 200.

The phase error signal 198 and the delay error signal 200 are applied to the phase and group delay register subsystem 196 to form both a phase tracking feedback loop, i.e., a phase-locked loop, and a delay tracking feedback loop, i.e., a delay-locked loop.

As explained in more detail with reference to FIG. 7, the in-phase video signal 82 and the quadrature video signal 84 represented by the complex video phasor 83 are correlated with the 308-fo carrier phase estimate φest 192 by the phasor counter rotator 220 to produce the evolved complex video signal U represented by the evolved complex video phasor 223. , and which is then cross-correlated in correlator 226 with the 1-bit C/A code local model 228 to produce the phase error signal 198. The 1-bit C/A code local model 228, which is specific to the GPS satellite 12, is generated by the C/A code generator 230 with the appropriate delay under the control of the C/A code group delay estimate τest 194.

Referring now again to Figure 6, the phase and group delay register subsystem 196 further provides a fo carrier phase estimate 204 to the L1 band fo carrier phase detector 206 and the L2 band fo carrier phase detector 208. The fo carrier phase estimate 204 is in the form of modulo one-half cycle of oscillation at fo, i.e. one cycle at 2 fo, and has a 2-bit digital form similar to the 30 8 fo carrier phase estimate φest 192. The fo carrier phase estimate 204 provides the L1 band fo carrier phase detector 206 and the L2 band fo carrier phase detector 208 with an estimate for the quadrant of the L1 band reconstructed fo carrier phasor 8912 shown in Fig. 8D.

The L1 band fo carrier phase detector 206 produces the L1 fo carrier residual phase measurement 210 equal to the difference, averaged over 100 seconds, and modulo one-half cycle, between the L1 band implicit fo carrier phase observable ψL1 implicitly contained in the L1 band portion of the GPS signals 15 from the GPS satellite 12 and the fo carrier phase estimate 204. Similarly, the L2 band fo carrier phase detector 208 produces an L2 fo carrier residual phase measurement 212 equal to the difference, modulo one-half cycle, between the L2 band implicit fo carrier phase observable ψL2 implicitly contained in the L2 band portion of the GPS signals 15 from the GPS satellite 12 and the fo carrier phase estimate 204.

Referring now to Figure 13, the phase and group delay register subsystem 196 is shown in more detail. The phase and group delay register subsystem 196 includes two main subsystems, the phase register 840 and the delay register 842.

The phase register 840 is divided into two parts, the phase register lower part 844 and the phase register upper part 846. The phase register 840 contains the 48-bit binary 308-fo carrier phase estimate Φest 856, which includes the 16 least significant bit phase estimate 852 generated by or contained within the phase register lower part 844, and the 32 most significant bit phase estimate 854 generated by or contained within the phase register upper part 846. The most significant bit (MSB) 848 generated by or contained within the phase register lower part 844 represents one-quarter of a cycle of the 308-fo implicit carrier phase observable Φest 856. The least significant bit (LSB) 850 generated by or contained in the phase register top 846 represents one-half of a cycle of the 308-foimplicit carrier phase observable Φ.

The phase register upper portion 846 generates the half cycle bit 812, which is applied to the pointer counter rotator 220 shown in Figure 9 and is equal to the least significant bit (LSB) 850. Similarly, the phase register lower portion 844 generates the quarter cycle bit 814, which is applied to the pointer counter rotator 220 and is equal to the most significant bit (MSB) 848.

The delay register 842 is formed by three cascaded, reversible up/down counters: a modulo-77 counter 858, a modulo-40 counter 860, and a Modulo 130944 counter 362. Therefore, unlike phase register 840, delay register 842 cannot be considered a binary numeric register. Instead, delay register 842 contains a mixed-based numeric representation of the C/A code group delay estimate τest 876, including the fo carrier phase estimate 204 and the C/A code group delay estimate τest 194.

The modulo-77 counter 858 within the delay register 842 contains a 7-bit number in the range 0 to 76 (decimal) representing the group delay in units of a half-period of oscillation at a frequency of 308 fo. Therefore, the modulo-77 counter 858 contains values for the group delay expressed in units corresponding to the phase unit of the least significant bit (LSB) 850 of the phase register top 846. These units are equivalent in the sense that each represents a change in position equal to one-half wavelength of a radio signal having a frequency equal to 308 fo propagating in a vacuum.

The modulo 40 counter 860 within the delay register 842 contains a 6-bit number in the range 0 to 39 (decimal) representing a group delay in units equal to 77 half-periods of oscillation at a frequency of 308 fo, which are the same units as one-quarter of an oscillation period at a frequency of 2 fo.

The modulo-40 counter 860 is formed by two stages in cascade: the 2-binary bit, modulo-4 counter 864, which serves as the input stage; and the 4-bit, modulo-10 counter 866, which serves as the next stage.

The 2-binary bit, modulo-4 counter 864 within the modulo-40 counter 860 generates or contains the 2-bit fo carrier phase estimate 204 in the range of 0 to 3 (decimal), representing the group delay in units of a quarter of an oscillation period at a frequency of 2 fo. As mentioned above with reference to FIG. 6, the 2-bit fo carrier phase estimate 204 is applied to the L1 band fo carrier phase detector 206 and the L2 band fo carrier phase detector 208.

Note that the fo carrier phase estimate 204 is derived directly from the C/A code group delay estimate τest 876, not from the 308 fo carrier phase estimate φest 856. This is because the temporal variations of the fo implicit carrier phases can be expected to more closely resemble those of the C/A code group delay observable τ than those of the 308 fo implicit carrier phase observable φ. It would be possible to use the 308 fo carrier phase estimate φest 856 instead, or in addition, if desired, as a basis for deriving the fo carrier phase estimate 204.

The 4-bit, modulo-10 counter 866 within the modulo-40 counter 860 generates or contains the 4-bit estimate 872 in the range of 9 to 9 (decimal) representing a group delay in units of an oscillation period at a frequency of 2 fo. This unit of group delay is also equal to a slice width of the P-code modulation, and therefore equal to one-tenth of a slice width of the C/A code modulation. As mentioned above with reference to FIG. 11, the 2-bit fo carrier phase estimate 204 and the 4-bit estimate 872 are applied to the C/A code generator 230 as parts of the C/A code group delay estimate τest 194.

The modulo 130944 counter 862 is formed by two stages in cascade: a 10-binary-bit, modulo 1023 invert counter 868 serving as the first stage; and a 7-bit, modulo 128 invert counter 870 serving as the second stage.

The 10-binary bit, modulo 1023 inversion counter 868 within the modulo 130944 counter 862 generates or contains the 10-bit whole piece group delay estimate 874 in the range of 0 to 1022 (decimal) representing a group delay in units of whole pieces of C/A code modulation. The 10-bit whole piece group delay estimate 874, the 4-bit estimate 872 from the 4-bit, modulo-10 counter 866, and the fo carrier phase estimate 204 together form the C/A code group delay estimate τest 194. The 7-bit, modulo-128 inverse counter 870 generates or contains a number in the range 0 to 127 (decimal) representing a delay in units of a millisecond, the period of the C/A code.

Before delay-locked tracking of GPS signals 15 from GPS satellite 12 begins, a program is executed by real-time computer 40 to calculate the group delay seed 37 to be applied to modulo 130944 counter 862 based on external information 44, as shown in Figure 2. If external information 44 is undetermined or unavailable, a series of trial values for group delay seed 37 may be formed by the real-time computer. Group delay seed 37 is applied to 10 binary bit, modulo 1023 inverse counter 868 and also to modulo 128 inverse counter 870 within modulo 130944 counter 862. It is noted that the least significant bit of the group delay initial value 37 represents a piece of the C/A code. This accuracy is sufficient to detection of the satellite signal by the 308 fo carrier phase and C/A code delay error detector 190. This accuracy, which is considerably greater than the repetition period of the C/A code, is also sufficient to eliminate integer period uncertainty in the interpretation of the delay.

The 30-bit C/A code group delay estimate τest 876 is applied to the real-time computer 40 and comprises the 7-bit contents of the modulo 77 counter 858, the 6-bit contents of the modulo 40 counter 860, and the 17-bit contents of the modulo 130944 counter 862. The total value of the 30-bit C/A code group delay estimate τest 876 is equal to the sum of the numbers contained in each stage of each counter in the delay register 842, each multiplied by the respective delay unit. The total value of the 30-bit C/A code group delay estimate τest 876 is therefore in the range of 0 to 128 milliseconds, and has a numerical accuracy of about 0.3 nanoseconds, the delay unit of the modulo-77 counter 858. Of course, the instantaneous value of the 30-bit C/A code group delay estimate τest 876 differs from the C/A code group delay observable τ by more than 0.3 nanoseconds, due to noise, multipath interference, etc.

The contents of the delay register 842, i.e. the value of the 30-bit C/A code group delay estimate τest 876, can be changed, other than by initialization by the real-time computer 40, by a delay register increment 878 from an OR gate 890 and by a delay register decrement 880 from an OR gate 892, both of which are applied to the modulo-77 counter 858. The delay register increment 878 increments the value in the modulo-77 counter 858 by one and the delay register Decrement 880 decreases the value in modulo-77 counter 858 by one.

If the delay register increment 878 is received by the modulo-77 counter 858 while the value 858 contained in the modulo-77 counter is equal to 76, then the value changes to zero and the modulo-77 counter 858 generates an overflow signal 882 which is applied to the modulo-40 counter 860 to cause the value in the modulo-40 counter 860 to be incremented by one. If the delay register decrement 880 is received by the modulo 77 counter 858 while the value in the modulo 77 counter 858 is equal to zero, then the value changes to 76 and the modulo 77 counter 858 generates an underflow signal 884 which is applied to the modulo 40 counter 860 and causes the value in the modulo 40 counter 860 to be decremented by one.

In the same manner, the modulo 40 counter 860 generates an overflow signal 886 and an underflow signal 888, which are applied to increment and decrement the count value in the modulo 130944 counter 862. During normal operation, the modulo 130944 counter 862 does not exhibit an overflow or underflow.

The OR gate 890 generates the delay register increment 878 in response to either the delay register increment signal 414 within the delay error signal 200 or the phase adder carry signal 894 from the phase adder 898. Similarly, the OR gate 892 generates the delay register decrement 880 in response to either the delay register decrement signal 418 within the delay error signal 200 or the phase adder borrow signal 896 from the phase adder 898.

The delay error signal 200 is generated by the correlator 226 shown in Figure 10 to indicate the sign of the difference between the C/A code group delay observable τ and the C/A code group N delay estimate τest 194, i.e. whether the energy detected by the early signal detector 252 is greater or less than the energy detected by the late signal detector 358. The delay register increment signal 414 within the delay error signal 200 indicates that the 30-bit C/A code group delay estimate τest 876 should be increased to better match the C/A code group delay observable τ , whereas the delay register decrement signal 418 indicates that the 30-bit C/A code group delay estimate τest 876 should be decreased to better match the C/A code group delay observable τ.

The phase adder 898 generates the phase adder carry signal 894 which is applied to an input of the OR gate 890 and to the increment input to the phase register upper portion 846, and the phase adder borrow signal 896 which is applied to an input of the OR gate 892 and to the decrement input to the phase register upper portion 846, in response to the sum of its three input signals: the phase error signal 198 from the two quadrant inverse tangent function generator 334, the phase estimate 852 of the 16 least significant bits from the phase register lower portion 844, and the carrier frequency register output signal 900. The phase adder 898 adds at least two of these three 16-bit binary numbers 110,000 times per second in response to a clock signal contained in the real-time digital display 34 from the real-time clock 36.

The phase adder 898 receives the phase error signal 198 only at time intervals corresponding to the real-time digital indication 34 from the real-time clock 36 which are integer multiples of 0.01 seconds. As explained with reference to FIG. 11, the phase error signal 198 is only available at such time intervals, and then only when the concurrent signal energy indication 348 exceeds the threshold 406, such that the indication 410 is TRUE for a value above the threshold, indicating that a valid phase error measurement was possible.

The phase adder 898 receives the phase estimate 852 of the 16 least significant bits from the phase register sub-portion 844 exactly 110,000 times per second, at uniformly spaced time intervals corresponding to the real-time digital display 34. The phase adder 898 also receives the carrier frequency register output signal 900 from the 16-bit frequency register 902, exactly 110,000 times per second, at uniformly spaced time intervals corresponding to the real-time digital display 34.

The carrier frequency register output signal 900 is also applied to the real-time computer 40 as part of the phase and delay measurements 202.

Therefore, 109,900 times per second, the phase adder 898 can receive and add together only two numbers, namely the 16 least significant bit phase estimate 852 and the carrier frequency register output signal 900; and 100 times per second, the phase adder 898 can receive and add together three numbers, including the phase error signal 198. At a non-variable rate of 110,000 times per second, the phase adder 898 provides the 16-bit phase sum 904 and, if required, the phase adder carry signal 894 or the phase adder borrow signal 896.

The phase adder carry signal 894 indicates that the addition by the phase adder 898 produced a sum that overflowed 16 bits and the phase adder borrow signal 896 indicates that the addition produced a sum that underflowed 16 bits.

The 16-bit phase sum 904 is generated 110,000 times per second, and is applied to the phase register lower portion 844 each time to replace its contents. The 16-bit phase sum 904 always includes a contribution from the 16 least significant bit phase estimate 852 and the carrier frequency register output signal 900, and may include the phase error signal 198. Whenever the phase adder carry signal 894 is generated from the phase adder 898, the phase register upper portion 846 is incremented by one. Each time the phase adder borrow signal 896 is generated, the phase register upper portion 846 is decremented by one. As mentioned above, the phase adder carry signal 894 is also applied to the OR gate 892 to increment the delay register 890 and the phase adder borrow signal 896 is also applied to the OR gate 892 to decrement the delay register 842.

The 16-bit frequency register 902 receives the frequency sum 906 from the frequency adder 908 100 times per second and replaces its contents with the frequency sum 906 at that rate.

The frequency adder 908 can receive three 16-bit binary number input signals, namely the carrier frequency register output signal 900 from the 16-bit frequency register 902; the phase error signal 198 from the two-quadrant inverse tangent function generator 334; and the frequency estimate 39 from the real-time computer 40.

The inclusion of the phase error signal 198 in the inputs to the frequency adder 908 helps to improve the dynamic response of the phase locked loop (PLL) formed by the 308-fo carrier phase and C/A code delay error detector 190 and the phase and group delay register subsystem 196, discussed with reference to Figure 6. The phase error signal 198 must be properly scaled before it is input to the frequency adder 908.

The frequency estimate 39 is included in the inputs to the frequency adder 908 for two separate purposes, for initialization and due to satellite loss.

The frequency estimate 39 is required in the frequency adder 908 during initialization so that the signal from the GPS satellite 12 can be fetched before the phase-locked loop has begun tracking the 308 fo implicit carrier phase observable Φ. As mentioned above with respect to Fig. 10, the 0.01 second integrators 324 and 328 in the correlator 226 have an effective bandwidth of about 100 hertz. The in-phase developed video signal 222 and the quadrature developed video signal 224 applied to the correlator 226 must therefore be accurately generated by the phasor counter rotator 220 such that the signal to be integrated lies within the effective bandpass of these integrators. In other words, the 308-fo carrier phase estimate Φest 192 applied to the pointer counter rotator 220 as shown in Fig. 7 and Fig. 9 must exactly reflect the 308-fo implicit carrier frequency of the signal from the GPS satellite 12 as received to within about 50 Hertz.

Accordingly, the frequency estimate 39 maintains the 30-bit C/A code group delay estimate τest 876 generated by the phase register 840 and the 308 fo carrier phase estimate φest 856 generated by the delay register 842 sufficiently accurate even if the GPS satellite 12 happens to be momentarily occluded. Operation in this manner is analogous to navigation by locking, namely, maintaining an estimate of position based on a previously known position and an estimate of position change determined from an assumed rate of change of position multiplied by elapsed time. The frequency adder 908 may be configured to add the phase error signal 198 and the carrier frequency register output signal 900 together to produce the frequency sum 906 if and only if the above-threshold indicator 410 indicates that the phase error signal 198 is available. If the above-threshold indicator 410 indicates that the phase error signal 198 is not available, then the frequency estimate 39 must serve as a substitute for the phase error signal 198. Even if the phase error signal 198 is available, the frequency estimate 39 may be used by the frequency adder 908 to "help estimate."

The carrier frequency register output signal 900, which as mentioned is contained in and generated by the 16-bit frequency register 902, represents the amount by which the phase register 840 would be incremented or decremented each time, namely 110,000 times per second, if there were no input to the phase adder 898 from the phase error signal 198. Therefore, the 16-bit frequency register 902 may properly be referred to as a phase rate register. When the phase locked loop formed by the 308-fo carrier phase and C/A code delay error detector 190 and the phase and group delay register subsystem 196, is phase locked, then the carrier frequency register output signal 900 represents a measurement of the frequency of the 308 fo L1 band center frequency carrier implicitly contained in the received GPS signals 15.

Fig. 14

In Fig. 14, a simplified schematic signal flow diagram is shown to explain the operation of the terminal 23, including the operation and inter-connections between the various phase and delay locked loops and other associated feedback control and measurement loops within the signal processing circuits.

In this schematic signal flow diagram, some details that have been described in discussions of other figures have been omitted for simplicity. In particular, some of the signals discussed with reference to the previous figures were described as having an analog form and others were described as having a digital form. Some of the digital signals were represented by different numbers of bits, and some of the signals were complex signals with both in-phase and quadrature parts, i.e. real and imaginary parts. However, in this Figure 14, such details are not shown. Signals are discussed with reference to analog or digital form, multiple parts or bits.

Furthermore, for the sake of simplicity, multiple satellite tracking channels 32 are not shown in Fig. 14. The operation of such a satellite tracking channel 32 is discussed based on the signals from only one GPS satellite. It should be noted that parallel operations are performed simultaneously or concurrently for the signals from each of the multiple GPS satellites being observed.

Antenna 22 receives a composite signal of overlapping, broadened spectrum L-band radio frequency signals simultaneously from multiple GPS satellites. The composite signal received by antenna 22 in the L1 and L2 frequency bands is applied via transmission line 600 to a pair of mixers 602 and 605. In mixer 602, the L1 band signals are down-converted by mixing with a locally generated 308 fo local model 604 from a local oscillator 606. Note that all of the oscillators and other time-critical circuits and operations shown in Figure 14 are governed by a single frequency standard, not shown, similar to frequency standard 28 shown in Figure 2.

The video frequency output signal 608 from the mixer 602 is applied to the counter rotator 610, which subtracts the phase of the phase estimate 612 from the video frequency output signal 608 to produce the output video signal 630.

If the frequency of the center frequency carrier wave, nominally at a frequency of 308 fo, implicitly contained in the signals in the transmission line 600 is greater than the frequency of the 308 fo local model 604, then the phase of the carrier implicitly contained in the satellite signals increases faster over time than the phase of the 308 fo local model 604. The phase of the center frequency carrier wave implicitly contained in the video frequency output signal 608 produced by the mixer 602 as the product of the satellite signals and the 30 8 fo local model 604 therefore increases over time.

The phasor representing the video frequency output signal 608 therefore rotates in the positive, i.e. counterclockwise, direction in the complex plane. The rotation rate in revolutions per second is equal to the frequency difference in Hertz between the frequency of the implicit center frequency carrier of the satellite signals and the frequency of the 308-fo local model 604. The Doppler shift of the received satellite signals, due to the relative motion between the satellite and the receiving antenna 22, is usually the main reason for this frequency difference. Typically, the Doppler shift of the 308-fo carrier, which is implicit in the received GPS satellite signals, is on the order of a few kilohertz. In other words, the rotation rate of the pointer representing the video frequency output signal 608 is several thousand revolutions per second.

The counter rotator 610 can be viewed as rotating the coordinate axes of the complex plane such that the signal pointer does not rotate as quickly with respect to the new axes of rotation.

The phase estimate 612 represents an angle in the range between zero and one cycle, or 0º to 360º. This angle is the instantaneous angle, modulo one cycle, by which the rotated coordinate axes to which the output video signal 630 refers are rotated with respect to the stationary coordinate axes to which the video frequency output signal 608 refers by the counter rotator 610. The phase estimate 612 changes over time and may increase or decrease.

The phase estimate 612 is stored in the phase register 616 from a combination of externally and internally derived information The term "internally derived information" refers to information derived from measurements made within the satellite tracking channel 32. Such measurements include measurements of signal properties in addition to the 308 fo implicit carrier phase observable Φ. The term "externally derived information" refers to information derived from sources external to the satellite tracking channel 32. Externally derived information may include a priori information derived from signals received at other locations, such as a shore station 18 shown in FIG. 1, as well as information derived from measurements within other satellite tracking channels 32 within the same terminal 23 of signals received from other satellites by the antenna 22.

Therefore, the phase register 616 receives a phase feedback 628 containing information from within the satellite tracking channel 32 about the phase value 680 and from the real-time computer 40 about the rate information 684, which may be generated by the real-time computer 40 based on information from a wide variety of other sources.

The goal is for the phase angle of the phase estimate 612 to rotate at as close a rate as possible to the phase angle of the video signal pointer representing the video frequency output signal 608. Ideally, the phase angle of the phase estimate 612 would rotate at the same rate as the phase angle of the video frequency output signal 608, so that the phase angle of the output video signal 630 from the counter rotator 610 would be stationary. Ideally, the value of this stationary phase angle would be zero. If the phase angle of the output video signal 630 is close to zero, the phase estimate 612 is close to the phase of, and can be used to measure, the 308 fo center frequency carrier from the selected satellite which is implicitly contained in the video frequency output signal 608, i.e. the 308 fo implicit carrier phase observable Φ.

For normal operation, it is sufficient to reduce the rotation rate of the output video signal 630 to less than about 50 revolutions per second. The frequency, i.e. the time derivative of the phase estimate 612, should not differ by more than 50 hertz from the frequency of the 308 fo center frequency carrier implicit in the video frequency output signal 608 from the selected satellite. Once the rotation rate has been successfully reduced to less than about 50 revolutions per second, it is possible, as indicated below, to average the signal over a time interval of 0.01 seconds without the pointer rotating more than halfway around the complex plane during that interval. It is desirable to average the signal over a time interval on the order of about 0.01 seconds to reduce the effects of noise and interference that may be present. Averaging over about 0.01 seconds reduces the effective bandwidth of the terminal 23 to about 100 Hertz, which is the inverse of the integration time. A long integration time and therefore a narrow bandwidth are desirable to reduce noise and the like, but the integration time must not be so long that the pointer to be integrated or averaged rotates more than about half a cycle. If the pointer rotates a full cycle, its integrated value or average is zero.

The signals received by the GPS satellite are not continuous wave signals with smoothly changing phases over time. In fact, the phase of each orthogonal component of a GPS satellite signal is switched very rapidly between two different values which are 180° apart, due to the biphase modulation employed in the satellite transmitter. This switching rate is approximately 1,000,000 times per second for the C/A code related component of the satellite signal, and 10,000,000 times per second for the P code related component, and is in any case considerably faster than the rate of rotation of the pointer discussed above. The "instantaneous" pointer representing the video frequency output signal 608 at any given time therefore switches very rapidly between two points on opposite sides of the origin of the complex plane, and these two points, while remaining on opposite sides of the origin, move relatively slowly along a circle whose center lies at the origin of the plane.

The 308 fo center frequency carrier wave implicit in the satellite signals can be represented by a rotating but non-switching pointer pointing to one of the diametrically opposite rotating points. The carrier of a GPS satellite signal is suppressed by the modulation in the satellite because the resulting averaged position of the "instantaneous" pointer is exactly halfway between the two opposite points, and therefore at the origin of the complex plane.

If no distinction can be made between the two opposite points, then the phase angle of this suppressed, implicit carrier has an uncertainty of 180º. This half-cycle uncertainty can be compensated or eliminated if one of the opposite points can be distinguished from the other. Such a distinction can be done if a certain pattern in the biphase modulation is known which can serve to identify one of the two phase points as the "zero" phase point, as opposed to the phase point offset by 180º. The pattern of the C/A code function, in combination with a related pattern, for example the frame synchronization pattern of the GPS navigation data modulation, can be used for this purpose. Even a partial knowledge of such patterns can be sufficient to distinguish the two phase points. An example of sufficient partial knowledge is the knowledge that the C/A code sequence has unequal numbers of "1" and "0" bits, in combination with a knowledge of a fragment of the navigation data or a partial pattern therein.

The counter rotator 610 removes most of the rotation rate or angular velocity of the pointer, but does nothing to remove the very rapid, namely 1,000,000 times per second, wobble or jump of the current pointer between the two diametrically opposed points on the circle centered at the origin of the complex video plane. To remove this wobble, the received signal is multiplied by the C/A code replica 634, which jumps back and forth in almost perfect synchronization with the jumps of the received signal.

The C/A code replica 634 is a replica or "model" of the C/A code component of the modulation of the signal transmitted by the selected satellite. The C/A code replica 634 is generated in the C/A code generator 636 and is multiplied by the output video signal 630 in the correlator 632. The timing of the generation of the C/A code replica 634 is governed by the time delay estimate τest 696 from the delay register 672.

The time delay of the C/A code replica 634 must be continually adjusted to follow the time delay of the signal received from the satellite. The time delay of the C/A code replica 634 is controlled by the delay estimate 696 contained in and generated by the delay register 672.

Like the contents of phase register 616, the contents of delay register 672 are adjusted to track the satellite signal by combining externally derived information with information derived by observing the satellite signal. However, while phase register 616 is adjusted to track the center frequency carrier phase, which relates to the phase delay of the signal, delay register 672 is adjusted to track the group delay of the signal.

Separate but interconnected registers are used for the phase delay and group delay estimates, despite the fact that the phase delay and group delay of the satellite signal are affected in virtually the same way by a change in distance, i.e., by a shift in position of the satellite along the line of sight from the receiving antenna 22. A change in distance of this kind results in a change in the time required for the signal to propagate from the satellite through space to the receiver.

One reason for producing separate estimates for the phase delay or "phase" and the group delay or "delay" by separate registers is that the phase delay and the group delay of the signal are not affected in the same way by propagation through the Earth's ionosphere. Another reason for The reason for maintaining separate estimates is that signal multipath affects the phase delay and the group delay differently. Another reason is that the uncertainty about an integer cycle or half an integer cycle of the implicit carrier phase observable is quite different from any uncertainty of the group delay observable. Another reason is that the two observables have different offsets and should therefore be treated separately.

The 308 fo implicit carrier phase observable Φ has the highest accuracy of the observables. Its corresponding range accuracy is on the order of 1 centimeter fo less, mainly because the wavelength of the 308 fo implicit carrier wave is so short, about 19 centimeters, and because errors due to multipath and noise are usually a small fraction, typically about 1/20th, of the relevant wavelength. The accuracy of the 308 fo implicit carrier phase observable Φ is also the highest in the time sense. This phase can be accurately measured based on signal integration or averaging of only 0.01 seconds. Relatively fast, statistical movements of the receiving antenna 22 therefore do not prevent an accurate measurement, and a rapidly changing position can be determined as a function of time with excellent temporal resolution and also with excellent spatial resolution.

The main disadvantage of the 308-fo carrier phase observable is that, partly because the wavelength is so short that the inevitable integer wavelength uncertainty in the interpretation of the phase is difficult to resolve, the observed value of the 308-fo carrier phase has a offset of unknown magnitude, possibly of enormous size. This offset must be determined if the power of these observables is to be fully exploited. As long as observations are made continuously or in effect continuously, the temporal change of the phase can be tracked so that the continuous series of observations contain only a constant, additive offset. Therefore, a potentially long period of time is permissible for determining the offset.

The offset in the time series of 308-fo carrier phase observations can be determined by one or a combination of techniques. One technique takes advantage of the fact that the partial derivative of a phase observable with respect to a position coordinate, a clock synchronization parameter, a satellite orbit parameter, or any of many other parameters that may be unknown but must be determined, changes in time due to the satellite's movement across the sky.

Although these partial derivatives vary with time, reflecting the sensitivities of the observables to all other unknown variables, the partial derivative of the observables with respect to the additive offset remains constant, equal to one. Therefore, a set of coupled equations relating all these unknowns to the observations in the series of observation times can be solved to obtain a unique solution or a least squares solution that separates the solution for the offset from the solutions for the other variables.

Although this general procedure for determining the time-varying position of a moving ship, additional information is required to solve the offset problem. If no additional information were available, then for each time instant there would be one or more additional, independent unknowns related to position. The additional presence of an unknown offset may result in more unknowns than knowns, and thus a unique solution would be impossible. However, if there are a sufficient number of satellites observed simultaneously over a sufficiently long period of time, and there is sufficient continuity in both the observation series and the position series or trajectory, a useful solution can still be obtained for the set of relevant offsets, simultaneously with a set of variables representing the trajectory.

Another method of determining the offset in the time series of 308-fo carrier phase observations is to use observations of some other observable. This method can be used alone or in combination with the method described above. The other observable can be the phase of another implicit carrier of a different frequency. If the second carrier frequency is sufficiently low, then its wavelength can be large enough that its uncertainty, integer or half-integer, can be resolved, for example by using a priori position information. If the second carrier frequency is close to the frequency of 308-fo, then it may be possible to resolve the integer uncertainty associated with the beat frequency between the two carrier frequencies. The latter beat frequency approach is similar to the "wide roads" method used in navigation with the Omega system.

A major problem with using any other carrier phase observable to resolve the offset uncertainty is that the new observable has its own offset and uncertainty problems. If the effective wavelength associated with the second carrier phase observable, used either directly or indirectly as in the beat frequency case, is very large, then the uncertainty about the integer wavelength may be easy to resolve, but due to the same fact the accuracy of the resulting position fix is poor. The position fix is poor because in general the uncertainty increases proportionally to the wavelength.

In contrast, if the effective wavelength associated with the second carrier phase observable used either directly or indirectly as in the case of a beat frequency wavelength is very small, then the potential accuracy is correspondingly very high; unfortunately, however, it may happen that this potential cannot be exploited because it is impossible to resolve the uncertainty with respect to integer multiples.

Therefore, in the preferred embodiment, an observable other than a carrier phase observable is used to determine the offset in the carrier phase observation. In the preferred embodiment, the C/A code group delay observable τ is used for this purpose. The C/A code group delay observable τ is a group delay observable. Since the C/A code is a periodic function with a period of 1 millisecond, the so-called "code phase", which represents the delay measured in code periods modulo 1, has a potential uncertainty of 300 kilometers, namely the distance that a signal travels in 1 millisecond. Such widely spaced uncertainties can be easily resolved using very coarse a priori position information. This uncertainty can also be resolved by using the 50-bit-per-second navigation data modulation of the GPS signal, either by decoding the navigation message or by measuring the phase of the 25-hertz carrier associated with this modulation. These methods for resolving uncertainty about an integer multiple of a millisecond can be combined.

The accuracy of the C/A code group delay observable τ is limited, mainly by noise and multipath effects, to a value of typically about 15 meters. This is the level of accuracy that can be obtained with a relatively short signal integration or averaging time, on the order of 1 second or less. This level of accuracy is related to and is governed by the bandwidth of the C/A code modulation, which is on the order of 1 megahertz. Note that a wavelength at 1 megahertz is 300 meters, and that 15 meters is 1/20th of 300 meters.

The noise and multipath effects which cause the majority of the error in an "instantaneous" measurement of the C/A code group delay observable τ, averaged over 1 second, are relatively large, but usually do not exhibit a significant offset, in the sense that the error can be reduced to a relatively small value by averaging a series of observations over a long period of time. Quantities related to the observations, for example series of instantaneous position determinations, each such determination being derived from C/A code observations of several satellites at one time, can be time averaged. However, such an approach suffers from the difficulty that the desired position information may vary over time and that it may be desired to determine the position immediately, not on the basis of a long time averaging. Conventional time averaging would yield unacceptable data relating to the mean position of the vessel 10 during the integration period, not its current position.

In the preferred embodiment, measurements of the C/A code group delay observable τ are used to determine position, and a long-term average is performed in such a way as to reduce or average out fluctuating errors in these observations. However, the C/A code delay measurements are not simply time averaged directly, since the position of the receiving antenna 22 may change over time and it is not desirable to average the position information.

Instead, one determines the difference between the 308fo implicit carrier phase observable φ and the C/A code group delay observable τ for the same satellite at the same time, and this difference as a function of time is time averaged. Of course, before the difference is determined, both observables must be expressed in the same units, such as time delay or corresponding distance. Movement of the antenna 22 or the satellite does not cause this difference to change, except to the extent that movement may cause errors such as multipath to fluctuate. In this way, the offset of the difference between the 308fo implicit carrier phase observable φ and the C/A code group delay observable τ for each Satellites are determined without any unwanted averaging of the time-varying information regarding the position.

Since the series of C/A code delay observations does not have a significant offset, the result of time averaging the difference between the 308 fo implicit carrier phase observable φ and the C/A code group delay observable τ is just the offset of the 308 fo implicit carrier phase observable φ. After this determination, this offset is subtracted from the original series of observations of the 308 fo implicit carrier phase observable φ to obtain a "new" series of observations of the 308 fo implicit carrier phase observable φ which do not have a significant offset, yet retain their original, excellent accuracy, both in position and in time.

The ionosphere, however, increases the group delay and decreases the phase delay of signals received by a satellite if both delays are defined such that a shift of the satellite along the observation line to the receiving antenna would produce equal changes of the same sign in both observables. Therefore, if the difference is taken between the 308-foimplicit carrier phase observable Φ, which is a phase delay observable, and the C/A code group delay observable τ, which is a group delay observable, the effects of the ionosphere do not cancel. In fact, they add together, doubling the effect.

To eliminate the effect of ionospheric effects, simultaneous observations can be made of the phases of other carrier waves that are implicitly included in the received GPS signals. In the preferred embodiment, the other carrier waves, which are observed in part to eliminate ionospheric errors in position determination, are waves with frequencies equal to fo. One such fo carrier is implicitly included in the signals received from each satellite in the L1 band, and one is implicitly included in the L2 band signals received by each satellite. The phases of these carriers indicate the group delays of the signals in the two bands. These two group delays, and hence the respective fo carrier phases, are similarly affected by a shift in position of the satellite along the line of observation from the receiving antenna 22, that is, by a change in the time it takes for the signal to propagate through space. However, they are not similarly affected by the ionosphere.

The group delay indicated by the C/A code group delay observable τ derived from a signal component in the L1 band and the group delay indicated by the L1 band implicit fo carrier phase observable ψL1 are derived from the signals in the same band and have equal contributions from the ionosphere. However, the group delay indicated by the L2 band implicit fo carrier phase observable ψL2 is larger by a factor of (77/60)², namely the square of the ratio of the two band center frequencies. The difference between the group delays in the two bands due to the ionosphere is therefore (2329/3600) times the group delay in the L1 band.

The contribution of the ionosphere to the group delay in the L1 band is determined by taking the difference between the fo phase measurements in the L2 band and the L1 band are multiplied by (3600/ 2329). The fo carrier phase is converted to a group delay by simply dividing the phase by the carrier frequency, which is the same for each band.

The difference between the L2 and the L1 band fo phase measurements has a half-integer uncertainty which needs to be resolved. This uncertainty is easy to resolve because the distance-corresponding uncertainties are large, namely half the wavelength at a frequency of fo, one wavelength at 2 fo, or about 30 meters.

As mentioned above, the difference between the L2 band implicit fo carrier phase observable ΨL2 and the L1 band implicit fo carrier phase observable ΨL1 is sensitive to the ionosphere, but insensitive to the position of the ship 10. The effect of the ionosphere on this difference observable is known a priori to be less than about 15 meters, or less than half a cycle of the phase. Therefore, in practice there is no significant uncertainty in the difference between (L2 band implicit fo carrier phase observable ΨL2) - (L1 band implicit fo carrier phase observable ΨL1). The value of the half-integer offset is zero.

As mentioned above, the effect of the ionosphere on the 308 fo L1 band center frequency carrier phase delay is equal to and opposite to the effect on the L1 band group delay. Therefore, the effect of the ionosphere on the "fundamental" 308 fo carrier phase observations can be detected and removed.

Even if one did not have to take the ionosphere into account, the L1 and L2 band fo carrier phase observations would contribute usefully to the position determination, since each represents a group delay measurement useful for the purpose of determining the offset of the 308 fo carrier phase observations with respect to the C/A code group delay observations using the differencing and averaging procedures described above. In fact, each fo carrier phase observation may be more accurate than the C/A code group delay observation, since the satellite signal bandwidth contributing to each fo carrier phase observation is on the order of 2 fo.

As explained above, the equivalent positional uncertainty of a group delay measurement varies inversely proportional to the bandwidth of the signal contributing to the group delay measurement. The bandwidth, 2 fo, contributing to the fo carrier phase measurements is 10 times the bandwidth of 1 megahertz contributing to the C/A code group delay observation.

A further advantage of the 10 times greater bandwidth of the fo carrier phase observations is that these observations are therefore insensitive to multipath interference from reflected signals whose excess propagation path length, with respect to that of the directly received signal, is greater than about 30 meters, where the effective wavelength corresponds to the bandwidth of 2 fo. In contrast, the C/A code delay measurements are sensitive to reflected signals with excess path lengths of up to about 300 meters, where the effective wavelength corresponds to the bandwidth of about 1 megahertz. Therefore, the fo carrier phase measurements help significantly to reduce the effects of multipath interference on position determination. in particular in the case of a ship 10 in which the antenna 22 may be located at the top of a mast with a height of the order of 30 metres.

The fo carrier phase measurements can be used, as explained above, to determine the offset in the 308 fo carrier phase measurements whether or not the C/A code delay measurements are used and whether or not the delay locked loop comprising the correlator 632 and the C/A code generator 636 is locked by the action of the delay error signals 688 and 692. As explained above, the operation of the correlator 632, which is normally necessary to obtain accurate 308 fo carrier phase measurements from the correlator 632 and the phase register 616, requires that the delay estimate 696 from the delay register 672 be accurate to within less than one piece of the C/A code. However, it should be taken into account in this context that this patch width corresponds to approximately 300 meters of path length, and that position information accurate within the required 300 meters, as explained above, could be obtained from other sources.

The satellite tracking channel 32 includes means for producing measurements of the fo carrier phases relative to an estimate produced by the delay register 672. These measurements of fo carrier phases are fed to the real-time computer 40 in which closed loop tracking is performed by executing a stored program in accordance with standard procedures well known in the art. Preferably, closed loop tracking of the fo carrier phases is performed in the real-time computer 40 rather than by special hardware in the satellite tracking channel 32 since these phases change so slowly.

Another reason it is desirable to produce separate estimates for the phase delay or "phase" and for the group delay or "delay" with separate phase and delay registers is that, for a variety of possible reasons, a valid measurement of one observable but not the other may be possible at a given time. The phase and delay registers 616 and 672 have the ability to "free-run" or lock even in the absence of rate information from the real-time computer 40 due to the action of the phase adder 898 and the carrier frequency or 16-bit frequency register 902 shown in Figure 13.

As explained above, the phase-locked loops and delay-locked loops in the terminal 23 generally attempt to track carrier phases and C/A code delays using error detection and feedback control. Each loop is coupled to the other loops in the sense that its output can help the other loops and also in the sense that it can be helped by the outputs of the other loops. This coupling may be provided in the satellite tracking channel 32 and also in the real-time computer 40 which provides estimates 38 to the satellite tracking channel 32 and receives measurements 42 in response. This coupling may extend beyond the terminal 23, for example to the data processing center 20 which exchanges data with the terminal 23 through the data link 21 as explained with reference to FIG. 1.

The generation of accurate estimates for the phase and delay observables may be assisted by an external information input 44 of ship position and speed information 50, as described with reference to Fig. 2 The vessel position and speed information 50 may include both translation and rotation information, and may include not only estimates of the translation and rotation state of the vessel, but also statistical information related to that state, such as wind and sea conditions. The statistical information may be used to optimize the dynamic tracking performance of a loop. Rotation information may include information regarding the roll, pitch and yaw of the vessel. Such information may be combined in the real-time computer 40 with information regarding the directions of the satellites to predict or estimate occlusion of the line of sight to a satellite, and/or the presence of a reflection of the signal from the satellite by a portion of the vessel 10 or the sea surface.

Returning now to the details of the operation of the tracking loops in the satellite tracking channel 32 of the terminal 23, it should be noted that while the operation of each loop may be assisted by inputs from other sources, each loop may also operate autonomously. For example, the C/A code delay tracking loop in the satellite tracking channel 32 may operate to maintain a delay lock regardless of whether any associated phase tracking loop is locked. As previously explained, the "spin" angular rate of the phase estimate 612 used in the counter rotator 610 must be within plus or minus about 50 hertz of the pointer rotation rate of the received signal in order to allow constructive integration of the signal in the correlator 632 which forms part of the delay tracking loop. However, a sufficiently accurate rotation rate may be conveniently derived from external information calculated without requiring phase locking.

Accordingly, a phase tracking loop can operate to maintain phase lock whether an associated delay-locked loop is locked or not. Even in "open loop" or unlocked states, phase and delay estimates can be maintained with sufficient accuracy using supporting information inputs from internal and external sources, as discussed above.

Each feedback loop is further provided with means for identifying which of its multiple information inputs is likely to be valid and therefore should be used. In some cases, the validity decision is best made by the real-time computer 40 based on applying criteria of "reasonableness" and mutual consistency to the multiple available inputs. However, validity decisions are also made within the satellite tracking channel 32. For example, the correlator 632 may include a comparator, such as the concurrent signal energy indicator 348 shown in Figure 10, in which the received signal strength is compared to a calculated threshold every 0.01 seconds. Only when the signal exceeds the threshold is a phase error signal generated by the correlator 632 applied to the phase register 616 to adjust the phase estimate contained therein accordingly.

The contents of the delay register 672 are adjustable by direct input of a calculated value for the delay from the real-time computer 40, as shown in Fig. 2, and also by the feedback control action of the delay error signals 668 and 670 generated by the correlator 632. and fed back to the delay register 672. The contents of the delay register 672 are also adjusted by the increment and decrement signals 620 and 622, respectively, from the phase register 616. These increment and decrement signals can maintain an accurate delay estimate in the delay register 672 even in the absence of the delay error feedback signals 668 and 670. As mentioned above, the phase register 616 itself continually receives adjustment values derived from both external information and from measurements of the received signal.

The output video signal 630 from counter rotator 610 is applied to the correlator 632 where it is applied to mixers 638, 640 and 642 for multiplication or cross-correlation with the C/A code replica 634 generated by the C/A code generator 636. The C/A code replica 634 is applied directly to mixer 642 and then to mixer 640 after being delayed in time by an amount equal to half the width of a C/A code piece.

The C/A code is the satellite-specific, pseudo-statistical binary code based on which the instantaneous pointer representing the GPS satellite signal is switched between the two points on the circle, which are 180° apart. The code is binary, so that the phase switches between only two values, and the timing pattern of the switching is periodic with a period of 1 millisecond. This can be expressed as the code representing a binary function that has a value of zero when the instantaneous signal pointer is in one position and a value of one when the pointer is in the other position.

The function is a periodic function of time with a period of 1 millisecond.

During the 1 millisecond period, the C/A code function switches back and forth at times, and only at times, that are integer multiples of 1/1023 of a millisecond. Each of the intervals between these times, i.e., each 1023rd of a millisecond, is known as a "slice" of code or a "code slice." In delay circuit 644, C/A code replica 634 is delayed by half a slice to obtain half-slice delayed version 646, which is applied to mixer 640 for correlation with output video signal 630. The output of mixer 640 is integrated for 0.01 seconds by integrator 674 and applied to arctangent generator 678 which produces phase signal 680, a measure of the phase of output video signal 630 with the fast biphasic switching removed from the signal and the signal averaged over 0.01 seconds. Phase signal 680 is the output of the simultaneous correlation channel and is applied to summer 632 along with rate estimate 684 from real-time computer 40.

The channel of the correlator which includes the mixer 640, the integrator 674 and the arctangent generator 678 is referred to as the coincident channel because the half-delayed replica 646 correlated with the output video signal 630 in that channel is intended to be "concurrent" so that the half-delayed replica 646 switches back and forth in synchronism with the desired satellite signal contained in the output video signal 630. The other two of the three coincident channels of the correlator are called the "early" channel and the "late" channel because the C/A code replicas 634 and 650 are earlier and later, respectively.

later than the "simultaneous" half-delayed replica 646. The primary purpose of the early and late channels is to enable detection of an error in the delay estimate 696.

The half-delayed replica 646 is delayed by half in delay circuit 648 to produce the full-delayed replica 650 which is applied to mixer 638. Therefore, in mixers 638, 640 and 642, the output video signal 630 is correlated to the same code sequence except for the half-delay differences in delay. Mixer 638 produces mixer output 652 which is integrated by integrator 654 for 0.01 seconds before being applied to squarer 656. Similar operations are performed on the output of mixer 642 by integrator 660 and squarer 662.

If the delay of the switching function produced in the C/A code generator 636 after being delayed by one-half in the delay circuit 644 exactly matches that of the output video signal 630, or in other words, if the half-delayed replica 646 swings or switches in synchronism with the output video signal 630, then the late correlation output signal 658 is substantially equal to the early correlation output signal 664. Therefore, the early and late channels of the correlator 632 produce equal output signals, except, of course, for a slight variation due to noise.

The late correlation output signal 658 indicates the signal energy caused by the cross-correlation between the output video signal 630 and the one-bit delayed replica 650 during each 0.01 second integration period. Similarly, early correlation output 664 indicates the signal energy captured by the cross-correlation between output video signal 630 and C/A code replica 634. If output video signal 630 undergoes a phase switch simultaneously with "in-time" half-delayed replica 646, then the correlation with full-delayed replica 650 and the correlation with C/A code replica 634 will yield the same value.

On the other hand, if the received signal arrives slightly earlier, the early correlation output 664 is greater than the late correlation output 658. Whether the locally generated code is generated too early or too late with respect to the received signal is determined by comparing the late correlation output 658 with the early correlation output 664 in the comparator 666. The outputs 668 and 670 of the comparator 666 indicate whether the early correlator channel or the late correlator channel had a higher output. Therefore, the output signals 668 and 670 of the comparator 666 indicate whether the C/A code replica 634 should be generated by the C/A code generator 636 earlier or later in order to match the simultaneous, half-delayed replica 646 to the signal in the output video signal 630.

If the late correlation produces a higher output such that the locally generated code appears to be too early and should therefore be delayed to bring it into alignment with the received signal, the comparator output 668 is "true". The comparator output 668 is applied to the delay register 672 and causes the delay estimate 696 stored in the delay register 672 contained in and generated by the same and applied to the C/A code generator 636 to control the delay of code generation.

Increasing the value of delay estimate 696 results in C/A code replica 634 being generated sooner, not later, as might otherwise be inferred from the term "delay." Delay register 672 is referred to as the "delay" register to distinguish its function from that of phase register 616. Through feedback control, the contents of delay register 672 are caused to track the group delay of the C/A code component of the received satellite signal. The term "group delay," commonly used in physics as well as in radio interferometry in geodesy and astronomy, indicates the negative value of the derivative of the phase of the signal with respect to frequency. "Group delay" is also known as "envelope delay" and "modulation delay" in electrical engineering and communications engineering. If a signal arrives later, its group delay is increased. Group delay observables are sometimes referred to as "pseudorange".

It should also be remembered that some observables called "phases", for example the C/A code phase and the fo carrier phase, are basically group delay observables; whereas some observables that are sometimes called "delay" may be basically phase delay observables.

In the preferred embodiment, if the satellite signal arrives earlier, its group delay is increased, but the delay estimate 696 decreases, and therefore also the phase estimate 612 which is contained in and generated by the phase register 616. The signs of the contents of the delay estimate 696 and the delay register 672 are defined such that both contents change in the same direction in response to a change in the distance between the satellite and the antenna 22. As the satellite moves closer, i.e. the distance of the satellite decreases, both the phase estimate and the delay estimate should increase.

Special considerations must be made regarding the special effects of the ionosphere on the propagation of radio signals. At frequencies in the GPS L1 and L2 bands, the ionosphere acts to increase the apparent range of the satellite indicated by the C/A code group delay observable τ, which is tracked by the delay register 672 in cooperation with the correlator 632. At the same time, however, the same ionosphere acts to decrease the apparent range of the satellite indicated by the 308 fo implicit carrier phase observable φ, which is tracked by the phase register 616 in cooperation with the correlator 632. Therefore, the 308 fo implicit carrier phase observable φ and the C/A code group delay observable τ give conflicting indications of the apparent change in distance from the satellite to the receiver caused by the ionosphere.

An increase in ionospheric density therefore causes an increase in the phase estimate 612 as if the satellite had moved closer, and the same increase in ionospheric density causes a decrease in the L1 band implicit carrier phase observable ψL1 and the L2 band implicit carrier phase observables ψL2. These phase observables refer to the group delays, not the phase delays, of the signals received in the L1 and L2 bands.

The group delay for each band, and hence the phase shift of the fo carrier implicitly contained in the signals received in each band, is inversely proportional to the square of the center frequency of each band. A given amount of ionization along the path through the upper atmosphere between the satellite and the receiver therefore causes a reduction in the phase of the L1 fo carrier equal to (77/60)2 times the reduction in the phase of the L2 fo carrier phase. Both fo carrier phases are reduced, whereas the center frequency implicit carrier phase in each band, i.e. the 308 fo carrier phase for the L1 band, is increased.

Further detailed discussion of Figure 14 reveals that if the locally generated code model appears to have too little delay, or switches earlier than the satellite signal in the output video signal 630, the output signal 686 would be true and would cause the contents of the delay register 672 to be reduced. Therefore, the correlator 632, together with the delay register 672, forms a closed feedback loop which maintains a number in the delay register 672 which tracks the variations in the phase switching delay of the received signal and therefore represents a measurement or an estimate of the delay in the output video signal 630.

As explained above, the C/A code function has 1023 pieces. In the early and late channels of the correlator 632, only two possible delays of the code, spaced by one piece, were correlated with the signal. If Of course, if the delay of the signal differs from that contained in delay register 672 by more than one slice, then neither the late channel formed by mixer 638, integrator 654, and squarer 656, nor the early channel formed by mixer 642, integrator 660, and squarer 662 will yield a significant correlation. This situation, in which no correlator channel yields a significant correlation, can arise during initialization if the initial value of delay register 672 does not sufficiently match the current signal delay. If sufficient power is not detected in any of the correlator channels, then delay register 672 must be incremented slightly at any time until all 1023 possible slice delays have been tried, or until a correlation with the signal is established.

The delay locked loop just described provides a measurement of the time delay of the phase switching pattern of the received signal. Another measurement of interest is, as previously described, a measurement of the phase angle of the received signal pointer at specific times. This measurement is performed in an analog manner using feedback so as to cause the contents of the phase register 616 to follow the phase of the signal pointer in the video frequency output signal 608. The phase register 616 contains a "true" number, with a fractional cycle portion and an integer or signed integer portion. The fractional cycle portion is continuously updated and small phase increments, significantly smaller than a full cycle or rotation of the pointer, are added to the fractional cycle portion of the phase register at a rate of 110,000 times per second. A phase feedback 628 is applied to the phase register 616 and represents the sum formed by the summer 682 of the phase signal 680 and the fractional rate related cycle value 684 from the real time computer 40. Even in the absence of a phase error, the fractional rate related cycle value 684 causes the contents of the phase register 616 to vary with time. The fractional rate related cycle value 684 may be derived from a combination of a value contained in a rate register (not shown) provided in the satellite tracking channel 32 and a value from the real time computer 40.

The rate related fractional cycle value 684 is added 110,000 times per second and is calculated according to the information regarding the orbit of the satellite and the positions and movements of the receiving antenna and the satellite and further from the phase signal 680. The amount added to the fractional cycle sub-portion 614 of the phase register 616 110,000 times per second is simply equal to the expected frequency shift of the received signal in hertz due to Doppler shift and other reasons divided by 110,000.

The phase register 616 is constructed to have two parts, a fractional cycle lower part 614, which contains all the bits of the binary phase number up to and including the quarter cycle bit, and the full cycle upper part 618, which contains the more significant bits of the number.

When the lower part of the phase number reaches and exceeds an integer value, the fractional cycle lower part 614 of the phase register 616 generates a carry signal 620 which causes the whole cycle upper part 618 to be incremented by one is incremented. If the expected frequency shift of the received signal is negative, then the amounts added 110,000 times per second to the fractional cycle sub-portion 614 are negative, and in this case the lower portion of the number may decrease until it reaches and exceeds a half-integer value.

In this case, the borrow signal 622 is generated and applied to the whole cycle top portion 618 of the phase register 616 and causes its contents to be reduced by one. The total value of phase contained in the phase register 616 is indicated by the phase number 626, which represents the sum of the fractional cycle bottom portion 614 and the whole cycle top portion 618 formed by the summer 624.

The most significant bit of the fractional cycle lower portion 614, together with the least significant bit of the full cycle upper portion 618 of the phase register 616, forms the phase estimate 612, modulo one cycle and with a resolution or quantization of one quarter cycle, thereby rotating the video frequency output signal 608 counterclockwise in the counter rotator 610. The counter rotator 610 is not required to receive the integer portion of the phase estimate, and the quantization error associated with the coarse quarter cycle resolution of the phase estimate 612 is effectively averaged out since the phase changes rapidly and non-uniformly. It is hoped that the rate at which the contents of the phase register increase or decrease will match that of the selected satellite's component in the video frequency output signal 608 in magnitude to within less than 50 revolutions per second. If this condition is actually fulfilled, then the angle of the pointer rotates by no more than half an angle in 0.01 seconds. revolution, and the signal survives integration in the integrators 654, 660 and 674.

The output video signal 630 is mixed with the "prompt" half-delayed replica 646 in mixer 640 and integrated in integrator 674. The phase angle 680 is calculated by calculating the arc tangent of the imaginary part divided by the real part of the complex signal 676 in arc tangent generator 678. The phase error 680 is applied as a phase error feedback signal to summer 682 which produces phase feedback 628 which is applied to phase register 616 in small increments 110,000 times per second.

Through this phase feedback path, the contents of the phase register 616 are increased or decreased as necessary to bring the phase estimate contained in the phase register 616 into alignment with the pointer of the currently received signal. As the pointer of the received signal rotates forward and/or backward, the angle of the pointer is continuously tracked by the phase register 616 and all revolutions of the pointer around the complex plane are also counted within the total cycle top 618 of the phase register 616.

The 308 fo carrier phase of the received signal provides a very accurate and very fast measure of the distance or path length from the satellite to the receiver. Any increase in path length is immediately reflected, within about 0.01 seconds, in a decrease in the contents of phase register 616. For each movement of the satellite by one 308 fo carrier wavelength, the phase of the received carrier signal changes by one cycle of phase and the contents of phase register 616 changes by one unit.

Of course, fractions of a wavelength of the movement are also detected.

The phase number 626 is therefore an indication or measurement of the distance to the satellite, despite the offset explained above. The phase in cycles is therefore equal to the distance in wavelengths plus an offset. The offset occurs due to the unavoidable fact that when the device is first put into operation, the correct initial value of the phase number 626 is not known. Any changes in the received signal phase are faithfully reflected by changes in this number, but the initial value is arbitrary or meaningless.

Although the phase is offset as a measure of the distance from the satellite to the receiver, the offset remains constant throughout the time the satellite signal is being tracked by the phase register 616 unless an interruption in tracking occurs. During such an interruption, the number of cycles of phase that would have been received had the signal not been interrupted may not be correctly known or may not be correctly estimated, so that after the interruption, the contents of the phase register 616 may have a different offset. However, the difference in offset must be an integer number of cycles. If the interruption is short, there is a good chance that this integer change will be zero, since the increments, 110,000 times per second, of the phase feedback 628 received during the interruption may have increased the phase estimate at a rate very close to the current phase rate of the signal. In other words, the phase register 616 performs a freewheel or coupling in an exact manner, whereby the error increases by less than one cycle if the duration of the failure is short.

To minimize the likelihood that the phase offset will change, the rate-related fractional cycle value 684 applied to summer 682 is calculated at any time from the best instantaneous information regarding the motion of the satellite and ship 10, and therefore the Doppler shift and other shifts that the received signal might exhibit. During a brief interruption of the signal, these increments of summer 682 represent very accurate predictions of the increments that phase register 616 should actually have in order to maintain phase alignment of phase register 616 with the received signal.

The delay locked loop formed by the correlator 632 and the delay register 672 provides an indication or measurement of the range to the satellite antenna 22 since changes in range are reflected not only in the phase register 616 but also in the delay register 672. As the satellite moves farther away, the signal propagation time is increased and the phase variations arrive later. This delay is immediately detected by an increase in the late correlation output 658, or indicated by the late correlation output 658 becoming greater than the early correlation output 664. As long as this disparity exists, the comparator 666 would cause the delay register 672 to be increased or decreased until finally the delay estimate 696 has caused the code to be generated by the C/A code generator 636 to be generated with the correct delay. However, it should be noted that the delay register 672 may also received increment or decrement input signals via carry signal 620 or borrow signal 622 from phase register 616, and that in the absence of any effects such as ionospheric dispersion which cause the group delay to vary differently than the phase delay, signals 620 and 622 would have inherently maintained a correct estimate of the delay in delay register 672.

As explained above, it is possible to determine position directly from the phase number 626 of the phase register 616. It is also possible to determine position by the C/A code delay measurement 698 of the delay register 672. As already mentioned, an advantage of the phase number 626 is that it is very accurate: even a very small movement on the order of a centimeter can be detected, since the wavelength of the signal is about 19 centimeters, so a movement of 1 centimeter would cause a phase change of 1/20th of a cycle, which can be detected.

However, as also explained above, these phase measurements are offset by an unknown amount, whereas the C/A code delay measurements 698 from the delay register 672 do not suffer from such an offset.

As explained above, the L1 band implicit fo carrier phase observable ψL1 and the L2 band implicit fo carrier phase observable ψL2 implicitly contained in the GPS signals 15 are measured and their difference is used to determine and calculate the effects of the ionosphere from the measurements of the 308 fo implicit carrier phase observable φ and the C/A code group delay observable τ. The codeless dual-band phase detector 751 shown in Fig. 14 gives measurements of the L1-band implicit-fo-carrier phase observables ψL1 and the L2-band implicit-fo-carrier phase observables ψL2.

As previously explained, the GPS signals 15 from the GPS satellite 12 are received by the antenna 22 and applied via the transmission line 600 to the mixer 602 for correlation with the 308 fo local model 604 from the local oscillator 606 to produce the video frequency output signal 608. In addition, the signals in the transmission line 600 are also applied to the mixer 605 for correlation with the 240 fo local model 724 from the L2 band local oscillator 726 to produce the L2 band video output signal 722. The video frequency output signal 608 and the L2 band video output signal 722 represent equivalent correlation products, except that they have been correlated with replicas of L1 and L2 band center frequency carriers, respectively.

The codeless dual-band phase detector 751 receives the video frequency output signal 608 and the L2 band video output signal 722 from mixers 602 and 605, respectively. Mixer 605 is like mixer 602 and downconverts the L2 band portion of transmission line 600 to the video frequency band by mixing transmission line 600 with the 240 fo local model 724 from L2 band local oscillator 726.

Within the codeless dual-band phase detector 751, the video frequency output signal 608 is applied to the L1-band fo carrier reconstructor 728, which produces the L1-reconstructed carrier composite signal 738. The L2-band video output signal 722 is applied to the L2-band fo carrier reconstructor 750, which produces the L2-reconstructed carrier composite signal 756. The L1-band fo carrier reconstructor 728 and the L2 band fo carrier reconstruction device 750 are exactly the same, the only difference being that one receives an input signal and produces an output signal related to signals received in the L1 band and the other receives an input signal and produces an output signal related to signals received in the L2 band.

The L1 reconstructed carrier composite signal 738 is applied to the phasor counter rotator 740 to produce the output signal 744 which is integrated for 100 seconds by the 100 second integrator 746. The output of the 100 second integrator 746 is applied to the arctangent generator 748 which produces the L1 fo residual phase measurement 718.

Accordingly, the L2 reconstructed carrier composite signal 756 is applied to the pointer counter rotator 758, the output of which is integrated for 100 seconds by the 100-second integrator 760. The output of the 100-second integrator 760 is applied to the arctangent generator 762, the output of which represents the L2 fo residual phase measurement 720.

The L1 reconstructed carrier composite signal 738 from the L1 bandfo carrier reconstructor 728 is a composite signal of continuous wave signal components with frequencies near 2 fo. In the L1 reconstructed carrier composite signal 738, a continuous wave component represents the reconstructed carrier wave implicitly contained in the signals received from each satellite in the L1 band. The frequency of each continuous wave, or reconstructed carrier component, is exactly twice that of the respective fo carrier which is implicitly contained in the received signals. The phase of each reconstructed carrier is also doubled. The frequency of each reconstructed carrier is offset from 2 fo, mainly due to the Doppler shift of the received signal from the satellite, by an amount which varies with time between plus and minus 30 Hertz.

The reconstructed carrier component from the particular satellite to which the satellite tracking channel 32 has been assigned is selected from the L1 reconstructed carrier composite signal 738 by the phase counter rotator 740 and the 100 second integrator 746. The basis for this selection is provided by the fo carrier phase estimate 742 generated by the delay register 672 and applied to the phasor counter rotator 740. The frequency, i.e., the time derivative, of the fo carrier phase estimate 742 is extremely close, within less than about 0.005 hertz, to the frequency of the reconstructed carrier component from the selected satellite within the L1 reconstructed carrier composite signal 738. The counter representing the reconstructed carrier component related to the selected satellite in the output signal 744 from the pointer counter rotator 740 therefore rotates less than one-half cycle during the 100 second period of each integration performed by the 100 second integrator 746. The reconstructed carrier component for the satellite associated with the satellite tracking channel 32 therefore accumulates coherently during the integration, but the reconstructed components for all other satellites do not. The reconstructed carrier components from other satellites do not accumulate coherently, so they will zero out during the integration performed by the 100-second integrator 746. because their phasor components in the output signal 744 from the phasor counter rotator 740 rotate one or more cycles during the 100 second integration time. The phasors of the other satellites rotate because their reconstructed carrier frequencies are generally not equal to within 0.01 hertz of the frequency of the fo carrier phase estimate 742.

Therefore, after completion of each 100 second integration, the arctangent generator 748 produces the L1 fo residual phase measurement 718, which is the four-quadrant inverse tangent function of the output of the 100 second integrator 746. The L1 fo residual phase measurement 718 therefore represents an exact measurement of the L1 band implicit fo carrier phase observable ψL1 for the selected satellite with respect to the fo carrier phase estimate 742.

The fo carrier phase estimate 742 is applied to both phasor counter rotators 740, which otherwise operate in the same manner for L1 and L2 band signals, respectively.

Accordingly, the 100-second integrator 760 and the arctangent generator 762 process the L2-band signals from the phasor counter rotator 758 to produce the L2 fo residual phase measurement 720 in the same manner as the 100-second integrator 746 and the arctangent generator 748 process the L1-band signals from the phasor counter rotator 740 to produce the L1 fo residual phase measurement 718, which represents an exact measurement of the L2-band implicit fo carrier phase observable ψL2 for the selected satellite with respect to the fo carrier phase estimate 742.

The satellite tracking channel 32 provides the C/A code delay measurement 698, the phase number 626, the L1-fo residual phase measurement 718 and the L2-fo residual phase measurement 720 for the observable combiner 701, which is described below with reference to Fig. 15.

Fig. 15

Two embodiments of the observable combiner 701 are shown in Fig. 15. A simplified embodiment of the observable combiner 701 is shown in Fig. 15A to facilitate description of its operation. The alternative embodiment of the observable combiner 701 shown in Fig. 15B is formed by a Kalman filter according to well-known techniques in the art. The embodiment shown in Fig. 15B is advantageous because it makes greater use of the information content of the observables.

In the embodiment shown in Fig. 15A, the observable combiner 701 receives a measurement of the C/A code group delay observable τ, for example, the 30-bit C/A code group delay estimate τest 876 from the delay register 842 shown in Fig. 13, or the C/A code delay measurement 698 from the delay register 672 shown in Fig. 14; a measurement of the 308-fo implicit carrier phase observable φ, for example, the 48-bit binary 308-fo carrier phase estimate φest 856 from the phase register 840 shown in Fig. 13, or the phase number 626 from the phase register 616 shown in Fig. 14; a measurement of the L1 band implicit fo carrier phase observable ΨL1, for example the L1 fo carrier phase measurement 210 from the L1 band fo carrier phase detector 206 shown in Fig. 6 or the L1 fo residual phase measurement 718 from the codeless dual band phase detector 751 of Fig. 14; and a measurement of the L2 band implicit fo carrier phase observable ΨL2, for example, the L2 fo carrier phase measurement 212 from the L2 band fo carrier phase detector 200 shown in Fig. 6, or the L2 fo residual phase measurement 720 from the codeless dual band phase detector 751 shown in Fig. 14.

As explained above, the measurement of the C/A code group delay observable τ is unique. This means that the integer millisecond uncertainty which can potentially affect a C/A code delay measurement has been resolved, for example as explained above with respect to the delay register 842 and the group delay initial value 37 shown in Figure 13. In this sense, therefore, the C/A code group delay observable τ is not significantly offset. However, it is affected by statistical fluctuations with a mean value close to zero of the order of 15 metres due to noise and multipath interference, and by an error, typically of the order of 0 to 30 metres, due to refraction in the ionosphere along the signal propagation path from the GPS satellite 12 to the antenna 22.

In fact, since the contribution of the ionosphere to the measurement of the C/A code group delay observable τ can only have one sign, it can be said that the C/A code group delay observable τ does indeed have an offset. On the other hand, a correction of an expected value for the ionosphere effect could be applied to the measurement of the C/A code group delay observable τ, for example by an estimate from external information used in the real-time computer 40 or by some other means not shown, so that the uncorrected remaining effect has an average value of zero when considered statistically. Methods for producing an expected value for the ionospheric effect on a C/A code delay measurement are known. In fact, the GPS transmit navigation data includes data from which such a value can be produced. However, this data is generally not very accurate and may not be available. Therefore, it is desirable to detect and remove the ionospheric effect by using as a basis the observed phase and/or delay characteristics of the signals received from a satellite.

As explained with reference to Fig. 14, the measurement of the 308 fo implicit carrier phase observable φ represents an offset measurement of range with an uncertainty of an integer or half an integer cycle, corresponding to an uncertainty of an integer or half an integer wavelength, which is difficult to resolve. However, the measurement of the 308 fo implicit carrier phase observable φ represents a very accurate indicator of range changes. The measurement of the 308 fo implicit carrier phase observable φ involves very little noise and multipath errors, compared to the measurement of the C/A code group delay observable τ. However, the error in the measurement of the 308 fo implicit carrier phase observable φ is due to the atmosphere is exactly the same magnitude as the error in the measurement of the C/A code group delay observable τ due to the ionosphere, if both measurements are expressed in equivalent units, for example in units of distance. This error is typically of the order of a distance of 5 to 30 meters.

As explained above, in the preferred embodiment, the signs of the measurement of both the The C/A code group delay observables τ and 308 fo implicit carrier phase observables φ are defined such that a shift of the satellite along the observation line toward the antenna 22 produces an increase in each of these measurements. Therefore, the effects of the ionosphere in these measurements were of opposite signs. Therefore, the ionospheric effects on the propagation of GPS signals 15 might typically be on the order of an equivalent distance of 20 meters, which would appear as an increase in group delay in a measurement of the C/A code group delay observables τ, whereas the same effect would appear as a decrease in phase delay in a measurement of the 308 fo implicit carrier phase observables φ.

For example, as shown in Figure 14, the signs of the L1 fo residual phase measurement 718 and the L2 fo residual phase measurement 720 are also defined such that each of these measurements would also increase if the satellite moved closer, currently assuming that the fo carrier phase estimate 742 has not changed. Since these subcarrier phase measurements are actually group delay measurements rather than phase delay measurements, such changes would have the same sign as the changes in the C/A code group delay observables τ (meaning they would appear as increased values) rather than having the sign of the changes in the 308 fo implicit carrier phase observables φ. Of course, the usual response of the fo carrier phase estimate 742 to a movement of the satellite along the observation line is for the value to increase, so that normally neither the L1 fo residual phase measurement 718 nor the L2 fo residual phase measurement 720 would change.

As shown in Figure 15A, the measurement of the C/A code group delay observable τ is applied to the scaling converter 920, which accepts the C/A code group delay observable τ in the numerical representation in which it was generated and generates the delay range 922, which is a number representing the C/A code group delay observable τ in units of distance.

The measurement of the 308 fo implicit carrier phase observable Φ is applied to the scaling converter 924, which accepts the measurement of the 308 fo implicit carrier phase observable Φ in the numerical representation in which it was generated and generates the phase range 926, which is a number representing the 308 fo implicit carrier phase observable Φ in the same distance units as the delay distance 922.

Accordingly, the measurements of the L1 band implicit fo carrier phase observable ψL1 and the L2 band implicit fo carrier phase observable ψL2 are applied to a scaling converter 928 and 932, respectively, which produce the L1 residual distance 930 and the L2 residual distance 934, respectively, in the same distance units as the phase distance 926 and the delay distance 922.

The phase distance 926 and the delay distance 922 are applied to the L1 distance subtractor 936, which subtracts the delay distance 922 from the phase distance 926 to produce the L1 distance difference 938, which in turn is applied to the averaging device 940. The averaging device 940 produces the offset estimate 942 as an average of the L1 distance difference 938 over a period of time. The purpose of this temporal averaging is to compensate for the fluctuations in the L1 range difference 938 that occur due to noise and multipath errors so that the offset estimate 942 is a good estimate of the offset present in the phase range 926. The length of the averaging period may be the entire time period over which continuous observations are available.

On the other hand, it may be preferable to use a shorter averaging time, for example to limit the maximum possible time period during which the unoffset exact range 936 could be affected by a random "cycle slip" or by a step change in the offset of the phase number 626 due to a momentary loss of phase tracking in the satellite tracking channel 32.

The phase distance 926 and the offset estimate 942 are applied to the offset subtractor 944, which subtracts the offset estimate 942 from the phase distance 926 to produce the exact, unoffset distance 935, which therefore represents a measurement of the distance having the spatial and temporal exactness of the phase distance 926 and freedom from uncertainty or offset of the delay distance 922. It is noted that both the phase distance 926 and the delay distance 922 contribute to the exact, unoffset distance 946.

In one sense, the terminal 23 can be said to derive position information from observations of the C/A code group delay observable T with the aid of observations of the 308-fo implicit carrier phase observable Φ. In another, equally valid sense, the terminal 23 can be said to derive position information from observations of the 6, and as can also be seen from the interconnection of the phase register 616 and the delay register 672 in FIG. 14, the 308 fo implicit carrier phase observable Φ and the C/A code group delay observable δ are derived from observations of the C/A code group delay observable δ. In fact, the terminal 23 receives position information from a multiple combination of the C/A code group delay observable δ and the 308 fo implicit carrier phase observable Φ, both during measurement and after measurement in the satellite tracking channel 32. Within the phase and group delay register subsystem 196 shown in FIG. 6, and as can also be seen from the interconnection of the phase register 616 and the delay register 672 in FIG. 14, the 308 fo implicit carrier phase observable Φ and the C/A code group delay observable δ are measured interactively within the satellite tracking channel 32. Each measurement of one helps the measurement of the other. This interaction during the measurement of the 308 fo implicit carrier phase observables φ and the C/A code group delay observables τ is different from and separable from the interaction during the post-measurement combination of these observables performed in the codeless dual-band phase detector 751.

In fact, this interactive acquisition can be viewed as a measurement of a new and different observable, namely the group phase delay observable. Measurements of the group phase delay observables combined as shown above determine a new type of ranging distance between the satellite and the receiver. The process can be referred to as a "macro-ranging" measurement, and the range so measured can be referred to as a "macro-ranging" distance.

Only the change, and not the mean or constant component of the phase range 926 during the observation period contributes to the exact distance 946 without offset. The mean or constant component of the exact range 946 without offset is determined by the delay distance 922. Therefore, the terminal 23 can be said to derive position information from observations of the C/A code group delay observable τ.

However, the measurement of the 308 fo implicit carrier phase observable Φ applied to the offset subtractor 944 increases the measurement of the phase distance 926 by removing its offset, which if not removed would then represent an error. The exact distance 946 without offset, which is the output of the offset subtractor 944, is the phase distance 926 with reduced offset error. The delay distance 922 contributes to increasing the phase distance 926 by helping to reduce its offset error. Therefore, the terminal 23 can also be said to obtain position information from measurements of the 308 fo implicit carrier phase observable Φ. using measurements of the C/A code group delay observable τ.

The exact range 946 without offset is affected by ionospheric errors due to the ionospheric errors contained in both the delay distance 922 and the phase distance 926. To correct the exact range 946 without offset, it is combined with a correction derived from the L1 residual distance 930 and the L2 residual distance 934, as explained below.

The L1 residual distance 930 and the L2 residual distance 934 are applied to the L1-L2 subtractor 948, which subtracts the L2 residual distance 934 from the L1 residual distance 930 to produce the L1-L2 difference 950, which measures the difference in the ionospheric delays of the GPS signals, received in the two bands. The L1-L2 difference 950 is time averaged or filtered in the L1-L2 difference averager 952 to produce the ionospheric estimate 954, which is a smoothed estimate of the effect of the ionosphere on the signals. The reason for the averaging or filtering performed in the L1-L2 difference averager 952 is to improve the estimate by smoothing out any rapid fluctuations that may occur in the L1 residual range 930 or the L2 residual range 934 due to noise or multipath and the like. Typically, such fluctuations are faster than actual ionospheric fluctuations.

However, the characteristic time scale of multipath variations may vary depending on environmental factors such as antenna height, horizontal distance to a reflector, sea conditions, ship speed, etc. Therefore, the averaging time or filter characteristics used in the L1-L2 difference averaging device 952 should be adapted to the current situation. Faster changes in the relative position of the antenna 22 and possible reflectors such as the sea surface, and equally rapid changes in the position of the antenna 22 with respect to the satellite caused, for example, by rough seas, reduce the coherence of multipath errors and allow integration periods as short as, for example, one minute. Less rapid changes may require longer integration periods on the order of several minutes. Certain ionospheric changes, known as ionospheric propagation disturbances, are quasi-periodic with quasi-periods on the order of half an hour to three quarters of an hour. The length of the Integration periods should therefore typically be less than 10 minutes.

The ionosphere estimate 954 is scaled by multiplying it by a factor of 3600/2329 in the scaling device 956. The product of this operation is the L1 ionosphere estimate 958, which is an estimate of the ionosphere contribution to the exact range 946 without any offset. The L1 ionosphere estimate 958 is subtracted from the exact range 946 without offset by the ionosphere subtraction device 960 to produce the vacuum range 962 in which the effect of the ionosphere on the exact range 946 without offset has been substantially removed, leaving the vacuum range 962 which maintains the temporal and most of the spatial accuracy of the phase range 926 and the freedom from uncertainty or offset of the delay range 922.

In Fig. 15B, the alternative embodiment of the observable combiner 701 is shown. This embodiment is formed from a conventional Kalman filter, or a sequential minimum squares estimator, in accordance with well-known techniques.

As shown in Fig. 15B, the Kalman filter 703 in the combiner 701 receives measurements of the C/A code group delay observable τ, the 308 fo implicit carrier phase observable φ, the L1 band implicit fo carrier phase observable ψL1, and the L2 band implicit fo carrier phase observable ψL2 from all of the satellite tracking channels 32, and associated time lag information, such as the real-time digital display 34 from the Fig. 2. Furthermore, the Kalman filter 703 receives external information 707, which may include orbit and time correction information 46, phase and delay observations from shore stations 48, and ship position and speed information 50, as shown in Fig. 2.

In the Kalman filter 703, the preferred values for the standard deviations of the input measurement errors, expressed in distance units, are on the order of 10 meters for the C/A code group delay observables τ, of 1 meter for the L1 band implicit fo carrier phase observables ψL1, and the L2 band implicit fo carrier phase observables ψL2, and of 3 millimeters for the 308 fo implicit carrier phase observable φ.

The Kalman filter 703 produces position information 705, which may include the instantaneous estimates and the estimated covariances of the associated estimation errors of the variables listed in the first column of the table shown in Fig. 15C. The variables and associated information shown in Fig. 15C are typical of those suitable for determining the position of a medium-sized vessel, such as a seismographic survey vessel, the terminal 23 of the vessel being equipped with a clock controlled by a frequency standard of the commercially available rubidium vapor type.

In the Kalman filter 703, each estimator variable is modeled by a first-order Gauss-Markov process, an integrated Gauss-Markov process, or the sum of any two of these processes. As is known to those skilled in the art of filtering and estimation, the Auto-covariance of a first-order Gauss-Markov process is an exponential function of the time lag, and can be conveniently characterized by two parameters: the square root of the variance for zero lag, SIGMA; and the 1/e lag or correlation time, T. Preferred values for these two model parameters are shown for each variable in the second and third columns of the table presented in Fig. 15C. Note that in some cases the preferred correlation time T is considerably longer than any possible observation period. In such cases, the Gauss-Markov process resembles a Monte Carlo process.

It is noted that different sets of state variables, model and/or statistical parameterizations may be more appropriate in other situations. The set presented in the table is almost a minimum set in the sense that a more complicated parameterization may be desirable. For example, the wave-induced motion of the ship 10 could be modeled using damped harmonic oscillators. It may also be desirable to introduce variables representing the satellite orbit and clock parameters.

The first three variables, latitude, longitude and altitude, describe the short-term mean position of antenna 22 with the wave-induced motion removed. The next two variables are the time derivatives of latitude and altitude, respectively. The latitude and altitude of vessel 10 are modeled only by the integrals of the respective derivative processes, given in the table. The altitude of vessel 10 is simply modeled by the first order process with the properties listed after "altitude".

The instantaneous position of antenna 22 is described by the sums of the short-term averages of the latitude, longitude, and altitude variables, plus the respective offsets delta-latitude, delta-longitude, and delta-altitude. It may be desirable to introduce such "delta" or position offset variables to model the wave-induced motion of vessel 10. Of course, these offset variables tend to track or absorb any short-term motion of antenna 22.

The epoch error of the real-time clock 36 as a function of time is modeled by the sum of the "clock epoch" process and the integral of the "clock rate" process, which are given in the table.

The offset for the 308-fo implicit carrier phase observable Φ for each observed satellite is explicitly modeled, by a procedure equivalent to a very slow Monte Carlo approach.

The ionospheric range effect in the 308-fo implicit carrier phase observable φ and the C/A code group delay observable τ is modeled in a standard way, as the product of a local "zenith" value and a zenith angle dependent "mapping factor" which is a function of the local zenith angle of each satellite. The Gauss-Markov parameters given in the table refer to the zenith value.

As explained above, the a priori uncertainty of the ionospheric parameter can be so small that it is not necessary to estimate offset parameters related to the L1-band implicit fo carrier phase observable ΨL1 and the L2-band implicit fo carrier phase observable ΨL2 However, offset parameters for these observables may be included in the set of estimated state variables for each satellite. Very quickly after the cessation of observations of a particular satellite, the estimates for these offsets should converge to the correct half-integer values, usually to zero. Once such convergence has been achieved, an offset parameter can be removed from the variable set and set to the appropriate half-integer value.

The Kalman filter 703 preferably includes means for monitoring the validity of its input data. Cycle slip in phase tracking, which appears as half-integer or whole-integer step changes in the phase measurements, can be detected by known methods. The rms values of the differences between the observed and the model-calculated values of the observables can also be monitored and checked against a priori assumptions. Any individual observation that differs by more than a small multiple of the appropriate rms value should be rejected.

Claims (46)

1. Method for deriving position information from a signal with Doppler shift, spectrum expansion and carrier suppression (15) received (22, 24) by a satellite (12-14) orbiting the earth, wherein the signal includes a first spectrum-expanded and carrier-suppressed component modulated by a first spectrum-expanding code that is known, and a second spectrum-expanded and carrier-suppressed component that is not modulated by the first code but is modulated by a second spectrum-expanding code that has a code sequence that may be unknown, the method comprising deriving position information by correlating (226, 632) the first known code (228, 634) with the first component, characterized in that the method comprises: (a) selection (146, 148, 730, 752) of the second component; (b) reconstructing a carrier (150, 728, 750) from the second component independently of the code sequence of the second code; (c) deriving an estimate (204, 742) of the Doppler shift of the carrier from the correlation of the first code; (d) using (740, 758) the estimated value to determine the phase of the carrier (206-212, 718, 720); and (e) Improving the position information by combining the phase with it (701). 2. Method according to claim 1, which comprises selecting the second component by means of a filter (146, 148) adapted to the spectrum of the second code. 3. Method according to claim 1 or 2, which comprises reconstructing (150) the carrier by doubling the phase of the component. 4. The method of claim 3, wherein the phase doubling is carried out by correlating the component with itself. 5. The method of claim 4, wherein the component is correlated with itself by dividing it into a plurality of portions (142, 144) and mixing one of these portions with another portion. 6. The method of claim 5, wherein the components are spectral components. 7. The method according to claim 6, wherein the spectral components are side bands. 8. The method of claim 1, 2 or 3, comprising selecting a plurality of signal components (142, 144) modulated by the second code and reconstructing the carrier by correlating (150) one of the plurality of signal components with another. 9. The method of claim 8, wherein the plurality of signal components are different spectral components. 10. A method according to any one of claims 1 to 9, comprising selecting a plurality of signal components, each modulated by the second code; Reconstructing a plurality of carriers from the plurality of signal components independently of the second code; estimating the phase of each of the plurality of carriers; and Combining the phase estimates of the plurality of carriers to improve the position information. 11. A method according to any preceding claim, wherein the satellite is one of a plurality of satellites simultaneously transmitting signals with overlapping, extended spectra. 12. The method of claim 11, wherein the satellite is a NAVSTAR GPS satellite. 13. The method of claim 12, wherein the first code is a C/A code of the NAVSTAR GPS. 14. The method of claim 12 or 13, wherein the second code is a P-code of the NAVSTAR GPS. 15. A method according to any preceding claim, wherein information about the second code is not accessible, and the method enables the derivation of improved position information from the signal component despite the inaccessibility. 16. A method according to any preceding claim, wherein the phase is used to reduce the ionospheric error in the position information. 17. A method according to any preceding claim, wherein the phase is used to reduce the multipath error in the position information. 18. Method according to one of the preceding claims, wherein the phase is used to reduce the systematic error in the position information. 19. A method according to any preceding claim, wherein the phase is used to reduce ambiguity in the position information. 20. A method according to any preceding claim, wherein the position information includes an estimate of the distance from the satellite to a point at which the signal is received. 21. A method according to claim 20, comprising: Estimating the phase delay of the signal; Estimating the group delay of the signal; and Combine the phase and group delay estimates to derive the range estimate. 22. The method of claim 21, wherein the point is moving and the combination of phase and group delay estimates are time averaged to derive an estimate of the instantaneous range. 23. A method according to any preceding claim, wherein the carrier is a subcarrier and the phase represents modulation delay. 24. Apparatus for deriving position information from a Doppler-shifted, spectrum-expanded, carrier-suppressed signal (15) received (22, 24) from an Earth-orbiting satellite (12-14), the signal including a first spectrum-expanded, carrier-suppressed component modulated by a first spectrum-expanding code that is known, and a second spectrum-expanded, carrier-suppressed component that is not modulated by the first code but is modulated by a second spectrum-expanding code that has a code sequence that may be unknown; the apparatus comprising means (226, 632) for correlating the first code for use in deriving position information by correlating the first known code (228, 634) with the first component, characterized in that the apparatus comprises: (a) selection means (146, 148, 730, 752) responsive to the signal for selecting the second component; (b) reconstruction means (150, 728, 750) responsive to the second component and reconstructing a carrier therefrom, independent of the code sequence of the second code; (c) Doppler estimation means (632, 616, 672) responsive to the means for correlating the first code and deriving an estimate (204, 742) of the Doppler shift of the carrier; (d) phase determining means (740, 758) responsive to the estimated value and determining the phase of the carrier (206-212, 751, 718, 720); and (e) means (701) for combining the phase with the position information to improve the position information. 25. Apparatus according to claim 24, comprising filter means (146, 148) responsive to the signal, adapted to the spectrum of the second code and serving to select the signal component. 26. Apparatus according to claim 24 or 25, comprising a phase doubling device (150) responsive to the signal component and reconstructing the carrier by doubling the phase of the component. 27. Apparatus according to claim 26, wherein the phase doubling means comprises an autocorrelation means for correlating the component with itself. 28. Apparatus according to claim 27, wherein the autocorrelation means comprises means for dividing the component into a plurality of portions (142, 144) and means for mixing the one portion with the other portion. 29. The device of claim 28, wherein the components are spectral components. 30. Device according to claim 29, wherein the spectral components are side bands. 31. Device according to one of claims 24 to 26, which comprises a device for selecting a plurality of signal components modulated by the second code, and means for reconstructing the carrier by correlating one of the plurality of signal components with another. 32. The apparatus of claim 31, wherein the plurality of signal components are different spectral components. 33. Device according to one of claims 24 to 30, which comprises means for selecting a plurality of signal components each modulated by the second code; means for reconstructing a plurality of carriers from the plurality of signal components independently of the second code; and means for estimating the phase of each of the plurality of carriers; and Means for combining the phase estimates of the plurality of carriers to improve the position information. 34. Apparatus according to any one of claims 24 to 33, wherein the satellite is one of a plurality of satellites simultaneously transmitting signals with overlapping, extended spectra. 35. Apparatus according to any one of claims 24 to 34, wherein the satellite is a NAVSTAR GPS satellite. 36. The apparatus of claim 35, wherein the first code is a C/A code of the NAVSTAR GPS. 37. Apparatus according to claim 35 or 36, wherein the second code is a P-code of the NAVSTAR GPS. 38. Apparatus according to any one of claims 24 to 37, wherein information about the second code is not accessible, and the apparatus enables the derivation of improved position information from the signal component despite the inaccessibility. 39. Apparatus according to any one of claims 24 to 38, comprising means for using phase to reduce ionospheric error in position information. 40. Apparatus according to any one of claims 24 to 39, comprising means for using the phase to reduce the multipath error in the position information. 41. Apparatus according to any one of claims 24 to 40, comprising means for using the phase to reduce the systematic error in the position information. 42. Apparatus according to claims 24 to 41, comprising means for using the phase to reduce ambiguity in the position information. 43. Apparatus according to any one of claims 24 to 42, comprising means for estimating the distance from the satellite to a point at which the signal is received. 44. The apparatus of claim 43, further comprising: Means for estimating the phase delay of the signal; Means for estimating the group delay of the signal; and a combination device for combining the phase and group delay estimates to derive the distance estimate. 45. Apparatus according to claim 44, wherein the point is moving and the combining means comprises time averaging means to derive an estimate of the instantaneous range to the moving point. 46. Device according to one of the preceding claims, wherein the carrier is a subcarrier and the phase represents modulation delay.