DE2907085A1 - METHOD AND ARRANGEMENT FOR PROCESSING PHASE-MODULATED SIGNALS - Google Patents

Description

from Schlumberger Technology Corporation, 5ooo Gulf Freeway, Houston / Texas, Ό3Ά,

concerning:
"Procedure and arrangement for processing phase-modulated signals"

The invention relates to communication systems and more particularly to improved methods and arrangements for receiving and Interpräation of data signals femgemessen in a system and have been transferred to the earth surface from a wellbore, where the measurements were not obtained during drilling. In particular, the invention relates to methods and arrangements for processing phase modulated signals.

Carrying out borehole investigations during the drilling process itself includes the transmission of measurements obtained underground to the surface of the earth, the measurements generally being made by instruments placed immediately behind the drill head. The possibility, continuously with information during the drilling of a borehole Obviously, getting the entire drill string up and running is tempting. Nevertheless, such systems have not yet found widespread use, in particular because of the problems involved in the transmission of the measurement information viaren connected to the earth's surface through the noisy and narrow borehole. Various schemes have been proposed for performing the Transmission of measurement information to the earth surface. According to a suggestion for example, the readings are to be transmitted by means of insulated electrical conductors extending through the drill string. This scheme

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however, requires adaptation of the drill string including the provision of electrical connections to the drill pipe couplings. Another proposed scheme uses an acoustic wave traveling up through the metallic drill string, but the apparently high levels of interfering noise in a drill string are problematic. Another scheme, which appears particularly promising, uses a drilling fluid within the borehole as a transmission medium for acoustic waves that are modulated with the measurement information. Typically, drilling fluid is circulated down through the drill string and drill head and up through the annulus delimited by the portion of the borehole which encloses the drill string. This is conventionally done in order to remove the debris and maintain a desired hydrostatic pressure in the borehole. In the technique described, a subterranean acoustic transmitter known as a rotary valve or "drilling fluid siime" repeatedly interrupts the flow of drilling fluid and this results in an acoustic carrier signal generated in the drilling fluid at a frequency that depends on the Rate of interruptions. The acoustic carrier is modulated as a function of the digital measured value data. In a phase modulation technique known as "phase shift keying" or PSK ₇ , the acoustic carrier is modulated between two or more phase states. A wide variety of coding schemes are possible using PSK modulation. In a "not back to zero" coding scheme, a change in phase represents one binary status (e.g., a logical 1), while the absence of a phase change represents the other binary status (e.g., a logical 0). The phase changes are achieved mechanically by temporarily modifying the drilling fluid siren interrupt frequency to a higher or lower frequency until a desired phase lag (or phase lead) is achieved, after which the drilling fluid siren returns to its nominal frequency. For example, if the nominal frequency of the drilling fluid siren is 12 Hz, a phase change of 18o can be achieved by temporarily lowering the frequency of the drilling fluid siren to 8 Hz for 125 milliseconds (which corresponds to a period at 8 Hz and 1 1/2 periods at 12 Hz), after which the drilling fluid siren frequency restored to 12 Hz

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will. One can easily see that an 18o phase shift is also achieved could be achieved by temporarily increasing the drilling fluid siren frequency during a corresponding period of time (i.e. until a desired Phase lead), after which one returns to the nominal frequency.

In confessional (PSK) communications, the bearer phase is commonly used changed in alternating directions (i.e. alternating phase lead and phase lag), so that the remaining change in carrier phase is close to O over a long period of time. In a system for transmitting logging data while the borehole is being driven, using an electromechanical device, such as a drilling fluid siren, to transmit acoustic waves to the drilling fluid, it is too prefer that all phase changes be made in the same direction (i.e., either phase lag only or phase lead only) because then driving the drilling fluid siren is more effective and much easier can be made. (For example, if all phase changes are achieved by momentarily lowering the frequency, it is never necessary to to increase the frequency above the nominal frequency and less drive power is required for the drilling fluid siren. In addition, the control circuitry can be made less complicated.) The term "unidirectional" PSK modulation means this type of modulation all phase changes take place in the same direction.

The modulated acoustic signal is transmitted to the surface of the earth one or more transducers that convert the acoustic signal into an electrical signal. It is then necessary to have the digital information recover contained in the modulation of the received signal. In short, this is accomplished by first doing the The received signal is processed to extract the carrier signal. The recon & ruined carrier is then used to generate the modulated electrical Demodulate signal synchronously.

In the conventional implementation of the system described, a band pass filter is typically used in the receiver, the filter

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has a bandpass spectrum centered with respect to the nominal carrier frequency and is used to detect the modulated carrier. As part of the present invention, however, it has been found that the use of a filter centered on the nominal carrier frequency does not produce optimal behavior. The unidirectional type of modulation leads namely to the fact that the mean carrier frequency deviates from the nominal carrier frequency. It has also been found that another problem arises using conventional existing filters for phase shift systems of the type described. A typical conventional filter strives for a symmetrical spectral characteristic with respect to the filter center frequency at. The unidirectional type of modulation, however, shows that a syirmetric filter characteristic does not mean optimal adaptation with respect to the frequency characteristic of the bypassed signal.

It is an object in one aspect of the present invention to provide an improved filter for use for detection in a phase shift transmission system of the kind in which the modulation is achieved by temporarily changing the carrier frequency unidirectionally.

In one known type of the system described, typically a carrier location loop is used in the receiving part, its task consists in locking on the carrier of the received signal and to generate timing signals that can be used in the emodulation process. It is desirable to have an interlocking on the carrier like this as quickly as possible in order to avoid a possible loss of information. It is also desirable once locking has been achieved has been to have a carrier loop that is relatively stable, i.e. not adversely affected by transient error component signals in the loop at different frequencies. These two objectives are somewhat contradicting each other, as they are relatively quick to do interlocking requires a relatively wide loop width, while stability of the loop generally requires a relatively narrow loop width would dictate. It is known that the loop widths can be varied manually once the lock has been made, but this technique is used

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i would not be particularly comfortable. Furthermore, result from the above-mentioned application in connection with the transmission of log values during the Sinking the borehole with the relatively low-frequency acoustic signals are practical problems when trying to determine the loop width of the Modify carrier tracking loop. The change in the width of the loop namely generally involves switching on different capacitors into the loop filter circuit and at the same time the JMbdification of the loop gain. At the frequencies of interest here, the condensers in the circuit generally have relatively large values and are therefore used Electrolytic capacitors that have relatively large capacitance without overly large dimensions, as is typical of non-electrolytic capacitors is. If a previously inactive capacitor is switched into the circuit, a problem arises because of the introduction of an offset voltage, resulting from the previous voltage across the new capacitor that does not match the voltage applied to it once it has been switched into the circuit. It's someone else's job It is an aspect of the present invention to provide an improved variable loop width carrier locator loop that addresses the problems noted above are avoided.

In the known types of system described, the carrier is generally extracted using a carrier location loop. The carrier location loop is a phase locked loop with a voltage controlled oscillator (VCO) that is responsive to Error signals resulting from the difference between the phase of the signal, derived from. VCO, and the phase of the carrier signal. According to the present However, the invention has shown that the unidirectional type of phase modulation tends to create a problem in the operation of the phase locked loop in the system described above. Namely there Phase changes are realized by momentary variation of the frequency, error pulses in the phase lock loop will always occur if there is a data transfer or data jump. Since the PSK-MDdulation is unidirecticnal (i.e. momentary frequency modification always takes place in the direction of a lower frequency or always in the direction of a higher one

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Frequency), these error pulses have the same polarity. These error pulses can tend to introduce undesirable frequency deviations in the carrier location loop.

It is another aspect of the object of the present invention to provide an improved carrier location loop for use in the Detect in a PSK phase modulated transmission system where the modulation is is effected by temporary unidirectional modification of the carrier frequency.

The solution to the problem described above within the framework the invention is defined by the claims.

One aspect of the present invention is directed to a method using a PSK signal that is associated with digital information has been modulated by momentarily unidirectional the nominal frequency of a carrier signal as a function of the digital information for the purpose of causing a Phase change has either increased or decreased; that will modulated carrier signal filtered with a filter that has a bandpass center frequency which is shifted from the nominal carrier frequency in the direction of the unidirectional increase or decrease in frequency. The digital signal is recovered from the filtered signal.

Another aspect of the present invention is directed to a method directed using a PSK signal that has been modulated with digital information by either decreasing or unidirectional instantaneous Increasing the nominal frequency of a carrier signal as a function of the digital information to effect a phase change; the following steps are provided: filtering the modulated carrier signal with a filter with a passband characteristic, which is symmetrical in the direction of the unidirectional decrease or increase of the frequency, after which the digital information is recovered from the filtered signal.

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Another aspect of the present invention is directed to a method directed to reducing the transition signals resulting from switching of another capacitor in a circuit for use in an electronic system having a pair of terminals, a plurality of Capacitors, each of which is coupled with one of its pads to one of the terminals, which furthermore has a circuit arrangement for coupling the other pad of a selected one of the capacitors to the other terminal, and which ultimately has variable gain circuitry, synchronized with the switch arrangement for influencing the potential of the other terminal. In this case, the characteristics according to the invention are: Continuously generating a reference voltage associated with each of the plurality of capacitors that are not currently connected between the pair of terminals where each of the reference voltages generated is a function of the voltage that would be present across the associated capacitor in the event that if the associated capacitor were instantly switched between the pair of terminals by the circuitry, and permanent application each generated reference voltage to the associated capacitor.

Another aspect of the present invention is directed to a method directed to stabilize a carrier locating loop e, used in conjunction with a device that receives a PSK signal containing digital information had been modulated by momentary unidirectional either lowering or increasing the nominal frequency of a carrier signal as a function the digital information to effect a phase change. The apparatus includes a carrier locating loop for determining the carrier of the received Signal. The loop circuit comprises a locked oscillator with a control terminal, and the frequency of the oscillator is determined by a Signal that is applied to the control terminal. A comparator to generate of a control signal is provided, the phase of a signal derived from the received PSK-modulated signal being compared with the phase a signal derived from the output of the controlled oscillator. Circuits are provided for applying the control signal to the control terminal of the oscillator. The method comprises the following steps: Generation of

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Koitpensaticnsiirpulsen in response to transitions (jumps) of the received Signal and application of the compensation pulses to the control terminal, so the Difference between the nominal frequency of the carrier and the mean frequency of the received signal to take into account which difference arises from the unidirectional type of carrier modulation.

The present invention also includes another aspect directed on an arrangement designed as an Eirpfänger for a PSK signal is modulated with digital information. The arrangement serves to provide the digital information recover from the carrier. The PSK modulated signal was modulated with the digital informatics by momentary unidirectional either lowering or increasing the nominal frequency of the carrier signal in Function of the digital information to effect a phase change. The order includes a filter for use in selective filtering of the modulated Carrier signal. According to the invention, the filter has a passband center frequency which is offset from the nominal carrier frequency in the direction the unidirectional decrease or increase of the frequency.

The present invention includes another aspect directed to an arrangement for demodulating a digital signal. The arrangement receives a PSK signal modulated with digital information and serves to display the digital information recover from the signal. The PSK modulated signal had been modulated with the digital information by current unidirectional either lowering or increasing the nominal frequency of a carrier signal as a function of the digital information to effect a phase change. Thereafter the carrier returns to the nominal frequency after the phase change has been effected. The arrangement surrounds a filter for use in the ffiLective Filtering the modulated carrier signal. According to the invention, the filter a bandpass characteristic that is asymmetrical in the direction of unidirectional lowering or increasing of the frequency.

Another aspect of the present invention is to an electronic one Circuit arrangement directed to a pair of terminals, a plurality of Capacitors, each of which has one of its pads on one of the terminals

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is coupled, a switch arrangement for coupling the other covering a selected one of the capacitors to the other terminal and variable gain circuits has, synchronized with the switch arrangement for influencing the potential of the other terminal. According to the invention Such a circuit arrangement is characterized by a circuit for reducing the transfer signals resulting from switching on another Capacitor between the pair of glues, wcte are provided: circuit components for the continuous generation of a reference voltage, assigned to each of the plurality of capacitors that are not currently between the pair are coupled by terminals, each reference voltage generated being a function of the voltage across the associated capacitor in would appear if the associated capacitor were instantly switched between the pair of terminals by the switch arrangement, and Circuit components for the permanent application of the generated reference voltage to the assigned capacitor.

Another aspect of the present invention is to a carrier location loop with variable loop width directed to lock onto the carrier of an input signal. The loop comprises a phase locked loop with an oscillator with a control input, an error signal generator for generating an error signal as a function of the phase difference between a signal derived from the oscillator and the input signal and a variable filter having a plurality of different ones Bandwidths for coupling the output of the error signal generator to the control input of the oscillator. Such a loop is in accordance with the invention characterized by a loop width control circuit coupled to the variable filter for automatically changing the loop width of the Filters in function of the input signal.

The embodiments of the invention are given below

explained in more detail with reference to the accompanying drawings.

1 shows a simplified, largely schematic representation a device for the transmission of measured values while drilling,

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Fig. 2 shows diagrams to explain the conventional PSK-MDdulaticn. and the unidirectional ramp phase PSK-MDdulaticn according to the invention,

Fig. 3 is a block diagram of the surface receiving system for the apparatus of Fig. 1;

FIG. 4 shows idealized waveforms for explaining the nature of the signals which occur at various points in the receiving system according to FIG. 3,

Fig. 5 illustrates the nature of a phase change according to the PSK-MDdulaticn from Fig. 2,

Fig. 6 illustrates the nature of the frequency spectrum of a conventional PSK modulated Signal compared with the spectrum of a unidirectional PSK-modulated signal,

Fig. 7 shows an example of a filter that is used in the subject matter of the invention,

Figure 8 is a block diagram of a phase locked loop with variable loop width according to an embodiment of the invention,

shows the basic principle of a loop filter,

shows a variable loop width filter according to the invention;

shows a traditional carrier tracking loop,

shows an embodiment of a carrier locating loop improved according to the invention and

Fig. 13 illustrates one type of waveform that is generated as an input for the loop filter of the carrier location loop according to FIG. 11.

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Fig. 9 Fig. 1o Fig. 11 Fig. 12th

Fig. 1 shows largely schematically a device for the borehole investigation during the return of the borehole according to a Embodiment of the invention, being from a conventional drilling unit Use is made. A platform with turntable 1o is above a borehole 11 drilled into the ground by itotary drilling has been. A drill string 12 is suspended within the borehole and carries a drill head 15 at its lower end. The drill string 15 and the Drill head 15 attached to it are rotated by a turntable 16 offset (drive not shown), which is in engagement with a wedge bar 17 at the top of the drill string. The drill string is hanging on one Hook 18 attached to a moving block (not shown). The wedge bar is connected to the hook via a shackle 19, the one Rotation of the drill string relative to the hook allows. Drilling fluid 26 is located in a sump 27 in the earth. A pump 29 pumps the drilling fluid into the drill string through an opening in the shackle 19 so that they can flow down through the center of the drill string 12. The drilling fluid emerges from the drill string through openings in the drill head 15 and flows then upwards in the annular gap between the outside of the drill string and the periphery of the borehole. As is well known, the drilling fluid promotes this Formation particles to the surface of the earth, and the drilling fluid becomes Recirculated sump 27 for recirculation. The small arrows in Fig. explain the typical flow direction of the drilling fluid.

Within the drill string 12, preferably near the drill head 15, there is a measuring and transmitter assembly 5o. The assembly includes 5o a logging device 55 containing any borehole characteristic of interest measures, for example the resistance of the ground, gamma radiation, drill head load, Tool cutting angle, etc. It will be understood, however, that the measuring device 55 can be used for measuring any parameters of interest we can be employed. The transmitter portion of the assembly includes an acoustic transmitter 56 which transmits an acoustic signal into the drilling fluid, this is representative of the measured borehole parameters. A useful type of acoustic transmitter known per se

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uses a device called the "drilling fluid siren", which has a slotted rotor and a slotted stator which revolves and repeatedly interrupts the flow of drilling fluid to a desired one generate acoustic wave signal in the flush. The transmitter 56 is controlled by a transmitter control and drive electronics 57, which comprises an analog-to-digital converter, which the measured parameter signals converts into digital form. The control and drive electronics 57 also have a phase shift modulator (PSK modulator), which is a Anrfebssignal generated for application to the transmitter 56 on.

With conventional phase shift (PSK) communication, the Changed the phase of the carrier signal in accordance with a digital data signal having two or more levels to produce a modulated carrier, which has two or more phases. The carrier phase is conventionally changed in alternating directions (i.e. alternating phase lead and phase lag), so that the remaining change in the carrier phase is close to zero over an extended period of time. In a drilling and measuring system with an electromechanical device, such as a drilling fluid siren, for the transmission of acoustic waves to the drilling fluid, it is preferable move to make all phase changes in the same direction (i.e. always lagging or leading), because then the technology for driving the Drilling fluid siren is more effective and simpler. The term "unidirectional" PSK modulation, which is used below, is intended accordingly for this type of Denotes modulation in which all phase changes occur in the same direction. Techniques for driving a drilling fluid siren to generate a PSK-modulated acoustic carrier wave in drilling fluid and for achieving a unidirectional PSK modulation are for example in U.S. Patents 3,789,355 and 3,820,063. It goes without saying, however, that other suitable means for such unidirectional PSK modulation can be used as described below. Fig. 2 illustrates the different PSK conventional PSK modulation and the unidirectional PSK modulation for use in a simultaneous measuring and drilling system. diagram 2A shows an unmodulated carrier signal with a period of T / 4, where

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T is the bit period of the modulating information. A bit pattern intended as an example is shown in diagram 2B, where the "O" to "1" transitions occur at times 2T and 5T and. "1" to "O" transitions to time points T, 4T and 6T. When a conventional differential encoded PSK coding scheme is used, a phase change in the bit time epoch (T, 2T, 3T, 4T ...) is indicative of a "1" bit, while the lack of a phase change in the bit time epoch is indicative of a Bit "O". It will be understood, however, that an opposite convention could be made, or that any suitable coding scheme compatible with the invention could be used. Accordingly, in diagram 2C, which shows the conventional PSK modulation, a phase change of θ inside is provided when the next bit is a "1", which means that phase changes occur at times 2T, 3T and 5T. Accordingly, the diagram 2C shows phase changes that occur at these times with phase changes that alternate in direction. The diagram 2D shows the nature of the PSK-MDdulation when using the unidirectional MDdulation as it is provided here. Phase changes take place at the same time points, but in this example, each Phasenähderung negative (ie, results in a phase lag), and _r can be seen that the phase changes accumulate.

Referring again to Figure 1, the generated acoustic wave propagates (i.e., the primary compciente of those that are received is said to go up in the drilling fluid through the center of the drill string at the speed of sound in the fluid in question. The acoustic wave is received at the earth's surface by transducers 31. The transducers where For example, it can be a piezoelectric transducer, convert the received acoustic signals into electronic signals. The outcome of the Transducer 31 is coupled to the surface receiving system 100, which is used to demodulate the transmitted signals and those in the borehole to display the measured value information obtained on the display and / or recording unit 5oo.

Fig. 3 shows a block diagram of the at the earth's surface Receiving system with an improved filter according to one aspect of the invention.

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The waveforms of Fig. 4 depicting an exemplary bit pattern "loll" are used from time to time for explanatory purposes. The acoustic signals in the borehole fluid are picked up by transducers 31 (Fig. 1) comprising transducers 31A and 31B according to the invention. In this embodiment, this pair of transducers is used in conjunction with a differential detection system is used which comprises a delay element 1o3 and a differential amplifier 1o4. The output of converter 31B is Via a buffer amplifier 1o2 and delay element 1o3 to the negative input terminals of the differential amplifier 1o4 applied. The converter 31A is connected to the positive input terminal of the differential amplifier via buffer amplifier Ιοί 1o4 laid. This differential detector arrangement is used, for the purpose used to suppress noise that is in one direction of propagation runs, which runs counter to that of the primary acoustic carrier wave. For example, when the distance between the transducers 31A and 31B is chosen so that it is equal to a quarter wavelength at the carrier frequency, and if the delay element 1o3 also to a quarter wavelength is set at the carrier frequency, the acoustic waves that propagate in the direction of the primary signal (arrow A) are subject to one total phase delay of half a wavelength. If the output of the delay element 1o3 is affected by the instantaneous signal, the converter 31A, is subtracted, the phases of the signals running in the direction of arrow A add up. Acoustic signals, however, that are in opposite Running direction (arrow B) lead to inputs on the differential amplifier 1o4 which are in phase so that these signals cancel each other out. this is readily understood without considering the fact that in such a case the input is at the positive input terminal of the differential amplifier 1o4 a quarter wavelength delay due to the transducer spacing, while the input is at the negative output terminal of the differential amplifier 1o4 is subject to a quarter wavelength delay because of the electrical delay element 1o3.

The output of the differential amplifier 1o4 is sent to a bandpass filter Ho coupled with a filter characteristic according to the principles of a Aspect of the invention, which are explained in detail below.

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The output of filter 11o is fed to a regulated amplifier 115 coupled, which has a Schnellanspredi / Langsaraausgleichscharistlk. The fast response mode is useful for achieving stability and sync lock in a minimal amount of time, while the slow release mode maintains gain during momentary loss of signal or level change. The output of the regulated amplifier 115 (shown in idealized form in diagram 4A) is sent to a synchronous demodulator 13o and a carrier location loop 12o with a variable loop width are coupled. The carrier tracking loop with variable loop width may include one phase lock loop and biüfet another important aspect of the present invention. The variable loop width can be operated either manually or automatically. In manual operating mode the carrier tracking loop works in a certain specified Loop width (e.g. wide, medium or narrow) depending on the choice made by the operator. These loop widths can for example, 0.3 Hz, 0.1 Hz and 0.03 Hz correspond, with which a range is covered more of an order of magnitude. The wide or medium loop width is typically used when interlocking is sought, and the narrow loop width is turned on once interlocking a has been aimed at once to improve the loop stability. In the automatic operating mode, the loop first strives for synchronization using the widest loop width (or the middle Loop width, if this is desired under certain conditions). After synchronization has been achieved, the loop width is switched to a narrower value. If a signal loss occurs, indicated by an output from a loss of signal detector in circuit 12o the loop width is switched back to the widest Einsehaitaanfstellung. Both in the manual and in the automatic mode of operation, the carrier location loop with a variable loop width can be used Circuitry may be equipped to pre-charge certain capacitors that will be switched in or out of service when the loop widths are switched. This capacitor precharge technique is advantageous to prevent a possible loss of lock if, for example, a switch is made to a narrower bandwidth, which may be the cause

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could be caused by transient voltages originating from the original voltage across capacitors that are put into operation in the circuit. Further details of the carrier location loop with variable loop width are given below.

As will also be explained in detail below, the output ναι of the carrier locating loop 12o is from the output with a variable loop width a voltage controlled oscillator (VCO) in the phase locked loop of the circuit. This oscillator typically works at a multiple of the nominal carrier frequency. A clock generator with a frequency divider accordingly routes a clock signal from the VGO output from which derived clock signal (shown in diagram 4B) is at the carrier frequency and has a shape that is suitable for demodulating the filtered input signal. The clock generator in circuit 12o may have a clock correction circuit according to one another aspect of the present invention. As in detail below described, the unidirectional nature of the PSK-modulated carrier signal leads to build up fault signal components in the carrier location loop. If this is not taken into account, for example by using the Clock correction circuit described below, the structure of the error component signals can cause an undesirable drift of the voltage controlled Cause oscillator in the carrier location loop. This undesirable build-up of error components can be eliminated by providing Displacement pulses, which tend to compensate for the error signals, which would otherwise accumulate. Since this type of error signal, which is under discussion here occurs with every bit transition, the output of a bit transition detector 15o (to be explained in detail) used to regulate the generation of correction pulses.

The output of the storage location loop 12o (diagram 4B) is sent to the Synchronous demodulator 13o coupled, which, as mentioned above, to its other Input is fed to the output of the regulated amplifier 115, whose output signal is to be demodulated. The synchronous demodulator can be an analog multiplier, for example. Its demodulated output is

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shown in wave form in diagram 4C. The output of the synchronous demcdulator 13o is coupled to a matched filter 14o. The filter 14o is matched to a Hechteckinpuls with the bit RAte. As known, the matched filter has the effect of integrating on a data transition at its input for a time equal to one bit period. Accordingly, at the end of each bit period is the output of the matched filter at an extreme positive or negative value (waveform of diagram 4D) at which the sampling can be carried out with the best efficiency. the The output of matched filter 14o is sampled by means of a and hold circuit 16o, the output of which is coupled to an analog-to-digital converter 17o is used to generate a signal in digital form. (The output of matched filter 14o is also passed to bit transition detector 15o coupled, which may comprise a zero crossing detector for detecting zero crossings of the output signal from the matched filter, so as to output pulses to be generated with a phase that is synchronized with the Bit transitions, the use of this transition detector output signal is detailed below). The signal that is used to trigger the scan through the sample and hold circuit 16o and for the definition of the Conversion period of the analog-to-digital converter 17o is used is generated from a sampling generator 18o. The sampling signal generated by the sampling generator (Waveform in diagram 4F) is, as can be seen, at the bit or Symbol rate. IM to get this relatively accurate signal at the bit rate a carrier-supported symbol location loop 19o can be used. the Carrier-assisted SynfoL location loop is in US patent application S.N. 684,6o4. In short, circuit 19o is a squaring one Type of phase locked loop with a voltage controlled oscillator and a frequency divider in the loop. Regarding this the circuit is constructed like a conventional bit synchronization circuit. However, as described in the aforementioned US patent application, is in addition to the tracking loop received time information, when a transition is detected in the received signal (i.e. the output of the Bit transition detector 15o in Fig. 3) the output of the carrier location loop 12o used to support the Syitibolortungsschleife 19o (output shown in Diagram 4E), during those periods during which

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Symbol passages are missing. This is made possible by the coherent relationship between the carrier and the bit rates. If there are no bit transitions after a number of bit periods, a signal derived from the carrier is used, to maintain synchronization.

The bit pattern output of the analog-to-digital converter 17o is for this Example shown in Diagranm 4G, and it can be seen that it results from sampling the output from the matched filter (Diagram 4D) with the scanning signal (diagram 4F) and subsequent analog-digital conversion. Since the data was originally in an unconventional Differential Coded PSK form have been encoded (as described above), a differential decoder 199 is used to retrieve the data in its original form. Since in individual a phase change indicative of a "1" in the coding scheme was, a bit change in the output of the analog-digital converter 17o (Diagram 4G) interpreted as a "1" by the differential decoder 199. The opposite is true for the lack of a bit change in the analog-digital converter output interpreted as an "O". Accordingly, and as is known per se, the contains Differential decoder has an exclusive-OR gate that responds to successively received Bits responds and produces a 1 output if consecutive bits are different, and an 0 output if consecutive bits are the same. The output of the differential decoder 199 is illustrated in FIG. 4H for the selected example.

It goes without saying that in illustration 4A from FIG. 4 the PSK Msdulaticn in idealized form has been shown with "instantaneous" phase changes in order to understand how the circuit works Fig. 3 to simplify. The actual phase changes are made in the manner explained in connection with diagram 2D. Fig. shows such a phase change, caused by a momentary lowering of the carrier frequency until the desired phase shift is reached. The dashed Line shows what the carrier waveform would look like without the frequency change being made.

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In one embodiment of the invention, the carrier frequency is 12 Hz and the bit rate is 1.5 Hz. Oil-directional PSKH modulation is implemented by momentarily lowering the carrier frequency to 8 Hz until a 180 ° phase lag has been achieved and then returning the carrier to its nominal 12 Hz frequency. (The desired lag is one-half the period of the nominal carrier frequency. One frequency of 8 Hz has a period which is 1 1/2 times the period of the nominal carrier frequency amounts to. Accordingly, the desired phase lag is achieved after a complete cycle at 8 Hz (125 milliseconds). this is readily apparent from FIG. 5, in which the solid waveform line changes to 8 Hz during one cycle, while the dashed waveform line illustrates continuation at a 12 Hz frequency. There However, it takes a finite time for the change between the two frequencies, and since during the transition the mean frequency is lower is than 12 Hz, the time actually consumed at 8 Hz is slightly shorter as 125 milliseconds, techniques for driving drilling fluid sirens in this manner are known per se, and may be exemplified see the above-mentioned IB patents 3,789,355 and 3,820,063, respectively will.

After the entire reception system has been globally explained, hereinafter certain aspects thereof which are in accordance with the invention are formed, explained in detail. According to one aspect of the invention has already been stated that the unidirectional phase shift of the carrier causes the spectrum of the modulated signal to be shifted in frequency relative to the nominal carrier frequency. The frequency shift or offset is accompanied by an asymmetry in the spectrum. Figure 6B illustrates the nature of the unidirectional PSK spectrum and can be compared with the frequency spectrum of a conventional PSK modulated signal with the same nominal carrier frequency f. The use of a band-pass filter (e.g. filter 11o of Fig. 3) which allows this offset and asymmetry of the frequency spectrum of the Modulated signal taken into account is advantageous for the witeamere Separation of the signal from the noise and for minimizing the

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Distortion of the signal of the signal by the filter. The exact degree of Spectrum shift and asyrrmatic depends on the data pattern of the modulatics. For example, an alternating "3", "O" data pattern would result in an offset by an amount equal to about the bit rate. Another data pattern will result in a displacement that is slightly lower would be than the bit rate. When the data pattern is not known from the start (as is usually the case), the data pattern can be random are assumed, and such a data pattern leads to an offset with respect to the carrier frequency of about half the bit rate. In the described Embodiment with the carrier at 12 Hz, the bit rate is 1.5 Hz, and the PSK modulation is achieved by unidirectional momentary lowering of frequency so that the preferred filter center frequency of the bandpass filter would be 11.25 Hz, i.e. minus the nominal carrier frequency half the bit rate. (It goes without saying that if the phase shift were achieved by unidirectional momentary increase in frequency, the offset was in the direction of the higher frequencies and would be at 12.75 Hz.)

There are several in accordance with the principles of the present invention Ways in which the bandpass filter can be designed. The bandwidth of the filter is chosen so that the modulated signal with a minimum of distortion, while at the same time noise and interference be suppressed. The minimum bandwidth (saintts -3dB to - 3dB) for filtering in a PSK system is typically equal to the bit rate, although generally a slightly larger bandwidth, for example 1.5 times the bit rate is recommended. When designing the B-pass filter one can follow the following steps: First, a low-pass prototype filter is selected and calibrated so that it has a bandwidth equal to half of the desired bandpass filter bandwidth. The low pass filter design is then extended to a band pass filter, centered on a frequency which is offset from the carrier frequency according to the rules given above. The BanφaB filter transmission zeros are then selected to the desired filter symmetry (or Asyirne-trLe) to effect. A certain filter configuration is then assumed and

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909843/0622

the filter component values for this are calculated. Since the details of a such an interpretation belong to the knowledge of the average skilled person they are left aside here in the interests of brevity.

For example, a filter can be implemented using a cascade connection of two active RC biquadrate filter compartments. One feedforward circuit configuration described in "Design Formulas for Biquad Active Filters Using Three Operational Amplifiers" published in IEEE May 1973 can be used. The finished filter can be assembled be composed of two active RC biquadratic sections connected in cascade, represented by the transfer functions according to FIG. 7 with

b ₁ = 2Ο.764 =
b _Q = 5987.63
d _Q = 4292.79
f

for the first biquadratic filter section. The interpretation formulas of the mentioned article can be used to calculate the values of the filter components. For example, for the first section

R ₀ and C ₁ = C ₀ as well as

OI Δ

K ₁ = K ₂ chosen by the designer. Then: 1

h =

^K 2 ^C 1

ρ = R = co (ie with an open circuit) ⁵ 6

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- 30-290708a

Above, only one example of how to design a bandpass filter has been given may that are useful in accordance with the principles of the invention, but numerous alternative construction techniques can be used.

FIG. 8 shows an embodiment of the carrier locating loop 12o (FIG. 3) with a variable loop width according to the invention. A Squaring circuit 2o1 receives the output from regulated amplifier 115 (Fig. 3) i.e. the filtered, gain-regulated PSK-modulated Signal. The squaring is used, essentially the modulation of the carrier and also doubles the frequency in the press of the wearer. The output of the squaring circuit 2o1 is an input for a phase detector 2o2. The other input of the phase detector 2o2 is the output a frequency divider (or clock divider) 2o3. The output of the phase detector 2o2 is coupled to a new type of filter 3oo with a variable Loop width, which will be explained in detail later. The output of the filter 3oo is coupled to the voltage controlled oscillator (VCO) 2o4, and the output of the VCO 2o4 in turn is coupled to the clock divider 2o3.

The loop width of the filter 3oo with variable loop width can either be manual or automatic under control by the loop pulp control circuit 2o5. In the automatic mode of operation, the loop width control circuit 2o5 receives the output of the signal loss detector 2o6. The loss of signal detector 2o6 includes a comparator that detects the loss of lock in the loop by comparing the input signal (from regulated amplifier 115) with an adjustable threshold level recorded. When the input signal is lower than the threshold level means this is a loss of locking. The loop width control circuit 205 is responsive to a loss of signal indication for a loop width modification for the filter 3oo with a variable loop width, namely in the direction of greater loop width. When the lock again has been established, or, for example, after a predetermined period of time if there is a high probability that it will be locked again has been, the loop width control circuit 2o5 causes a loop

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90-9843 / 0622

width modification of the filter 3oo with variable loop width in the direction narrower loop width. Switching takes place in manual operating mode manually with switch 2ο5Ά.

The loop width (or bandwidth) of the phase locked loop generally determines the time required for the loop to lock and also determines the stability of the loop; i.e. their ability maintain the lock in the presence of a noisy input. As mentioned above, a larger loop width is advantageous for achieving locking quickly, but once locking has been established, the larger loop width is disadvantageous because it leads to a lower stability than a phase-locked loop with a narrower loop width. It is therefore advantageous to have a larger loop width for the interlock itself and then to switch to a narrower loop width after the interlock has been established is to improve the stability of the loop. According to the invention can the modification of the grinding width can be carried out automatically. A an important feature of the invention is prevention of introduction of offset voltages in the loop when switching between different Loop width, which could otherwise lead to a loss of the lock.

In order to better understand the invention it is useful to begin with consider the basic loop width filter according to FIG. 9. Of the The output of the phase detector 2o2 (Fig. 8) forms an input for the positive Input terminals of an operational amplifier 4o1. The negative input sound of the operational amplifier 4o1 is coupled backwards to the output of the amplifier via a capacitor C. The output of the operational amplifier 4o1 is also connected to the positive input of a further operational amplifier via a gain control resistor network 4o2 (framed by a dashed line) 4o5 coupled. In this simplified representation, the gain control network comprises a series resistor R_ and a resistor. R ^, which is connected to the ground reference potential. The output of the operational amplifier 4o5 is fed back to the negative input of the same. The

- 24 -

909843/0622

The output of the operational amplifier 4o5 is also applied to a voltage divider, Consists of series resistors 99R and R, which is at ground reference potential. The junction between the resistors is the voltage divider fed back to the negative input of the operational amplifier 4o1. The transfer function of the loop width filter according to Fig. 9 is:

_F (S) = A (S + 1) / RC)
(S + A / I00RC)

If integrated into the phase lock loop according to FIG. 8, the The closed loop transfer function can be expressed as:

H (S) =

(S + AKS + AK / RC)

where A is a gain factor that is less than or equal to 1, regulated through unit 4o2, and K is a loop gain constant which changes in proportion to the VCO frequency. It can easily be shown that the loop width can be changed without affecting the damping factor the loop when A and either R or C are changed in inverse proportions to each other. Typically, A and C can be used individually Steps can be changed. However, as stated in the introductory section, Switching the loop width during operation can lead to data loss as a result of a lock failure caused by a Displacement voltage in the loop width filter when the loop width is switched will. For example, assume in Fig. 9 that a certain Voltage across the capacitor C in the loop filter. To change the loop width, another capacitor is typically added to the loop filter circuit (instead of C), and at the same time the gain of the loop filter is changed. When this happens, a different voltage applied across the new capacitor. If the If the initial voltage across the new capacitor does not have a correct value, the change in the gain factor can lead to a disturbing error signal in the Lead loop through which the lock is lost.

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909843/0622

In Fig. 10 an embodiment of an adaptive loop width fliter is shown with a feature of the invention, according to which capacitors are precharged to prevent the loss of locking when Ua switching to different loop widths. The operational amplifiers 4o1 and 4o2 and the resistors, which were marked with 99 R and R in FIG. 9, remain unchanged. The resistor R _{1 of} the gain control network from FIG. 9 is offset by three individual resistors which are coupled to ground via three position pole sections 48oA of a switch 48o. Depending on the switch position, one of the three resistors R ₁₁ , R ₁ ″ or R_ is placed between the positive input terminals of amplifier 405 and ground reference potential. The capacitors C ₁₁ , C _1? or C ₁ ^. can be viewed as a substitute for capacitor C from Fig. 6. By operating switch sections 48oB, 48oC and 48oD of switch 48o, one of these capacitors is connected between the negative input terminal of operational amplifier 4o1 and a point which is at a fixed voltage across the output of the operational amplifier 4o1 is. This fixed voltage can be, for example, 5.1 volts, due to the use of the zener diode 412 and current sources 415, 416. The positions of the various sections of the switch 48o in the embodiment of FIG. 1o are operated together. The three positions of the switch are labeled "w" (wide, "m" (medium) and "n" (narrow)), which denote the achievable loop width settings of the circuit for this embodiment. The control of the switch can be either manual or automatic , as effected by the loop width control circuit 2o5 (Fig.8). It can be seen that with the switch in the W position, the resistor R ₁₁ and the capacitor C .. are in the loop, with the switch in the M position Resistor R ₁₂ and capacitor C ₁₂ and the switch in the N position, the resistor R ₁ - and the capacitor C.3 At relatively low operating frequencies, as occur in the measuring / drilling process as described above, relatively high values are used for the For example, the capacitors C ₁₁ , C ₁₂ or C ₁ -, can have values of 10, 33 or 100 microfarads. In order to avoid extremely large capacitors, it is practical to use electrolytic probes nsators which require a bias, and this is taken into account in the circuit of Fig. 10 by the bias current sources

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290708B

415 and 416 as well as Zener diode 412. A filter capacitor 413, which typically has a high value of about 220 microfarads, is parallel to Zener diode 412. The individual resistors R ₁ -, R "or R ₁ - can have values between infinity (circuit open ), ^J 3.86 KOEftn and 1, oo KOHm and the resistor 414 can have a value of 9, o9 KOhm.

Based on the circuit section according to FIG. 10 described so far, it is assumed that the adaptive filter is operated in the operating status with a wide loop width, ie with resistor R ₁₁ (open) and capacitor C ₁₁ . If the output of the operational amplifier 4o1 is at a voltage V ₁ , and since the input impedance for the operational amplifier 4o5 is very high, the voltage at the input of the operational amplifier 4o5 is also approximately V ₁ . Assume now that the loop width toggle control for switch 48o is toggled to the middle loop width position. The resistor R ~ now forms a voltage divider with the resistor 414. Since R ₁ "only makes up 3/1 ο of the total resistance of resistor 414 plus R ₁₂ , the voltage at the input of the operational amplifier 4o5 would be reduced to a value of about (o, 3) V ₁ fall. The output of the operational amplifier 4o5 would accordingly be reduced immediately to 3/1 ο of its previous value. This jump itself causes a loss of locking, since the output of the amplifier 405 is coupled to the loop VCO (FIG. 5). The positive side of capacitor C ₁₂ which is switched into the circuit is 5.1 volts above voltage V ₁ (as is the positive side of capacitor C ₁₁ which was switched out of the circuit.) On a sudden jump at To avoid the output of the amplifier 4o5, the initial _{voltage across C 12 should be} greater than the voltage that was across C ₁₁ by a factor of 1o / 3. Accordingly, and as will be explained in the following, according to the invention a corresponding pre-charging of the capacitors is provided which are not currently in operation in the circuit. Furthermore, however, the following consideration should be taken into account: Two signal components are generally present in the loop filter circuit, namely an alternate signal component and a DC voltage or a very low-frequency error voltage. Since the positive side of all three capacitors C ₁₁ , C and

- 27 -

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.35-2907095

C ₁ -. are coupled to a common point (namely the 5.1 V above the output voltage of the operational amplifier 4o1), care must be taken not to reduce the inoperative capacitors (i.e. those that have just been extracted from the circuit) to a fixed gain factor times both components, since the alternate component is a continuous mode signal that should remain the same regardless of the loop width chosen.

In the circuit of Fig. Io, a voltage which is representative is for the voltage across the capacitor that just turned into the circuit to each of a plurality of gain control amplifiers 421, 423 and 425 applied. More precisely, the tension will the 5.1V below the voltage on the positive side of the straight in the circuit used capacitor is applied to the positive input terminal of each of these amplifiers 421, 423 and 425, and the voltage at the negative input of the op amp 4o1 (which also controls the voltage on the negative Side of the capacitor that is used in the circuit) is connected to the negative input terminals of each of amplifiers 421, 423 and 425 are applied. Three further levels of switch 48o, labeled 48oE, 48oF and 48oG, are used in addition, one of the three gain control inputs with a gain control terminal to connect each of the respective amplifiers 421, 423, 425. In the illustrated embodiment, the gain control multipliers are applied to the amplifiers 421 for the switch positions w, m and n, 1, o, o, 3 and o, 1, respectively. The gain control multipliers applied to the Amplifier 423 for switch positions w, m and η are 3,3, 1, o and o, 33. The gain control multipliers applied to the Vastärker 425 for the switch positions w, m and η are 1o, 3, ο and 1, o, respectively. ! fen recognizes that the gain control multipliers, applied to the gain controllers 421, 423 and 425 via the switch levels 48oE, 48oF and 48oG, respectively any means known per se can be generated / for example by switch on appropriately weighted resistors (not shown) in voltage divider circuits, to derive the desired multipliers.

The outputs of amplifiers 421, 423 and 425 are connected to the negative input terminals of the operational amplifiers 422, 424 or 426.

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-36- 2907Q8S

couples. The positive input terminals of these amplifiers are each connected to the output of the operational amplifier 401, so that each of them receives a signal which is 5.1 V below the voltage on the positive side of the capacitor which is currently used in the circuit. The outputs of the amplifiers 422, 424 and 426 are each coupled to two poles of the corresponding switch levels 48oB, 48oC and 48cD, respectively. The three switch levels are arranged in such a way that the negative shields of the capacitors, which have just been switched out of the loop filter circuit, are connected to the outputs of the corresponding amplifiers (422, 424 or 426). In detail, the capacitor C ₁ - is coupled to the output of the amplifier 422 for the m and n switch positions, capacitor CL "is coupled to the output of the amplifier 424 for the w and n switch positions, and the capacitor C ₁₃ is £
m-switch positions.

and the capacitor C ₁₃ is connected to the output of the amplifier 426 for the w and

In operation, the switch causes 48o the önschaltung the filter abrasive eribreite by simultaneous shifting to the corresponding gain factor (resistor R _11, R ..- or R ₁ O zusaimen with the corresponding capacitor (C _11, C ₁₂ or C _13). The switch planes 48oB , 48oC and 48oD are also used to apply the desired precharge voltages to those capacitors that are not currently used in the circuit. This is effected by the amplifiers 421 foew to 426. In detail, the positive terminals of these saäas amplifiers are coupled to a potential, the

5.1 V below the voltage across the positive blade of each of three capacitors C ₁₁ , C ₁₂ and C ₃ . The negative input wire of the amplifier 422, 424 or 426 is connected to the potential on the negative layer of the relevant capacitor C ₁ , C ₁ ″ or C ₁₃ , which is currently working in the circuit. Since the outputs of amplifiers 421, 423 and 425 are coupled to the negative input terminals of amplifiers 422, 424 and 426, respectively, it can be seen that the continuous mode change signal component in the output of amplifiers 422, 424 and 426 is canceled and is not applied as a precharge voltage.

An operational example is given below:

- 29 -

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-25 "-

Assume again that the circuit operates in the width loop mode, ie with R ... (open circuit) and capacitor C ₁₁ . As described above, switching to the middle loop width would require an initial voltage across Q. (the new capacitor in the circuit) that is 1o / 3 (= 3.3) times the value that was applied across C ₁₁ immediately before switching was. It can be seen that in this situation a gain control factor of 3.3 is applied to amplifier 423 via switch level 48oF. If instead a switchover to the narrow loop width were intended, the resistor R would switch to the

13 circuit itself cause the input voltage at the amplifier 505 to drop to a tenth of the value immediately before the switchover. Accordingly, the gain control factor applied to amplifier 425 (which affects the pre-charging of capacitor C _1. , Which in this case would be switched into the circuit) has a value of 10. The other gain control factors for the amplifiers 421, 422 and 423 can also easily be recalculated as being suitable for the respective situation.

Another feature of the improved carrier location loop (e.g. Block 12o of FIG. 3) according to a further aspect of the present invention shall be discussed below. 11 illustrates a conventional one Carrier location loop circuit. It is obvious that they are the circuit 8 corresponds to, but without the signal loss detector 2o6, loop width control 2o5A and filter 3oo with variable loop width. Accordingly, the same reference numerals have been used for corresponding components used. The modulated carrier is first squared by a squaring circle 2o1 to destroy the MDdulation information. The outcome of the Squared circle 2o1 is a signal at about twice the carrier frequency and provides an input for a phase comparator 2o2. The exit of the phase comparator is coupled to a loop filter 2o3, the output of which is connected to the control input terminal of a voltage-controlled Oscillator (VCO) 2o4 is coupled. The output of the VCO is via a frequency divider (or clock divider) 2o5 to the other input of the phase comparator 2o2 laid.

- 3o -

909843/0622

If in operation in a known manner once the lock has been achieved, the phase locked loop of FIG. 11 remains locked to the carrier because of phase differences between the generated clock signal (Output from clock divider 2o5) and the received carrier cause an error signal that has the tendency to adjust the VQO frequency in the correction direction to readjust according to a detected "error". However, as explained above, the unidirectional nature of the phase modulation has been described System tends to create a problem with the operation of the phase locked loop. Because changes (in the case of data transitions) caused by a momentary variation of the frequency (in the exemplary embodiment in the direction of a lower frequency), error pulses are generated generated in the output of the phase comparator whenever a data transition occurs is present. Since the PSK-Mdulation is unidirectional (i.e. instantaneous frequency modification is always in the direction of a lower frequency - as in the example - or always in the direction of a higher frequency), these error pulses have the same internal polarity. It was found that these error pulses may have a tendency to pull the carrier loop out of frequency.

Figure 12 shows an improved carrier locator loop circuit with Means responsive to transitions in the received signal that are provided for the compensation of the signal applied to the control terminal of the VCO the difference between the nominal frequency of the carrier and the actual frequency average frequency of the received signal. In Fig. 12, the squaring circuit, the phase comparator, the loop filter, the voltage controlled The oscillator and the clock divider all have the same reference numerals as in FIG. 11.

In the embodiment of FIG. 12, the output of the phase comparator becomes 2o2 is applied to the loop filter and the VCO via a summing circuit 21o. The other input of the summing circuit 21o receives Kompensaticnsimpulse by a random pulse generator 22o. The random pulse generator 22}, which can be a monostable multivibrator, is triggered by the output from Bit transition detector 150 (FIG. 3) via line 222 and generates a short Kortpensationsirrpuls always when a data transition occurs.

- 31 -

909843/0622

- JS \ -

In this way, the effect of the error pulses discussed above can be found do not accumulate and do not cause a frequency shift in the phase-locked loop. Fig. 13 shows the waveform as an output from the Suirmierkreis 21o. The error pulses 1, 2 and 3, which have data transitions occur are compensated by the pulses 1 *, 2 'or 3 'generated by the pulse generator 22o. The net receipt for the VCO, resulting from the frequency-determining nature of the phase modulation is therefore essentially at zero.

909843/0622

Claims (1)

Schlunberger Technology Corporation ^ '^ U / U QQ Claims 1. Method for processing a phase-modulated signal that with digital informaticn was modulated by momentary unidirectional ER Increase or decrease the nominal frequency of a carrier signal in operation the digital information for bringing about a phase change, characterized in that, that the modulated carrier signal is filtered with a filter having a passband center frequency which is opposite the nominal carrier frequency is offset in the direction of unidirectional increase or decrease in frequency, and that the digital information from the filtered signal is recovered. 2. The method according to claim 1, characterized by a frequency offset of the center frequency versus the nominal carrier frequency by an amount that is a function of the bit rate of the digital information. 3. The method according to claim 2, characterized in that the amount the frequency offset is equal to half the bit rate of the digital information. 4. The method according to any one of claims 1, 2 or 3, characterized in that that the passband characteristic of the filter is made asymmetrical in the same direction as the frequency offset. 5. Method of processing a phase modulated signal that modulates with digital information by momentary unidirectional either lowering or increasing the nominal frequency of a carrier signal as a function of digital information to effect a phase change by the step: filtering the modulated carrier signal with a filter with a pass characteristic that is asymmetrical in direction is the unidirectional decrease or increase in frequency, and Recovering the digital information from the filtered signal. _ ₂ _ 6. Method of reducing transient signals originating from the Inclusion of various capacitors in a circuit for use in an electronic circuit system comprising: a pair of terminals, a plurality of capacitors, each with one of its pads attached one of the terminals is coupled, a switch arrangement for the coupling of the other layer of a selected one of the capacitors to the other terminal and circuit components with variable gain, synchronized with 'fer switch arrangement for influencing the potential of the other clergy, characterized by the following steps: continuous generation of a reference voltage, assigned to each of the capacitors not connected between the terminals, each of the reference voltages generated being a function of the voltage that would appear across the associated capacitor in the event that when this - 'capacitor from the switch assembly immediately between the pair would be switched by terminals, and continuous application of the generated reference data to the assigned capacitor. P *. L>. A method of stabilizing a carrier location loop used in connection with an apparatus which receives a phase modulated signal that has been modulated with digital information by instantaneously unidirectional either lowering or increasing the nominal frequency of a carrier signal as a function of the digital information to effect a phase change, wherein The device contains a carrier locating loop circuit for locating the carrier of the received signal, which carrier locating loop circuit contains a regulated oscillator with a control terminal, and the frequency of the oscillator is stormed by a signal that can be applied to the control terminal, with a comparator also being provided for generating a control signal by comparison the phase of a signal derived from the received phase modulated signal with the phase of a signal derived from the output of the controlled oscillator, and wherein circuit components are provided for the A Applying the control signal to the control terminal of the oscillator, characterized by the following steps: Generation of compensation pulses depending on transitions in the received signal and application of the compensation pulses to the control terminal for the purpose of calculating the difference between the nominal frequency and the mean frequency of the received signal, originating from the unt 909843/0622 directional type of carrier modulation. β. ^. Arrangement for the reception of a signal phase-modulated with digital information to recover the digital information from the modulated signal, which has been modulated with the digital information by Moltentanes unidirecticnales either lowering or increasing the nominal frequency of a carrier signal as a function of the digital information to cause a phase change, which arrangement a filter for the selective filtering of the modulated carrier signal, characterized in that the filter has a passband center frequency which is offset from the nominal carrier frequency in the direction of the unidirectional decrease or increase of the frequency. β. ί. Arrangement according to Claim 8, characterized in that the center frequency is offset from the nominal carrier frequency by an amount which is a function of the bit rate of the digital information. y &. 9. Arrangement according to claim 9, characterized in that the center frequency is offset from the nominal carrier frequency by an amount equal to half the bit rate of the digital information. ;H. / C ". Arrangement according to Claims 8, 9 or 1o, characterized in that that the passband characteristic of the filter is made asymmetrical in the same direction as the frequency offset. Y Z. λ \ <assembly of claim 8 for use in a well drilling and Msßvorrichtung for performing subterranean measurements during the low bringing the wellbore in a fluid-filled borehole, and for the Koitmurikation of the measured values to the surface, with a downhole measurement and transmission assembly including in a drill string transducers having means for generating acoustic carrier waves at a nominal frequency in the borehole fluid, with means for phase modulating the generated acoustic carrier waves in accordance with digital data representing the measured values by momentary unidirectional either lowering or increasing 909843/0622 the frequency of the acoustic carrier signal and with one at the earth's surface located receiver assembly with the filter and with converters for converting the modulated acoustic carrier wave into electronic signals, characterized in that the center frequency of the filter compared to the Nominal carrier frequency is offset by an amount that is a function of the Bit rate of the digital information and that means are provided for extracting the digital data from the filtered electronic signals. > 3. \ Z 'Arrangement for the demodulation of a digital signal, to which a signal phase-modulated with digital information is applied and which serves to extract the digital information, wherein the modulated signal with the digital information by momentary unidirectional either lowering or increasing the nominal frequency of a carrier signal as a function of the digital information has been modulated to cause a phase change, after which the carrier is returned to the nominal frequency after the phase change has been effected, which arrangement includes a filter for selective filtering of the modulated carrier signal, characterized in that the filter has a passband characteristic which is asymmetrical in Direction of unidirectional decrease or increase in frequency. 14. Electronic circuitry with a pair of terminals, one A plurality of capacitors, each of which is coupled with one of its pads to one of the terminals, with a switch arrangement for the coupling of the other layer of a selected one of the capacitors to the other terminal and with variable gain Kcpigionenten, synchronized with the switch arrangement, to influence the-'iotentials at the other Kenne, which circuit arrangement is characterized by a circuit for the Reduction of transition signals' resulting from switching another Capacitor between the ^ Kleitnienpaar, with a circuit for the permanent Generation of a low voltage, assigned to each of the plurality of capacitors, which are not currently coupled between the pair of terminals, each such reference voltage being a function of the voltage across the assigned capacitor would occur if the assigned capacitor 909843/0622 sator instantly from the circuitry between the pair of terminals would be switched, and. with switching circuits for continuous application each reference voltage generated in this way to the associated capacitor. 15. Circuit arrangement according to claim 14, characterized in that dadupcn the circuits for generating a reference voltage are appropriately designed are responsive to the voltage just across the pair of clamps and are also designed to be a ratio of gain factors of the circuit components with variable gain. 16. Circuit arrangement according to claim 15, characterized in that the circuitry for generating a reference voltage a plurality of amplifiers encompass, each of which is designed responsive to the instantaneous tension between the pair of terminals and in terms of a gain is regulated according to a ratio of the gain factors. 17. Circuit arrangement according to claims 14, 15 or 16, characterized that the positive surface of each of the capacitors is coupled to one terminal of the pair of terminals and this one terminal to a positive one Tension is maintained. . Al, carrier location loop with variable loop width for locking onto the carrier of an input signal with a phase-locked loop which has an oscillator with a control input, with an error signal generator for generating an error signal as a function of the phase difference between a signal derived from the oscillator and the input signal and having a variable filter with a plurality of different bandwidths for coupling the output of the error signal generator to the control input of the oscillator, characterized by loop width control circuits, coupled to the variable filter for automatically changing the loop width of the filter as a function of the input signal. . , <<? - carrier location loop according to claim 18, characterized in, §09843 / 0622 that the variable filter comprises a plurality of capacitors operating or are switched out of operation in the filter under control by the loop width control circuit, the variable filter having circuit components for the continuous pre-charging of those capacitors, which are not currently in the filter mode, in order to prevent the loss of the lock in the phase-locked loop when the disconnection by the Loop width control circuitry is made. J2o. / t, - carrier location loop according to claim 18, wherein the variable filter comprises: a first amplifier having first and second input terminals, of to which the first input terminal B has an input signal applied to it, one second amplifier, a variable gain control circuit for switching through coupling of the output of the first amplifier an input of the second amplifier, the variable gain control circuit has at least first and second different gain actuators, variable capacitance components with at least first and second Capacitors, one of which is shared with the variable gain control circuit switchable is marked for the capacitive coupling of the output from the first amplifier to the second input terminal of the first amplifier by: circuitry for generating a reference voltage, assigned to the capacitor which is currently not effective in the variable capacitance components is which reference voltage generated is a function of the voltage that would appear across the capacitor, which is currently not in operation, if it would be put into operation instantaneously, and by circuitry for continuously applying the generated reference voltage to the capacitor, which is currently not used in the variable capacity components. £ 1. // £. Carrier location loop according to Claim 2o, characterized in that the circuitry for generating a reference voltage is responsive to the voltage above the current in the variable capacitance component in operation capacitor and is also designed responsive on a ratio of the gain factors. - 1 - 909843/0622 22 .. Ό-. Carrier locating loop according to Claim 2o, characterized in that the circuit for generating a reference voltage comprises first and second amplifiers, each associated with the first and second capacitor, respectively, the amplifier associated with the capacitor that is currently not in use being designed to respond to the voltage across the capacitor that is currently in use. , 23. \%. · Carrier location loop according to claim 22, characterized in that each of the amplifiers can be regulated with regard to its amplification according to a different ratio of the amplification factors. JM. Wt arrangement for receiving a signal phase-modulated with digital information, which arrangement is used to recover the digital information from the modulated signal, which signal has been modulated with the digital information by either lowering or increasing the nominal frequency of a carrier signal as a function of the digital information to bring about a phase change which arrangement comprises a carrier locating loop circuit with a controlled oscillator having a control terminal, the frequency of the oscillator being determined by a signal applied to the control terminal, with a comparator circuit for generating a control signal by comparing the phase of a signal derived from the received modulated signal with the phase of a signal derived from the output of the regulated oscillator, and with circuitry for applying the control signal to the control terminal of the oscillator, characterized by circuitry se that are designed to respond to transitions (jumps) in the received signal for the compensation of the signal applied to the control terminal with regard to the difference between the nominal frequency of the carrier and the mean frequency of the received signal, resulting from the unidirectional type of carrier decoding. , 25. ZCt circuit arrangement according to Claim 24, characterized in that the compensation circuits contain circuit components for generating a pulse for each data jump of the received signal and circuit components for adding the generated pulses to the control signal. 909843/0622 Zi has' circuit according to claim 24 for an apparatus for thtersuchen vcn holes during its sinking of for draining underground Msssungen in a fluid-filled borehole, and for the communication of ffeßwerte to the ground surface, with a downhole measurement and transmitter module, which: an assembly attachable to a drill string for deriving the Measurement information, means for generating acoustic carrier waves at a nominal frequency in the borehole fluid, a device for the phase modulation of the generated acoustic Carrier waves according to digital data that are representative of the measured values, by momentary unidirectional either lowering or increasing the frequency of the acoustic carrier signal, and with a receiver assembly located on the earth's surface, which has: Traveling circuits for converting the modulated acoustic carrier waves into an electronic input signal and a carrier locating loop circuit that includes the regulated oscillator, the comparator, the apply circuitry and the compensation circuitry characterized in that the compensation circuits are circuit components comprise circuit components for adding for generating a pulse at each data transition of the received signal scwie of the generated pulses to the control signal, such as circuit components for recovering the digital data by demodulating the input signal with a signal derived from the output of the controlled oscillator. 909843/0622