DE2036475C - Phase coherent receiver circuit - Google Patents

Description

The invention is concerned with a phase coherent Receiver circuit, consisting of a parametric preamplifier and a downstream Phase detector, with the help of which the difference between the phase of the input signal of the circuit and the phase of the reference signal of a voltage-controlled oscillator and formed via a control filter this is fed for readjustment. This reference oscillator is made by frequency multiplication the pump frequency for the parametric preamplifier is derived.

The invention and the advantages achieved thereby are intended to be based on an exemplary embodiment below Reference to the figures are explained in more detail below.

F i g. 1 shows the basic circuit diagram of a known phase-coherent receiver with a parametric Preamplifier;

F i g. 2 shows the circuit of a quasi-degenerate or degenerate non-reciprocal parametric Preamplifier;

F i g. 3 shows the directions of the currents and voltages across the reactance diodes of the parametric Preamplifier;

'F i g. Fig. 4 shows the circuit of the up-converter including the noise sources;

F i g. 5 shows the chain connection of up and down mixers;

F i g. 6 shows the neutralization of this mixer chain;

F i g. 7 shows the entire amplifier quadruple with the noise inflows at the input and output;

F i g. 8 shows the amplifier quadrupole with the total inflow of noise at the input.

In FIG. 1 shows the block diagram of a phase-coherent receiver circuit as it is used, for example, in satellite communication systems. With the aid of a phase detector, the difference between the phase of the input signal U _s and the phase of a voltage-controlled oscillator U _{p0 is} formed in this circuit, which is fed back to the oscillator via a control filter and readjusts its phase or frequency. If the phase q _{s of} the input signal corresponds to the phase Ψρο of the reference signal, the control signal U _r disappears at the output of the control filter, whereby the stable point of the control system is given. The voltage controlled oscillator serves as the basic oscillator; with the aid of a frequency multiplier, the pump frequency signal U _{p is} derived from it, which signal is fed to the parametric preamplifier as a pump voltage.

The invention is based on the object of this known phase-coherent receiver circuit to further develop that it has an increased signal-to-noise ratio, thereby increasing the safety of such Circuits is improved or the range for a secure message transmission is significantly is enlarged.

According to the invention, this is achieved in that, in the case of the initially described phase coherent Receiver circuit of the parametric preamplifier is not designed reciprocally and has a pump frequency which is equal to twice the signal frequency, and that the frequency multiplier as a frequency doubler works.

In a preferred embodiment of the invention this is a non-reciprocal parametric preamplifier from a known (German Aus Legeschrift 1 120 525) chain connection of an upward and down-converter formed, the diodes of which are switched through in different phases by a pump source will; the feedback conductance of the amplifier, the overall circuit of which is shown in FIG. 2 shown is is neutralized.

In the following, an attempt will be made to determine the degree of improvement which is achieved by the inventive ße Using the described non-reciprocal or degenerate parametric preamplifier results in to be recorded arithmetically.

The circuit

of the quasi-degenerate non-reciprocal parametric amplifier

First, the circuit of the quasi-degenerate non-reciprocal parametric amplifier should be considered, from which one can then move on to the degenerate case. In Fig. 2 shows the circuit of the amplifier. It contains a signal circuit at the input and output of the mixer chain, ar which the signal generator or the load are ruled out. If the frequency of the lower sideband: ρ - s is close to the signal frequency s, the signal circuits at the input and output can simultaneously serve to tune to the frequencies of the signal unc of the lower sideband. The connection between the parametric up and down mixer is then only represented by the parallel circle at the upper sideband p + s between the terminals 1 'and. The voltage of the pump frequency p, which is used for the phase-coherent operation with the help of a frequency multiplier from the reference oscillator

is to be conducted, the two reactance diodes assumed here as ideal with different Phase fed via an external phase shifter. In case that

^Δ Ψ = Ψρ, \ - 9> ρ, 2 = ± ^ 5-

is set, the then phase non-reciprocal mixer chain can be controlled by a purely imaginary neutralization conductance j ^ between the input and output terminals. Completely decoupled in the reverse direction

will. The conversion matrix of the quasi-degenerate build-up or

Down mixer

For the in F i g. 3 directions of the currents and voltages on the ideal reactance diode the conversion equations apply:

i;

l'p * s

jsO ^l) J (ps) C ^ll) * -j (ps) Ö ⁰⁾

C ⁽¹ > 0 j (p + s) C ^l0]

W- *

(2-1)

where the terms with C_ ^{a) have been} neglected. If, according to FIG. 4 adds the signal circuits at the input and output, as well as the uncorrelated noise inflow caused by the loss conductance values at the frequencies sup ± s, this is how the abbreviations give

y; + jsO ⁰⁾ = (2-2)

the conversion matrix of the quasi-degenerate upward or downward mixer including the noise inflows to:

Ip ₊ S

Ys jsC "

-j (ps) C ^l) * y "* _ _s -j (p + s) e ^u ο

The derailleur
of up and down mixer

By connecting up and down mixers in a chain as shown in FIG. 5 result on the 40 where Terminals / 'and i the conditions:

-jsÖ ^u * Us + L. 0 Up * -s Yp ₊ S Up _{+ S} _ Jp + S_ Ip + s.l + lp + s.2

(2-3)

(3-2)

X + s = X + sA +

Lp + sA - -lp

see 2

Llp + sA ~ l ± p

was set.

(3-1) For the tensions at the lower sideband on the

45 The input and output of the amplifier are obtained from the second line for the up and down mixer. The currents, voltages and conductance values of the on condition:
forward mixers are with the index 1, those of the
Down mixer designated with the index 2. the
Equation (2-3) applies to both the upward alj and 50
Door the down mixer.

With the conditions for the ladder connection according to equation (3-1), the third line gives the Equation (2-3):

U * - _S i =

Mp-s.2 = J ip-s.l - l * -s.2 -

ψ-s.i

, (3-3) • (3-4)

Vp + s - J

If you put the voltage U _{p ± s} in the first line (3-2) of the upward or downward mixer, you get

the via pole equations of the non-neutralized mixer chain. These equations are:

+ .s) IC ₁ '"1 ² .s- (p - s) ! C ₁ '" I ²

.v) C ₁ '"* Cj"

y _{p + s}

^u > - ²

(3-5)

. .SC ₁ ¹ "* _r . .- .. .SC, ¹ ". "

Ln + s Lp-S.

I _ ^ ^Ti ^ i ^ ' J /, IV + * (P + S) | C ₂ "' | ² S (p - S) ι W ₂ ; - ι ι

's.2 - y ys1 H- is. 2 + y TTi 1 · Η.ν.2

ip + s [ ± p + s Lp-s.2

sC ^ ^] * SC ₂ ⁾

Η "is. 2 +. / ^ [Ip + s. 1 + ip + s, 2] ~. / ^ I Ip-s. 2 ·

ip + s Lp-s.2

The signal properties _γ _ s (ρ + s) C ₁ "' * C ₂ '"

of the neutralized quasi-degenerate amplifier ^l0 -12- y ^ j ^ r,

The signal properties of the quasi-degenerate sic + slC "* C ⁽¹⁾

Amplifier is obtained with the help of the four-pole constant Y ₂₁ = —— '—-1 ! / j £

th, which upon neutralization according to the> ^

^ f ^ ^F 'S ⁶ JP! follow equations (3-5), 5. At resonance and with and (3-6) for the noise-free case ι ,, i _{p ± s} = 0 to:

_ Iy- (I) I J'n I / - (1) 1 ^ (1) 1 J? '_ T I

/ ^ d) _ Iy- (I) I J'n I / - (1) 1 ^ (1) 1

_20th

s (p - s) | C, ⁽¹⁾ | ² and the abbreviations

J22 = X.

% - ° · ¹ (4-1) _G _ S (P ₊ S) IC '"! ²

Gp-.i = G. + C. + C, (4-3) and

5? -S.2 ' _3o ^G ps.2 = G _L + G _C , _{2 +} G ₀ . (4-4)

The equations (4-1) for the quasi-degenerate case s χ ρ - s result in the following equations:

L, = [g. + Gc- ,, + G _D -

According to equation (4-5), for G ₀ = %, expressed in reverse and with the four-pole constants: in the downward direction, complete decoupling is achieved. for
this case follows from equation (4-5) with

G, = G _L , Gc ₁ = G _c . ₂ = G _c ** - IjIJ ^ P "

and the abbreviation

j _C (i) i For the neutralized quasi-degenerate

ß = _ (4-6) 50 stronger is then obtained with the four-pole constants

G _s + G _c + G _{D according to} equation (4-7):

the equation:

i, -Bf _{1 +} Oc ₊ GJD- ^ H. HT) _j5

L.2 = -J ² GdM ^ + [G _s + G _c + G _D [[1 - i? ² ] I / _sJ .

The Uberfaeuneseewinn ^The transfer profit in the reverse direction

des qS ^ d £ ^? VMikm _fc results in the same way as equation (4-9)

The transmission gain of the amplifier in the forward direction is due to the ratio of the active power at the load conductance and the available active ^ _. _ß " U _sl _ 4G _s G _t | ^ ₂ | ² · power of the signal generator given. You get ^ar * ^Ul J ^ ₂ ~ [Y _n J ₂₂ -Y ₁₂ Y ₂ J ^{2 i4} " ³¹ ^ for general: 65

= 4G _s G _t

(4-8)

Because G ₀ = %, ie Y ₁₂ = 0, L _ür = 0.

The noise characteristics of the quasi-degenerate amplifier

To calculate the noise inflow at the input and output of the quasi-degenerate amplifier, set UsA = -L4.2 = 0 in equations (3-5) and (3-6) and get:

The mean values of the noise current squares of equation (5-7) are given by the Nyquist relation

[T ^] ² = AKT (G ₅ + G ^ ₁ ) Af; O ² = AK.TG _c . ₂ Af

Il - IsA + J ~~ y "" - Lip + j.ι + Jp +,. 2 y

(5-1)

= L

, 2

ίίρ + sA + ip + s.2]

15 or

lip * -,, I ² = A KT [G, + G ^ ₁ ) Af;

il ^ ² = AKTG ^ ₁ Af

Up ₊ .,., I ² = AKTG ^ ₁ Af;

\ ϊ ^ \ ² = 4XTG _{p + s} , ₂ / l /

(5-8)

p ₊ .,. + Zp ₊₅ .21 ² = 4KTG _{p + s} Af

4 - p.2

J.J - ,. 2 ·

For the circuit according to FIG. 7 the relationships apply

(5-2) given.

With the conditions

20th

L ₁ = III! / .. I +1.2 Ex.2 + J1

(5-3) ,, ι

from which one can derive the total inflow of noise at _{and the relationships} at the input of the amplifier to ⁶

t ^{k (5} " ⁴⁾

receives, the door that is now shown in FIG. 8 the equations are: 35

Vp. 1 ~ <pp. 2 - - ~ 2 '

.3-Gp-s. ι = G _p _ _{s 2} = G _s + G _c + G _D

η
Yn Gf "* _ (G ₅ + G _c + G _D ) (1 - ß ² )

γ r <"*

± 21 kl

2Gr

(5-9)

IsA = Iu UsA + V, 2Ü.2 + Χ «is.2 = YlI UsA + Γ22 Ü.2 ■

(5-5)

Füri ", thus follows from equations (5-1) and 40

G ² _{p + S} G _{p + S} p + s

as well as β according to equation (4-6) one obtains:

IU ²

AKTAf

-G + C ι (Gs + G _c + G _D ) ² (1- /? ² ) ²

- «, +« c. 1 H 77 ^ 2 G _r . 2

ill. . * »V ^ J. _Ä

hot - ls.i γ is, 2 Jy ⁺ P ~

Ä2 \ Jp-s. ί

(5-10)

sd "

Lp-s.2

. SC} "*

The noise figure of the quasi-degenerate amplifier!

and thus for the mean value of the total noise current ^{is then 1} ^ ^the noise current square of the signal square with resonance tuning of the amplifier: source

[^ | 2 ₌

as a reference

(YOU

_t + G _D + G _c ) ² { 1- 4Eq

(5-7)

⁶⁵

{5-11,

If the signal circuit losses are neglected, the chain connection of up and down mixers

for the degenerate case
G ₁₁ = G _C2 = 0

For the chain connection of the degenerate up and down

one obtains the simple downward mixer from equation (5-11), the conditions according to apply again Relationship: Equation (3-1), which then results in the corresponding

Treatment as in Section 3 of the four-pole equations

(5-12) ^I0 L Yis U _sA J

those for large gain β - * 1 and G ₀ <G ₅ den, 3, S ² C ₁ "'* Cj"

Value Y _{3 s} ^y ' ²

Fs2 = 3db (5-13) (6-3)

accepts. ι _ ^JS ^ i

iV 2 - y

The conversion matrix of the degenerate - ^3v

Up or down mixer Compared to the quasi-degenerate case where

L " ² > 3 _S ^{JS 2} U _x . 2 J ~ ⁵ * ²

s - d % s result.

was and the signal voltages U _x , U _p ^ _s , U _{p + s} das

controlled nonlinear element, applies now in the degene- - «· '_ _{= e} ^ ² '. _v . 1

rieilen case: 25 Li. 1

s = ρ - s or ρ = 2s I ₇ *

/, _ and l / ₊ ; = U ₃ ° "'

U ₉ = (/, _, and l / _{p +} ;. = U ₃₅ ^l ° "'ii.2

For the noise-free case with [ _s = ^ ₊₅ = 0, the conversion equations (2-3) for the degenerate upward or downward mixer are:

Ic ₁ ¹¹¹ I = | c «'|

_{35 and G3 according to equation (4. 2) follow from equation} .

'' gen (6-3) with resonance tuning of Y ₁ ; Y-

/ j, = - j 3 _S C ⁽¹ > U ₅ +% _s U _3s . and χ ,: '

(6-4)

Ui =

/.,.2= G _D e '(VrV2)

If you go again

II

and neutralize the mixer chain with a purely imaginary conductance value Y _n according to Figure 6, it follows from equations (6-4) for resonance tuning and neutralization with G ₀ = Y _n :

(6-5)

The maximum negative conductance at the amplifier input is obtained from the phase condition

- (6-6)

and therefore because of U ₂ = ° from the second line of equation (6-5)

2G _D | t / .., | ;

(G ,, + G ₁ -. ₂ + G ₀ HU ₄ , ₂ 1

the condition

-(H

(6-7)

the for

G ₄ . = G ,. and G _r] = G, - ₂ = G ₍

with β according to equation (4-6) the value
'/.V.2 - VV. = arccotg / i value. With the phase conditions according to equations (6-6) and (6-8), as well as for

G ₄ = G ,. and G /,., = G ₍₂ = G ₁ follows from equations (6-5):
LsA = [G ,, + Gf + G ₀ ] [I - / Z] U ₄ ,,

(6-10)

(6-8) _{^"5/4, 2} = - / 2G _n U ,.

_ -y 2 (.rc colg / l- ~

accepts. For a large gain with ß - * \ it follows + [G ₄ . + G, - + G "] [l -ß from equation (6-8):

over transmission gain _ ₍ _ | _ Jj_ (6_9) ^{20 of} the degenerate amplifier

With the help of the four-pole constants from the equation

In this case, the negative Wirkleit- (6-10) is obtained for the transmission gain of the value at the output of the amplifier is its maximum degenerate amplifier according to the equation (4-9):

Windward (leg- = -

IG, + G _c + G ₀ T U-If] ² _ -J 2 (.n: c «g / I -Ji

1 - / ie

and with

(6-11)

arc cotg β % -j-

^; ur fi values close to one:

whereby

a -. (6-12) Pr.2 _ HÜ2 ² windward quasi

'r. 1 UsI

The improvement that is thus obtained through the phase-coherent operation is through the ratio the transfer gains between the degenerate and quasi-degenerate case by the drawing:

40 = Fquasi · Windward quasi (6-15)

represents the ratio of the power of the uncorrelated noise signals at the output and input. However, the transmission of the power of the correlated useful signals is due to the relationship

45

Lüvdeg. ₌
Windward, so to speak

(1 -

= (1 + ßf (6-13) windward dcg.

(6-16)

given according to equations (4-10) and (6-12). For large gains with ß- * i an improvement of V = 16 = 12 db is obtained.

The noise figure

of the degenerate amplifier

When calculating the noise behavior of the

degenerate amplifier care must be taken that

Noise signals uncorrelated, useful signals but corrected

be reinforced. The usual methods, which only take into account the noise contributions, therefore fail. One must now start from the definition equation of the noise figure, which is based on the ratio of the signal-to-noise ratios at the input and output of the amplifier
The following applies:

given. One then obtains from equations (6-14), (6-15) and (6-16) for the noise figure of the degenerate Amplifier the connection:

Fdeg. =

Fquasi Luv quasi _ Fquasi Luv deg. V

(6-17)

55 With Fquasi according to equation (5-11) and V according to equation (6-13) it then follows:

60 Fdeg. =

l + ß ²

(6-18)

Fdeg. =

PsA P ,:

(6-14)

P, 2 ' 3 G ₁ (I + Pf

According to equation (6-18), the noise figure of the degenerate amplifier has become smaller by the Fauqr V = (I + pf than the noise figure of the quasi-degenerate amplifier.

The signal-to-noise ratio at the output of the amplifier has therefore been significantly improved compared to the signal-to-noise ratio at the input of the amplifier due to the phase-coherent operation. For large gain with ß - * \ and for G _D <G _{s it} follows from equation (6-18) for

Fdeg. as 1.8 === -9 from. (6-19)

This means that a phase coherent receiver circuit implemented according to the invention is particularly useful is suitable for the transmission of messages with spacecraft, since the sensitivity of the Receiver for the range and security of the message transmission is of crucial importance is.

For this purpose 2 sheets of drawings

Claims (3)

Patent claims: 1. Phase coherent receiver circuit, consisting of a parametric preamplifier and a downstream phase detector, with the help of which the difference between the phase of the Input signal of the circuit and the phase of the reference signal of a voltage controlled oscillator formed and fed through a control filter to this for readjustment and that of this The oscillator derives the pump frequency for the parametric preamplifier by frequency multiplication is characterized in that the parametric preamplifier is not reciprocal is formed and has a pump frequency which is equal to twice the signal frequency and that the frequency multiplier acts as a frequency doubler. 2. Circuit according to claim 1, characterized in that that the parametric preamplifier consists of the daisy chain of an up and down mixer, the diodes of a Pump source are controlled in different phases, and that the feedback conductance of the amplifier is neutralized. 3. Circuit according to one of the preceding claims, characterized in that it is application takes place for communication with spacecraft.