DE102022106791A1 - MIMO RADAR DEVICE AND MIMO RADAR METHOD - Google Patents

Abstract

The present disclosure relates to a MIMO radar device that includes a transmitter circuit with a plurality of transmission channels. The transmitter circuit is designed to transmit a first sequence of FMCW radar chirps via a first transmission channel such that the start and/or stop frequencies of each of at least a portion of the FMCW radar chirps of the first sequence are different from one another. The transmitter circuit is designed to transmit a second sequence of FMCW radar chirps via a second transmission channel, the start and/or stop frequencies of each of at least a portion of the FMCW radar chirps of the second sequence being different from one another. The first and second sequences of FMCW radar chirps are transmitted simultaneously. The control circuit is configured to control the first and second transmit channels to adjust phases of the FMCW radar chirps of the first sequence and the second sequence according to a predefined Doppler Division Multiplex, DDM, scheme.

Description

Area

The present disclosure relates to radar systems and in particular to MIMO (Multi Input Multi Output) radar concepts that employ multiple transmit channels, one or more receive channels and multiple waveforms.

background

Automotive radars, along with other environmental sensors such as lidar, ultrasound, and cameras, are one of the backbones of self-driving cars and advanced driver assistance systems (ADASs). These technological advances are made possible by complex systems with signal processing paths from radars/sensors to one or more controllers. Automotive radar systems enable the detection of objects and obstacles, their position and speed relative to a vehicle. The development of signal processing techniques along with advances in millimeter wave (mm-wave) semiconductor technology plays a critical role in automotive radar systems. Various signal processing techniques have been developed to provide better resolution and estimation performance in all measurement dimensions: range, azimuth/elevation angle and speed of the targets surrounding the vehicles.

For example, frequency-modulated continuous-wave (FMCW) radar systems are known to obtain range, speed, and angle information by performing multiple Fast Fourier Transforms (FFTs) on samples of radar mixer outputs . A first FFT, also commonly referred to as a range FFT, provides range information. A second FFT over the distance-transformed samples, also commonly referred to as a Doppler FFT, provides velocity information. The first and second FFTs result in a so-called 2D range/Doppler map including range and velocity (FFT) bins. A third FFT, which includes phase information of signals from different antenna elements of a (virtual) antenna array, can yield additional spatial or angular information - so-called direction-of-arrival (DoA) information.

MIMO (Multiple Input Multiple Output) is widely used to increase the effective radar aperture size by synthesizing a virtual receiver array through a combination of physically implemented multiple transmitter channels and/or multiple receiver channels. To synthesize virtual array information from limited physical arrays, separation of reflected signals received at each receiver channel from different transmitters is an important procedure in MIMO technology. Equally important is the identification of the corresponding transmitters of each reflected signal in each receiver channel.

There is therefore a need for the largest possible virtual array synthesis with high resolution and good identification of corresponding transmitters.

Summary

This need is met by MIMO radar devices and methods according to the independent claims. Further advantageous embodiments are addressed by the dependent claims.

According to a first aspect, the present disclosure proposes a MIMO radar device. The MIMO radar device includes a transmitter circuit that includes a plurality of transmission channels. The transmitter circuit is designed to transmit a first sequence (or a frame) of FMCW radar chirps (or ramps) via a first transmission channel. The start and/or stop frequencies of each of at least a portion of the FMCW radar chirps of the first sequence are different from each other. Furthermore, the transmitter circuit is designed to transmit a second sequence (or a second frame) of FMCW radar chirps via a second transmission channel. The start and/or stop frequencies of at least a portion of the FMCW radar chirps of the second sequence are different from each other. The transmitter circuit is designed to transmit the first and second sequences of FMCW radar chirps simultaneously (simultaneously). The MIMO radar device further includes a control circuit configured as the first and control the second transmit channel to adjust phases of the FMCW radar chirps of the first sequence and the second sequence according to a predefined Doppler Division Multiplex, DDM, scheme.

Typically, the start and stop frequencies of each FMCW radar chirp within a chirp sequence are identical. However, embodiments of the present disclosure suggest varying start and/or stop frequencies within a chirp sequence. This allows the effective modulation bandwidth to be increased to achieve higher range resolution at longer distances.

In DDM, the FMCW radar chirps transmitted simultaneously from different transmission channels are modulated by different phase sequences, so that the respective received signals can be separated/decoded in a receiver based on the respective phase sequences. In embodiments, the respective phases of the FMCW radar chirps of the first sequence and the second sequence adjusted according to the DDM scheme may be maintained unchanged for a complete FMCW chirp period (interval), while for the following FMCW Chirp period, a respective new phase can be set according to the DDM scheme and maintained unchanged for the complete FMCW chirp period, etc. Thus, the DDM scheme can be a chirp-wise DDM scheme in which a transmit channel-specific phase adjustment can be made via the The entire period of the FMCW chirp is retained and can then be changed for the next chirp on a transmission channel-specific basis. DDM is sometimes referred to as code division multiplexing (CDM). However, those skilled in the art who take advantage of the present disclosure will recognize that embodiments are not limited to first and second broadcast channels. Any number of broadcast channels is possible. In particular, three transmission channels are suggested in some exemplary embodiments.

In some embodiments, the control circuit of the MIMO radar device is configured to control the first and second transmission channels such that the respective FMCW radar chirps of the first and second sequences have a same respective duration and/or a same respective frequency bandwidth between the respective ones Have start and stop frequency. Thus, an FMCW radar chirp of the first sequence and a simultaneously transmitted FMCW radar chirp of the second sequence can have the same duration and/or the same bandwidth. In particular, an FMCW radar chirp of the first sequence and a simultaneously transmitted FMCW radar chirp of the second sequence can have the same respective start and stop frequencies. In other words, the control circuit may be configured to control the first and second transmission channels to cause respective simultaneous FMCW radar chirps of the first and second sequences to have the same start and stop frequencies. However, the respective start and stop frequencies may change within or during a sequence of FMCW radar chirps.

In some embodiments, the control circuit is configured to control the first and second transmission channels such that the start frequencies of each subsequent pair of subsequent FMCW radar chirps of the first sequence differ by a predefined frequency offset and the start frequencies of each subsequent pair of subsequent FMCW -Radar chirps of the second sequence differ by the same predefined frequency offset. Thus, the starting frequency of a subsequent FMCW radar chirp in each of the sequences may increase or decrease by the predefined frequency offset with respect to a previous FMCW radar chirp. In other words, the control circuit may be configured to control the first and second transmission channels such that the starting frequencies of each subsequent pair of subsequent FMCW radar chirps of the first and second sequences increase or decrease by the predefined frequency offset. This can increase an effective frequency modulation bandwidth and thus the achievable distance resolution.

According to the predefined DDM scheme, the control circuit can be designed to assign each transmission channel a unique sequence of phases, which is applied to the respective sequence of FMCW chirps of the respective transmission channel. Thus, each transmit channel has its unique dedicated sequence of phases. On a receiver side, different transmission channels can then be identified based on the respective transmission channel-specific sequences of phases. As previously explained, a phase applied to an FMCW chirp can be understood as an initial phase offset that remains constant throughout the duration of the FMCW chirp. However, the phases (or phase offsets) of the different FMCW chirps within a sequence of FMCW chirps may vary.

In some embodiments, the control circuit is configured to select the phases for the predefined DDM scheme from an M-ary phase modulation alphabet, where M ≥ 2 is an integer. This means that the different phases for the FMCW chirps consist of a finite number of M alternative phases can be selected. Examples of modulation alphabets would be BPSK (Binary Phase Shift Keying), QPSK (Quaternary Phase Shift Keying), 8-PSK and the like. BPSK in particular can be useful. This can reduce ambiguity on the receiving end.

In some embodiments, the control circuit is designed to assign a first predefined phase to the FMCW chirps of the first sequence. The first predefined phase can, for example, be a constant phase of z. B. 0° or 180° and can be applied to any FMCW chirp of the first sequence (e.g. no phase modulation of the first sequence of FMCW chirps). Furthermore, the control circuit can be designed to assign the first phase and another second phase to the FMCW chirps of the second sequence. If the first phase is 0°, the second phase can e.g. B. be 180°. If the first phase is 180°, the second phase can e.g. B. be 0°. A phase difference of 180° between the first and second phases can be useful. For example, the first phase and the second phase can be alternately assigned to subsequent FMCW chirps of the second sequence (e.g. 0° | 180° | 0° | 180° | ...). However, other phase patterns are also conceivable (e.g. 0° | 0° | 180° | 180°| 0° | 0° | 180° | 180°| ...). Simultaneously transmitted FMCW chirps of the first and second sequences have predetermined phase differences. The predetermined phase differences may include first and second predetermined phase differences. For example, the first predetermined phase difference may be 0° if simultaneously transmitted FMCW chirps of the first and second sequences both use the first phase. On the other hand, when simultaneously transmitted FMCW chirps of the first and second sequences use different ones of the first and second phases, the second predetermined phase difference may be 180°.

In some embodiments, the control circuit is designed to assign the first phase to a first FMCW chirp of the second sequence and the second phase to a subsequent second FMCW chirp of the second sequence (e.g. 0° | 180° | ...), while two consecutive FMCW chirps of the first sequence with the first phase (e.g. 0° | 0° | ...) are sent. This corresponds to a continuous use of the first phase for FMCW chirps of the first sequence and an alternating use of the first and second phases for FMCW chirps of the second sequence. In this way, the first and second transmission channels can be identified and separated on a receiver side.

In some embodiments, the control circuit is configured to assign a sequence of first phases to the first sequence of FMCW chirps. The first phases include either a first phase value (e.g. 0°) or a second phase value (e.g. 180°, where the first phase value and the second phase value differ by 180°. On the first sequence of FMCW chirps So only one of the first and second phases can be used. Furthermore, the control circuit can be designed to assign a sequence of second phases to the second sequence of FMCW chirps. The second phases include a third phase value (e.g. 0 ° or 45°) and a fourth phase value (e.g. 180° or 225°), whereby the third phase value and the fourth phase value differ by 180°. So only one of the first and second phase values can be related to the first sequence of FMCW Chirps may be applied, and both the third and fourth phase values may be applied to the second sequence of FMCW chirps. At least one of the first phase value and the second phase values may be different from each of the third and fourth phase values. Alternatively, at least one of the first or second phase values may be identical to the third phase value. Thus, in some embodiments, the different transmission channels may use different binary phase modulation alphabets.

In some embodiments, the transmitter circuit of the MIMO radar device additionally includes a third transmission channel and is configured to transmit a third sequence of FMCW radar chirps via the third transmission channel such that the start and/or stop frequencies of each of at least a portion of the FMCW radar chirps of the third sequence are different from each other. Thus, start and/or stop frequencies of the FMCW radar chirps of the third sequence can be changed in a similar or identical manner to the start and/or stop frequencies of the FMCW radar chirps of the first and second sequences. The control circuit can be designed to assign a first phase (e.g. 0°) to each FMCW radar chirp of the first sequence. The control circuit may be configured to assign a sequence of second phases to be applied to the FMCW radar chirps of the second sequence, the sequence of second phases comprising the first phase (e.g. 0°) and another second phase (e.g. 180°) in a time order specific to the second transmission channel. The first and second phases can differ by 180°. Further, the control circuit may be configured to assign a third phase sequence to be applied to the FMCW radar chirps of the third sequence, the third phase sequence comprising the first phase (e.g. 0°) and the second phase (e.g . B. 180°) in a time sequence specific to the third transmission channel. The time order specific to the third transmission channel may differ from the time order specific to the second transmission channel. Therefore, a binary DDM scheme can be used for the three transmission channels. This can lead to good performance in identifying targets.

In some embodiments, the control circuit is configured while transmitting two consecutive FMCW chirps of the first sequence with the first phase (e.g. 0° | 0°), the first phase to a first FMCW chirp of the second sequence and the second phase to assign a subsequent second FMCW chirp of the second sequence (e.g. 0° | 180°). Furthermore, the control circuit can be designed to transmit four consecutive FMCW chirps of the first sequence with the first phase (e.g. 0° 0° | 0° | 0°), the first phase of two consecutive FMCW radar chirps third sequence, followed by the second phase being assigned to two more consecutive FMCW radar chirps of the third sequence (e.g. 0° | 0° | 180° | 180°).

In some embodiments, the MIMO radar device may further include a receiver circuit with at least one receiver channel. The at least one receiver channel is designed to receive a received signal that corresponds to reflections of the first sequence of FMCW radar chirps and the second sequence of FMCW radar chirps (and optionally the third sequence of FMCW radar chirps). This means that reflections from all simultaneously transmitted sequences of FMCW radar chirps can be superimposed on a receiver antenna. In some embodiments, more than one receiver antenna (receiver channel) can be used on the receiver side.

In some embodiments, the receiver circuit is configured to perform a first FFT (range FFT) of the received signal (receiver channel) to generate a plurality of first FFT bins (range bins). The receiver circuit may further be configured to perform a second FFT (Doppler FFT) using the first bins (range bins) to generate a plurality of second FFT bins (Doppler bins). The receiver circuit may further be configured to combine the first and second bins to obtain (per receive channel) a combined range/Doppler map for the plurality of transmit channels. The combined range/Doppler map includes combined range/Doppler information relating to all sequences of FMCW radar chirps from the various broadcast channels. This combined range/Doppler information can be separated and assigned to each of the transmission channels taking into account the predefined DDM scheme.

In some embodiments, the receiver circuit is configured to identify each of the transmit channels based on evaluation of amplitudes and/or mutual distances of a plurality of second bins (Doppler bins) that are the same first bin (range bin) of the combined range/Doppler card are assigned. In this way, the different transmission channels can be identified based on distances between second FFT bins in which signal amplitudes exceed a certain threshold value. For example, the DDM phase pattern of at least one of the transmit channels may cause target appearances in two second FFT bins (associated with the same first FFT bin) that are separated from each other by a predefined bin distance corresponding to a predetermined phase difference. The predetermined phase difference is, for example, 90°, which corresponds to a bin distance of N _r /2, where N _r denotes a number of FMCW radar chirps within a sequence.

In some embodiments, the receiver circuit may be configured to identify a first of the plurality of transmit channels based on a pair of second bins with a mutual bin distance corresponding to a predetermined phase difference associated with the predefined Doppler division multiplexing scheme and identify at least a second one of the plurality of transmission channels based on amplitudes of the remaining second bins associated with the same first bin.

• - Senden einer ersten Sequenz von FMCW-Radar-Chirps über einen ersten Sendekanal, derart, dass die Start- und/oder Stoppfrequenzen jedes von zumindest einem Abschnitt der FMCW-Radar-Chirps der ersten Sequenz voneinander verschieden sind,
• - Senden einer zweiten Sequenz von FMCW-Radar-Chirps über einen zweiten Sendekanal, wobei die Start- und/oder Stoppfrequenzen jedes von zumindest einem Abschnitt der FMCW-Radar-Chirps der zweiten Sequenz voneinander verschieden sind,
• - wobei die erste und die zweite Sequenz von FMCW-Radar-Chirps gleichzeitig gesendet werden; und
• - Steuern des ersten und des zweiten Sendekanals, um Phasen der FMCW-Radar-Chirps der ersten Sequenz und der zweiten Sequenz gemäß einem vordefinierten Doppler-Divisions-Multiplex-Schema einzustellen.

According to another aspect, the present disclosure proposes a MIMO radar method. The procedure includes:

• - sending a first sequence of FMCW radar chirps over a first transmission channel, such that the start and / or stop frequencies of each of at least a portion of the FMCW radar chirps of the first sequence are different from one another,
• - sending a second sequence of FMCW radar chirps over a second transmission channel, the start and / or stop frequencies of each of at least a portion of the FMCW radar chirps of the second sequence being different from one another,
• - wherein the first and second sequences of FMCW radar chirps are transmitted simultaneously; and
• - Controlling the first and second transmit channels to adjust phases of the FMCW radar chirps of the first sequence and the second sequence according to a predefined Doppler division multiplexing scheme.

In some embodiments, the start and stop frequencies of respective subsequent FMCW radar chirps of the first and second sequences are increased or decreased by a predefined frequency offset.

Short description of the characters

• 1 zeigt ein Blockdiagramm einer herkömmlichen Radarsignal-Verarbeitungskette;
• 2 zeigt eine gemeinsame 2D-Entfernungs-/Doppler-Schätzung mit FMCW-Radar; Figurindex für die y-Achse ist Meile/Stunde, während die x-Achse in Metern ist
• 3 veranschaulicht eine Azimut-Winkelschätzung unter Verwendung eines gleichmäßigen linearen Antennen-Arrays;
• 4 veranschaulicht ein Konzept virtueller Array-Synthese ;
• 5A zeigt ein Zeitmultiplexzugriff- (TDMA-; Time Division Multiple Access) MIMO-Radarübertragungsverfahren;
• 5B zeigt ein Codemultiplexzugriff- (CDMA-; Code Division Multiple Access) MIMO-Radarübertragungsverfahren;
• 6 zeigt ein FMCW-Radarsignal, das eine herkömmliche Sequenz von FMCW-Radar-Chirps umfasst;
• 7A zeigt gemäß Ausführungsbeispiel der vorliegenden Offenbarung eine MIMO-Radarvorri chtung;
• 7B zeigt gemäß Ausführungsbeispiel der vorliegenden Offenbarung ein FMCW-Radarsignal, das eine gestufte Sequenz von FMCW-Radar-Chirps umfasst;
• 8A veranschaulicht gemäß Ausführungsbeispiel der vorliegenden Offenbarung ein FMCW-Radarsignal, das eine gestufte Sequenz von FMCW-Radar-Chirps umfasst;
• 8B veranschaulicht gemäß Ausführungsbeispiel der vorliegenden Offenbarung phasengesteuerte Modulationsvektoren für verschiedene Sendekanäle;
• 9 zeigt einen Empfängerprozess für ein gestuftes DDM-MIMO-Radarübertragungskonzept gemäß Ausführungsbeispielen;
• 10 veranschaulicht den Effekt der DDM-MIMO-Radarübertragung auf ein Dopplerspektrum; und
• 11 veranschaulicht die verbesserte Entfernungsauflösung, die mit der vorgeschlagenen abgestuften Sequenz von FMCW-Radar-Chirps erreichbar ist.

Some examples of devices and/or methods are described below solely by way of example and with reference to the accompanying figures, in which the following applies:

• 1 shows a block diagram of a conventional radar signal processing chain;
• 2 shows a joint 2D range/Doppler estimate with FMCW radar; Figure index for the y-axis is mile/hour while the x-axis is in meters
• 3 illustrates azimuth angle estimation using a uniform linear antenna array;
• 4 illustrates a concept of virtual array synthesis;
• 5A shows a Time Division Multiple Access (TDMA) MIMO radar transmission method;
• 5B shows a Code Division Multiple Access (CDMA) MIMO radar transmission method;
• 6 shows an FMCW radar signal comprising a conventional sequence of FMCW radar chirps;
• 7A shows a MIMO radar device according to an embodiment of the present disclosure;
• 7B shows, in accordance with embodiments of the present disclosure, an FMCW radar signal comprising a stepped sequence of FMCW radar chirps;
• 8A illustrates, in accordance with embodiments of the present disclosure, an FMCW radar signal that includes a stepped sequence of FMCW radar chirps;
• 8B illustrates phased modulation vectors for various transmit channels in accordance with embodiments of the present disclosure;
• 9 shows a receiver process for a staged DDM-MIMO radar transmission concept according to exemplary embodiments;
• 10 illustrates the effect of DDM-MIMO radar transmission on a Doppler spectrum; and
• 11 illustrates the improved range resolution achievable with the proposed stepped sequence of FMCW radar chirps.

Detailed description

Various examples will now be described in more detail with reference to the accompanying drawings, in which some examples are shown. In the figures, the thicknesses of lines, layers and/or regions may be exaggerated for clarity.

Accordingly, while further examples lend themselves to various modifications and alternative forms, some specific examples thereof are shown in the figures and will be described in detail below. However, this detailed description does not limit further examples to the specific forms described. Additional examples may cover all modifications, equivalents, and alternatives that fall within the scope of the disclosure. Same or similar reference number Throughout the description of the figures, references are made to the same or similar elements, which when compared to one another may be identical or implemented in modified form while providing the same or similar functionality.

It is understood that when an element is referred to as being “connected” or “coupled” to another element, the elements may be connected or coupled directly, or through one or more intermediate elements. When two elements A and B are combined using an “or”, it is understood that all possible combinations are disclosed, i.e. H. only A, only B and A and B, unless explicitly or implicitly stated otherwise. An alternative formulation for the same combinations is “at least one of A and B” or “A and/or B”. The same applies to combinations of more than two elements.

The terminology used herein for the purpose of describing specific examples is not intended to limit other examples. If a singular form, e.g. For example, if "a, an" and "the, the, the" are used and the use of only a single element is neither explicitly nor implicitly defined as mandatory, other examples may also use plural elements to implement the same function. Similarly, if functionality is described below as being implemented using multiple elements, further examples may implement the same functionality using a single element or processing entity. It is further understood that the terms "comprises", "comprising", "comprising" and/or "having" when used, specify the presence of the specified features, integers, steps, operations, processes, acts, elements and/or components, but does not exclude the presence or addition of one or more other features, integers, steps, operations, processes, acts, elements, components and/or any group thereof.

Unless otherwise defined, all terms (including technical and scientific terms) are used herein in their usual meaning of the field to which the examples belong.

1 shows a block diagram of a conventional radar signal processing chain 100.

A radio frequency (RF) transceiver front end 110 is used to generate transmit (TX) radar signals that may be emitted via one or more transmit antennas 112. The radar signals can be in frequency bands ranging from 3 MHz to 300 GHz. Automotive radar systems typically operate on bands in the 24 GHz and 77 GHz portions of the electromagnetic spectrum, known as mm-wave frequencies, so that adequate speed and range resolution can be achieved. One or more receive (Rx) antennas 114 are used to receive electromagnetic waves (radar signals) reflected from targets. Radar operation includes range (distance), relative speed and possibly direction estimation. The latter can be performed when more than one receiving antenna is used in a receiving antenna array. Radar systems that use both multiple transmit and multiple receive antennas are commonly referred to as MIMO radars. For proper transmit antenna spacing, the multiple-input multiple output (MIMO) radar can emulate a larger aperture phased array radar. This larger array can be called a virtual array.

A distance processor 120 connected downstream of the RF transceiver front end 110 is designed to carry out distance estimation. A distance R to a target can be determined based on the round-trip time delay that the electromagnetic waves use to propagate to and from that target: R = (cτ/2), where τ is the round-trip time delay in seconds and c is the speed of light in meters per second. Thus, estimating τ enables distance measurement. For example, pulse-modulated continuous waves (CWs) may include periodic and short power pulses and rest periods. The quiet periods allow the radar to receive the reflected signals and serve as time markers for the radar to perform the range estimation. A pulsed radar configuration using CW frequency modulated (FM) pulses can provide simultaneous range/speed estimation in multi-target traffic scenarios. An FMCW radar transmits periodic FM chirps (also referred to as pulses or ramps) that can increase linearly in frequency during the pulse. The received signal reflected from a target is conjugately mixed with the transmitted signal to produce a low-frequency beat signal (also referred to as a baseband signal) whose frequency indicates the distance of the target. This operation can be repeated for P consecutive FMCW chirps. Two-dimensional (2D) waves molds 210 in 2 represent successive reflected chirps arranged over two time indices, p, n. The so-called slow time index p simply corresponds to the chirp number. On the other hand, the so-called fast time index n assumes that for each chirp, the corresponding continuous beat signal is sampled at frequency f _s to collect N samples within a chirp duration T.

The range processor 120 may be configured to perform a first discrete Fourier transform (e.g., FFT) over the fast time n to obtain the beat frequency f _b coupled to the Doppler frequency f _d . This operation is also commonly known as range transform or range gating, which allows the estimation of a Doppler shift corresponding to a unique range gate or bin by applying a second Fourier transform (e.g. FFT) over slow time. This can be done by a speed processing element 130. Thus, a range/Doppler map 220 can be generated using a 2D FFT, see 2 . An exemplary range/Doppler map 220 shown in 2 , shows two targets, a first at 10 m distance and 0 mph relative speed, and a second at 20 m distance and 20 mph relative speed. The targets may be subregions of interest of the range/Doppler map.

Previously, it was assumed that automotive radars only received reflection from targets of interest, such as vehicles in front. However, in addition to direct reflections from a target of interest, the radar also receives reflections from road debris, guardrails and walls, for example. This unwanted return to the radar is commonly called clutter. The number of clutters in the system changes as the environment surrounding the vehicle varies. Thus, adaptive algorithms such as Constant False Alarm Rate (CFAR) processing and Space-Time Adaptive Processing (STAP) can be used to reduce the effect of clutter reduce. To identify valid targets in the presence of clutter, a target detection threshold should be appropriately selected. For example, if the amplitude of the range/Doppler map at an estimated range is greater than some threshold, the target may be considered detected. Thus, the threshold should depend on the noise (e.g., clutter) in the given system. As clutter increases, a higher threshold can be selected. A simple CFAR method based on cell or bin averaging can use a sliding window to derive the local clutter level by averaging multiple range bins. This described threshold selection and target (peak) detection is carried out in processing block 140.

The use of broadband pulses, such as FMCW pulses, provides discrimination of targets in both distance and speed. The discrimination in direction can be made using a multi-antenna array, such as in multi-antenna radar systems. Multi-antenna radar systems can employ multiple transmitters, multiple receivers, and multiple waveforms to exploit all available degrees of freedom. To spatially resolve targets and provide a comprehensive representation of the traffic scene, the angular location of targets is estimated. Thus, in automotive radars, the location of a target can be described in terms of a spherical coordinate system (R, θ, ρ), where (θ, ρ) denote azimuth and elevation angles, respectively. A single antenna radar setup is sufficient to provide a range-velocity map but is not sufficient to provide angular information because the measured time delay lacks information regarding the angular locations of the targets. To enable direction estimation, the radar is designed to receive reflected waves with multiple antennas. For example, positioning a target using electromagnetic waves in two dimensions requires that the reflected wave data from the object be collected in two individual dimensions. These individual dimensions can be formed in many ways using combinations of time, frequency and space via receiving antennas. For example, a linear receive antenna array 114 and wideband waveforms such as FMCW form two unique dimensions. Additionally, smaller wavelengths in mm-wave bands correspond to smaller aperture sizes and thus many antenna elements can be tightly packed into an antenna array. The effective radiation beam, which is stronger and sharper, thus in turn increases the resolution of angle measurements.

wo d die Distanz zwischen Antennenelementen (normalerweise die Hälfte der Wellenlänge), angeordnet in einer linearen Konstellation, ist. Die Verzögerungslaufzeit τl_q erzeugt gleichmäßige Phasenprogression über Antennenelemente, was die Schätzung des Winkels θ_q durch FFT in einem räumlichen Bereich erlaubt. Somit können die 2D-Position (Entfernung und Winkel) und die Geschwindigkeit von Zielen durch eine 3D-FFT geschätzt werden. Die dritte Winkel-FFT (Ankunftsrichtungs-, DOA- (Direction-of-Arrival), Verarbeitung) wirdConsider an antenna array, positioned in plane z = 0, and let l be the abscissa corresponding to each receiving antenna position, see 3 . Let (R _q , θ _q ) be the position of the qth target in spherical coordinates, moving with velocity v _q relative to the radar. Using far-field approximation for the qth target, the round-trip time delay between a transmitter positioned at the origin and the receiver positioned at a coordinate l can be expressed by $${\tau }_{lq}=\frac{2\left(R_q+v_{q}t\right)+ld\text{sin}{\theta }_q}{c},$$

where d is the distance between antenna elements (usually half the wavelength) arranged in a linear constellation. The delay time τl _q produces uniform phase progression across antenna elements, allowing the estimation of the angle θ _q by FFT in a spatial domain. Thus, the 2D position (distance and angle) and velocity of targets can be estimated by a 3D FFT. The third angle FFT (Direction-of-Arrival (DOA) processing) is

in processing block 150 of the exemplary radar signal processing block diagram of 1 executed.

Further conventional automotive radar processing may include target clustering 160, target tracking 170, and optional sensor fusion 180 with sensor data from other environmental sensor types (e.g., camera, lidar, etc.).

MIMO radar systems can employ multiple transmitters, multiple receivers, and multiple waveforms to exploit all available degrees of freedom. MIMO radars can be classified as “widely spaced” or “co-located.” With a widely spaced MIMO radar, the transmit/receive antennas capture different aspects of a target's radar cross section (RCS). In other words, the target appears to be spatially distributed, producing a different RCS at each antenna element. This RCS diversity can be used to improve radar performance. On the other hand, for a co-positioned MIMO radar, the RCS observed by each antenna element is indistinguishable.

Motor vehicles typically use co-positioned MIMO radars that are compact in size. For proper transmitter spacing, the co-positioned MIMO radar can emulate a larger aperture phased array radar, see 4 . This larger array is called a virtual array. For MIMO radar processing, as in 4 shown, a 1-D receiver (-Rx) array with two transmit (Tx) antennas is considered. N _T and N _R denote a number of Tx and Rx antenna elements, respectively. It is assumed that d _T and d _R represent corresponding Tx and Rx antenna spacings. Furthermore, it is assumed that Tx and Rx antenna positions are given by l _T and l _R in Cartesian coordinates. The 2-D FMCW mixer output signal over a fast time and aperture can therefore be referred to as $$d\left(l_T,l_R,n\right)\approx {\displaystyle \sum _{q=0}^{Q-1}{\alpha }_p\text{exp}}\left\{\mathrm{<figure-callout\; id="2"\; label="j"\; filenames="DE102022106791A1\_0006.png,DE102022106791A1\_0009.png"\; state="\{\{state\}\}">j</figure-callout>}2\pi \left[\frac{2K{R}_q}{c}\frac{n}{f_s}+\frac{f_c\left\{\left(l_T{d}_T+l_R{d}_R\right)sin{\theta }_q\right\}}{c}+\frac{2f_c{R}_q}{c}\right]\right\}+\omega \left(lT,lR,n\right)$$

From the above equation, it can be seen that when d _T =N _R × d _R , the MIMO radar imitates a regular 1-D array radar with a single Tx and _NT × N _R Rx antenna elements. This is known as a virtual array representation. Therefore, the spatial resolution of the FFT-based target mapping can be improved by a factor of N _T.

A challenging aspect of MIMO radar is the selection of waveforms. The waveforms can e.g. B. can be made orthogonal in the frequency, time or code domain.

For a MIMO radar, the most intuitive and simple way to easily separate the signals sent by different TX antennas (Tx channels) is alternate sending, that is, each Tx channel sends its own waveform alternately, and there is no overlap between any two broadcasts. This is in 5A shown. With this alternative time division multiplexing (TDM) transmission approach, ideal orthogonality can be achieved and the traditional radar waveform (e.g. chirp waveform) can be used directly in all transmitters. Although this alternative TDM transmission approach is easy to use, it is obvious that the transmission capabilities of all TX antennas are not fully utilized. Compared to the MIMO radar, where all TX antennas can transmit simultaneously, this alternative transmission approach suffers from a loss in transmit power, resulting in a shorter target detection distance (processing gain remains the same or detection distances are the same, at the expense of a Doppler uniqueness range reduction). TDM).

Doppler division multiplexing (DDM) (also called code division multiplexing (CDM)) MIMO waveform means that the signals sent from different TX channels are modulated by different series of phase codes, either in fast time or in slow time so that these signals can be separated/decoded in a radar receiver. Since there is no ideal orthogonal code sequence with good auto- and cross-correlation properties, the DDM-MIMO waveforms can only approximately satisfy the orthogonality requirement. In the Fast Time DDM (CDM) waveform, the phase codes in each pulse/chirp are modulated by the carrier signal. The slow time DDM waveform uses the phase codes to modulate the initial phases of the various chirps. 5B illustrates an example of a slow-time DDM (CDM) waveform in which chirps from transmit channel Tx1 and transmit channel Tx2 are transmitted simultaneously. However, Tx2 applies a different phase code than Tx1. In the example shown, the initial phase of every other chirp of Tx2 is 180°, while Tx1 applies an initial phase of 0° for every chirp.

6A illustrates an FMCW radar signal that includes a conventional sequence 600 of FMCW radar chirps 602.

The chirp sequence 600 includes a frame of _NR FMCW radar chirps 602-1, 602-2, ..., 602-NR. The chirp sequence 600 thus includes N _R FMCW radar chirps in the slow time domain. An example would be N _R = 512. Each chirp 602 has a duration of t _dwell and includes a number of samples N _samples (fast time). Each chirp 602 therefore includes N _samples in the fast time range. An example would be N _samples = 1024. At a sampling frequency f _s , the number of samples per ramp is N _samples = t _dwell * f _s . An example would be f _s = 25 MSPS. A chirp repetition interval is denoted by t _R > t _dWell , which is the time between the respective start times of two consecutive chirps 602 within the sequence 600. The duration of the entire chirp sequence 600 is therefore t _R * N _R . Each chirp 602 of the sequence 600 has the same start frequency f _start and the same stop frequency f _stop . The difference between the stop frequency f _stop and the start frequency f _start denotes a waveform modulation bandwidth BW. An example would be BW = 220.3 MHz.

As shown in the table in 6B As can be seen, a range resolution R _res of the conventional FMCW radar signal depends on the modulation bandwidth BW according to R _res = c/2/BW, where c denotes the speed of light. This means that the range resolution R _res is improved when BW is wide. The maximum detection distance R _max of the conventional FMCW radar signal depends on the bandwidth BW according to R _max = c/2/BW*N _samples /2. This means that the maximum detection distance R _max is shorter when BW is wide. On the other hand, a narrow BW results in a poor range resolution R _res but a longer maximum detection distance R _max . R _max and R _res are therefore in a compensatory relationship.

The present disclosure proposes to increase an effective modulation bandwidth BW to achieve a higher range resolution R _res for MIMO radar systems. This can be combined with a predefined DDM scheme to optimize the time domain and frequency domain characteristics with limited frequency resource.

7A illustrates a block diagram of a MIMO radar device 700 according to an embodiment of the present disclosure.

The MIMO radar device 700 includes a transmitter circuit 710. The transmitter circuit 710 includes a plurality of Tx channels 712-1, 712-2, 712-3. Each Tx channel may include digital and/or analog hardware components such as: E.g. power splitters, phase shifters, power amplifiers and TX antennas. Although 7A illustrating an example of three TX channels, one skilled in the art, taking advantage of the present disclosure, will recognize that the increased effective modulation bandwidth BW concept described herein may also be applicable to fewer or more Tx channels.

The transmitter circuit 710 is designed to transmit a first sequence of FMCW radar chirps via a first Tx channel 712-1. In contrast to that in 6 In the conventional case described, the start and/or stop frequencies of each of at least a portion of the FMCW radar chirps of the first chirp sequence are different from each other. Furthermore, the transmitter circuit 710 is designed to transmit a second sequence of FMCW radar chirps via a second Tx channel 712-2. Here too, the start and/or stop frequencies of each of at least a portion of the FMCW radar chirps of the second chirp sequence are different from one another. Note that the first Tx channel 712-1 and the second Tx channel 712-2 operate simultaneously. The first and second sequences of FMCW radar chirps are therefore sent at the same time. In other words, the FMCW radar chirps of the first and second chirp sequences are transmitted simultaneously.

Since the first and second sequences of FMCW radar chirps are transmitted simultaneously, it is necessary that signals from the first Tx channel 712-1 and the second Tx channel 712-2 and the optional third Tx channel 712- 3 can be separated at a receiving section 730 of the MIMO radar device 700. As mentioned earlier, this can be achieved through Doppler Division Multiplexing (DDM). Doppler Division Multiple Access (DDMA) is an interchirp phase coding scheme (“slow time” coding). In this configuration, the signal spectrum of each Tx channel is slightly shifted, allowing waveforms to be separated in the Doppler range.

For this purpose, the MIMO radar device 700 further includes a control circuit 720 configured to control the first and second Tx channels 712-1, 712-2 (and the optional third Tx channel 712-3) to phase adjust the FMCW radar chirps of the first sequence and the second sequence (and an optional third chirp sequence) according to a predefined DDM scheme. As in 7A As shown, the control circuit 720 may include respective analog or digital phase shifters 722-1, 722-2 and 722-3 for the respective Tx channels 712-1, 712-2 and 712-3.

The MIMO radar device 700 may optionally additionally include a receiver circuit 730 for receiving reflections of the transmitted FMCW radar signals. The receiver circuit 730 includes at least one Rx channel 732. The at least one Rx channel 732 is designed to receive a received signal corresponding to the reflections of the first sequence of FMCW radar chirps, the second sequence of FMCW radar chirps and a optional third sequence of FMCW radar chirps of the optional third Tx channel 712-3. In the example shown, the receiver circuit 730 includes not only one Rx channel 732, but four Rx channels 732-1, ..., 732-4 for improved signal-to-noise ratio (SNR). ratio) and improved angular resolution. The receiver circuit 730 is explained in more detail below. Those skilled in the art who take advantage of the present disclosure will recognize that the transmitter circuit 710 and the receiver circuit 730 may be integrated or implemented separately and may include, but not include, digital and analog circuit components used in FMCW radar transceivers limited to, e.g. E.g. baseband circuits, mixer stages, RF circuits, digital-to-analog converters (DACs; digital-to-analog converters), analog-to-digital converters (ADCs; analog-to-digital converters), amplifiers, antennas and Similar.

A sequence 750 of N _R FMCW radar chirps 752 according to an embodiment of the present disclosure is shown in FIG 7B shown. The FMCW sequence shown can be referred to as a stepped FMCW sequence.

In the example shown by 7B Not only do the start and/or stop frequencies of some of the FMCW radar chirps 752 of the sequence 750 differ from one another. The start and stop frequencies of each of the successive N _R FMCW radar chirps 752 of the sequence 750 are different from each other. In the example shown, the control circuit 720 is designed to control the first Tx channel 712-1 such that the starting frequencies of each subsequent pair of subsequent FMCW radar chirps 752 of the first sequence 750 differ by a predefined frequency offset Δf. Likewise, the control circuit 720 is designed to control the second Tx channel 712-2 such that the starting frequencies of each subsequent pair of subsequent FMCW radar chirps 752 of the second sequence 750 differ by the same predefined frequency offset Δf. In the case of a third Tx channel 712-3, the control circuit 720 is designed to control the third Tx channel 712-3 such that the starting frequencies of each subsequent pair of FMCW radar chirps 752 of the third sequence 750 are by the same predefined frequency offset Δf differ. In the example shown, the control circuit 720 is designed to control the respective Tx channels 712-1, ..., 712-3 such that the starting frequency of each subsequent FMCW radar chirp 752-n (n = 2, .. ., N _R ) of the respective chirp sequence 750 increases by the predefined frequency offset Δf compared to its previous FMCW radar chirp 752-(n-1). Thus, the starting frequency f _start,n of the nth (n = 2, ..., N _R ) FMCW radar chirp 752-n of the respective chirp sequence is f _start,n = f _start,1 + (n- 1) · Δ _f , where f _start,1 denotes the start frequency of the initial chirp 752-1 of the respective chirp sequence. Likewise, the stop frequency f _stop,n of the nth (n = 2, ..., N _R ) FMCW radar chirp 752-n of the respective chirp sequence is f _stop,n = f _stop,i + (n- 1) · Δf, where f _start,1 denotes the stop frequency of the initial chirp 752-1 of the respective chirp sequence.

In the example shown, the stepped FMCW chirp sequence 750 from 7B It is assumed that the duration t _dwell and the bandwidth BW for each chirp 752-n (n = 1, ..., N _R ) of the sequence 750 are identical table are. Thus, with an increase in starting frequencies between each successive pair of subsequent FMCW radar chirps 752 by the predefined frequency offset Δf, an increase in an effective total bandwidth BW of the sequence 750 compared to the conventional case of 6 be received. The increase in the total effective bandwidth BW can be (Nr-1)*f _R -1)*Δf compared to the conventional case of 6 be. The stepped FMCW chirp sequences of all Tx channels 712-1, 712-2, 712-3 can be identical and sent simultaneously. This means that all N _R chirps of the different stepped FMCW radar chirp sequences of the different Tx channels can be transmitted simultaneously and with the same increasing respective start and stop frequencies.

In the example shown, the starting frequency increments by Δf per each chirp 752 within the chirp sequence 750. The number of samples N _samples and other conditions may be the same as the conventional FMCW waveform 6 . The larger total effective bandwidth BW of the resulting stepped FMCW waveform can result in increased range resolution. However, R _max can remain the same if the actual BW per chirp 752 does not change.

Those skilled in the art, taking advantage of the present disclosure, will recognize that the starting frequency may also decrement by Δf for each chirp 752 within a chirp sequence. In other examples, the starting frequency could only increment or decrement every other chirp within a chirp sequence. Numerous variations are conceivable in order to achieve a larger effective total bandwidth BW of the chirp sequence compared to the conventional case. Embodiments do not rely on the chirp sequence 750 of the example 7B limited.

According to the present disclosure, the stepped FMCW waveforms or sequences of the plurality of Tx channels 712-1, ..., 712-3 may be combined with a predefined DDM scheme for Tx channel separation at the receiving side. For this purpose, the control circuit 720 (comprising respective phase shifters 722) may be configured to assign each Tx channel 712-1, 712-2 and 712-3 a unique sequence of phases that correspond to the respective sequence of FMCW chirps of the respective Tx channel is applied. The phases for the respective unique sequence of phases for each Tx channel can generally be selected from an M-ary phase modulation alphabet, where M ≥ 2 is an integer. M = 2 would mean a binary phase modulation alphabet, e.g. B. with phases 0° and 180°, or 45° and 225°, or 90° and 270°, etc. M = 4 would mean a quaternary phase modulation alphabet, e.g. B. with the phases 0°, 90°, 180° and 270°. The higher M, the more Tx channels can be separated. A sequence of phases applied to a chirp sequence can also be called a phase modulation vector.

In one embodiment, three Tx channels 712-1, 712-2 and 712-3 may be separated using a DDM scheme with a binary phase modulation alphabet (ie, M = 2). An example of the DDM scheme for the respective chirp sequences of the three Tx channels 712-1, 712-2 and 712-3 is shown in 8th shown.

In 8A The frequency stepped FMCW chirp sequence 750 is repeated, which is transmitted simultaneously from all Tx channels 712-1, 712-2 and 712-3. The respective sequences of FMCW radar chirps each comprise N _R chirps, where a chirp start frequency is increased by Δf chirp for chirp, while individual chirp bandwidths BW remain unchanged (constant) during the respective sequences. The achievable distance resolution thus improves $${\text{R}}_{\text{res}}=c/(2*\text{BW}*\Delta \text{f}*\left({\text{N}}_{\text{R}}-1\right)).$$

8B illustrates an example configuration of the respective phase shifters 722-1, 722-2 and 722-3 for the Tx channels 712-1, 712-2 and 712-3. The control circuit 720 is designed to assign a (single) first phase φ ₁ to the FMCW chirps of the first chirp sequence of the first Tx channel 712-1 (Tx1). In the example shown, the first phase is φ ₁ 0°. All N _R stepped FMCW chirps 752-n (n = 1, ..., N _R ) of the first chirp sequence are sent with the first phase φ ₁ . The 1 × N _R phase modulation vector for the first Tx channel 712-1 (Txl) can thus be [φ ₁ , φ ₁ , φ ₁ , φ ₁ , ..., φ ₁ , φ ₁ ].

The control circuit 720 is further designed to assign the first phase φ ₁ and a second phase φ ₂ to the FMCW chirps of the second sequence of the second Tx channel 712-2 (Tx2). At the in 8B The example shown is the second phase φ ₂ 180°. A temporal order of the first phase φ ₁ and the second phase φ ₂ is specific to the second Tx channel 712-2. For example, every odd FMCW chirp of the second chirp sequence is sent with the first phase φ ₁ . Any even numbered FMCW chirp of the second Chirp sequence is sent with the second phase φ ₂ . This can also be the case the other way around. Therefore, the simultaneously transmitted FMCW chirps of the first and second sequences have predetermined phase differences, which include a first predetermined phase difference Δφ ₁ and a second predetermined phase difference Δφ ₂ . In the example shown, the first predetermined phase difference Δφ ₁ is 0° and the second predetermined phase difference Δφ ₂ is 180°. However, other first and second predetermined phase differences Δφ ₁ , Δφ ₂ are also possible in other exemplary embodiments. The 1 × N _R phase modulation vector for the second Tx channel 712-2 (Tx2) can thus be [φ ₁ , φ ₂ , φ ₁ , φ ₂ , ..., φ ₁ , φ ₂ ].

The control circuit 720 is further designed to assign the first phase φ ₁ and the second phase φ ₂ to the FMCW chirps of the third sequence of the third Tx channel 712-3 (Tx3). Here too, a temporal order of the first phase φ ₁ and the second phase φ ₂ is specific to the third Tx channel 712-3 and differs from the temporal order of the first phase and the second phase for the second Tx channel 712-2 . For example, the first and second FMCW chirps of the third chirp sequence are transmitted with the first phase φ ₁ . The third and fourth FMCW chirps of the third chirp sequence are transmitted with the second phase φ ₂ . The fifth and sixth FMCW chirps of the third chirp sequence are transmitted with the first phase φ ₁ . The seventh and eighth FMCW chirps of the third chirp sequence are transmitted with the second phase φ ₂ etc. The 1 × N _R phase modulation vector for the third Tx channel 712-3 (Tx3) can thus be [φ ₁ , φ ₁ , φ ₂ , φ ₂ , φ ₁ , φ ₁ , φ ₂ , φ ₂ , ..., φ ₁ , φ ₁ , φ ₂ , φ ₂ ].

In the example shown by 8B the control circuit 720 is formed while transmitting two consecutive FMCW chirps of the first sequence with the first phase φ ₁ (0°), the first phase φ ₁ (0°) a first FMCW chirp of the second sequence and the second phase φ ₂ (180°) to a subsequent second FMCW chirp of the second sequence. Furthermore, the control circuit 720 is designed to assign the first phase _{φ 1 (} 0°) to two consecutive FMCW radar chirps of the third sequence, followed by the second phase φ ₂ (180°), which is assigned to two more consecutive FMCW radar chirps of the third sequence.

Those skilled in the art who will benefit from the present disclosure will recognize that it is not necessary for the first phase φ ₁ 0°. Instead it could be any phase value φ. Then the second phase φ ₂ could z. B. can be selected to φ+180°. It is not necessary that a phase difference between the first and second phases is necessarily 180°. Other values are also possible.

As already mentioned, the in 7 MIMO radar device 700 shown also includes a (monostatic or bistatic) receiver circuit 730 for receiving reflections of the transmitted DDM-MIMO waveforms. The receiver circuit 730 includes at least one Rx channel 732-1 (including an Rx antenna and analog and digital Rx circuits). Typically, the receiver circuit 730 includes a plurality of Rx channels 732-1, 732-2, ..., 732-NRx. In the example shown, N _Rx = 4, resulting in a virtual array of 12 antenna elements. For high-resolution radar devices, e.g. B. a combination of N _Tx = 3 Tx channels and N _Rx = 16 Rx channels is conceivable, which leads to a virtual array of 48 antenna elements.

The receiver circuit 730 is designed to receive a respective Rx signal via the Rx channel 732-n (n = 1, ..., N _Rx ), which is a superposition of reflections of the first sequence of FMCW radar chirps (of Tx1). , the second sequence of FMCW radar chirps (from Tx2) and the third sequence of FMCW radar chirps (from Tx3). The following reception procedure is described for an Rx channel 732. However, this can be applied to a plurality of Rx channels for virtual array synthesis.

As in the example Rx process 900 from 9 shown, after analog-to-digital conversion (ADC) (see 902) of the received radar signal, a respective range FFT (over fast time) may be performed (see 904). That is, in step 904, the receiver circuit 730 may be configured to perform (for each Rx channel) a range FFT (over fast time) of the respective received signal to generate a plurality of range bins corresponding to the respective Rx channel and assigned to the plurality of Tx channels. Ranging FFTs are well known to those skilled in the art who will benefit from the present disclosure.

After the range bins are generated by the range FFT, a Doppler FFT (over slow time) can be performed using the previously obtained range bins to to generate a plurality of Doppler bins that include velocity information (see step 906). Doppler FFTs are well known to those skilled in the art who will benefit from the present disclosure. After performing Doppler FFT, combined 2D range/Doppler map data is available for the majority of Tx channels (Tx1 - Tx3). A respective combined 2D range/Doppler map can be obtained for each Rx channel.

When more than one Rx channel is used, the detection probability can be improved by summing or integrating the 2D range/Doppler map data from all Rx channels. For example, noncoherent integration (NCI) of the available range/Doppler maps may be performed (see step 908). To obtain a 2D range/Doppler reference map, adaptive threshold and peak detection algorithms such as Constant False Alarm Rate (CFAR) processing or Space-Time Adaptive (STAP) processing can be used to reduce the effect of to mitigate clutter (see 910, 912). The remaining peaks (above a certain signal threshold) in the 2D range/Doppler reference map may identify targets and indicate a respective distance (R) of the target from the MIMO radar device 700.

It is noted that the Doppler bins associated with a range bin of the 2D range/Doppler reference map may still be ambiguous due to the predefined DDM scheme. However, distance information can be unique. To obtain unique Doppler (i.e., velocity) information from the 2D range/Doppler reference map, the receiver circuit 730 may be configured to evaluate each of the Tx channels (Tx1, Tx2, Tx3) based on evaluation of amplitudes and/or identify mutual distances of a plurality of Doppler bins (peaks) associated with the same range bin of the 2D range/Doppler reference map for each Rx channel. This can be done in step 914, which can be divided into steps 914-1, 914-2 and 914-3.

10 shows the phase modulation effects corresponding to the different phase modulation vectors used for the different Tx channels 712-1, 712-2, 712-3 (Tx1, Tx2, Tx3).

10 shows a phase shift effect in Doppler FFT associated with a phase distance between elements of the phase modulation vectors. As a result of the (initial) phase modulation of an FMCW chirp sequence in a Tx channel, original frequency indices (bins) of the Doppler FFT of scatters corresponding to an in 10 shown phase shift effect shifted. For example, if three TX channels are used with different phase modulation vectors, shown in 8B , four different frequency indices (bins) appear after a Doppler FFT from the same range FFT bin. A bin distance between the Doppler bins depends on the modulation vectors of the Tx channels.

In the example shown, the (slow) frequency index 0 corresponds to the phase modulation vector of the Tx channel Tx1 = [0°, 0°, 0°, 0°, 0°, 0°, 0°, 0°, 0°]. The phase distance between adjacent elements of the modulation vector is 0°. The (slow) frequency index -N _R /2 corresponds to the phase modulation vector of the Tx channel Tx2 = [0°, 180°, 0°, 180°, 0°, 180°, 0°, 180°]. The phase distance between adjacent elements of the modulation vector is 180°. The (slow) frequency indices ±N _R /4 correspond to the phase modulation vector of the Tx channel Tx3 = [0°, 0°, 180°, 180°, 0°, 0°, 180°, 180°]. The (average) phase distance between neighboring elements of the modulation vector is 90°.

Out of 10 It can be seen that peaks in the Doppler bins of Tx1, Tx2 and Tx 3 (which are associated with the same range bin) whose values are above a threshold level, called the CFAR threshold, are within a (slow) frequency index window of - N _R /2 to N _R /4 occur. If the range bin index is denoted F _ind and the Doppler bin index of the leftmost peak is denoted S _ind , the bin positions of the peaks can be in an Rx channel-specific range/Doppler Map can be described as ( _Find , _Sind ) SNAP _{RX(i)_P1} (F _ind , mod(S _ind +1 *N _R /4)), SNAP _{Rx(i)_P2} (F _ind , mod(S _ind +1 *2N _R /4)), SNAP _{RX(i)_P3} (F _ind , mod(S _ind +1*3N _R /4)) SNAP _{Rx(i)_P4} . RX(i) denotes the i-th Rx channel.

The Doppler bin window size in which peak values corresponding to Tx1, Tx2 and Tx3 can be expected is therefore 3N _r /4 in the example shown. Each pair of peaks in the Doppler bin window has a bin distance of N _R /4. Thus, the DDM synthesis 914 may include a step 914-1 of identifying the Doppler bin window by pairing peaks in four Doppler bins associated with the same range bin with a Doppler bin distance of N _R /4 between adjacent peaks. Step 914-1 provides peak Doppler signals corresponding to Tx channels Tx1, Tx2 and Tx3.

Then, the DDM synthesis 914 may include a step 914-2 of identifying the Tx channel Tx3 based on the evaluation of the peak amplitudes and the mutual distances of the four previously identified Doppler bins associated with the same range bin and located in the Doppler -Bin window from -N _r /2 to N _r /4. How out 10 As can be seen, the third Tx channel Tx3 with the phase modulation vector [0°, 0°, 180°,180°,0°, 0°, 180°,180°] causes two peaks in two Doppler bins ±N _r /4 , the N _R /2 Doppler bins are spaced apart. A peak amplitude for the Doppler bins ±N _r /4 (Tx3) is less (1/sqrt(2)) than those of the Doppler bins 0 (Tx1) and -N _r /2 (Tx2). Thus, among the four Doppler bins assigned to the same range bin, Doppler bins N _r /2 bins are spaced apart and have a phase difference of 90°, these Doppler bins can be considered as belonging to the third Tx channel Tx3 Phase modulation vector [0°, 0°, 180°,180°,0°, 0°, 180°,180°] can be identified. Optionally, the lower amplitude of the Doppler bins can also be used to identify the third Tx channel Tx3. After the Tx channel Tx3 is identified, the DDM synthesis 914 may include identifying the Tx channels Tx1 and Tx2 in step 914-2 based on relative positions of the remaining Doppler bins (peaks) and/or their amplitudes. For example, a peak centered between the Doppler bins associated with the TX channel Tx3 may be associated with the (unmodulated) TX channel Tx1. The remaining fourth peak at -N _R /2 then belongs to Tx channel Tx2. Therefore, the receiver circuit 730 may be configured to select a first (e.g., Tx3) of the plurality of Tx channels based on a pair of Doppler bins with a predetermined bin distance (e.g., N _R /2) and a predetermined Identify phase difference (e.g. 90°) associated with the predefined DDM scheme. Their amplitudes may be lower than Doppler bins corresponding to other Tx channels (e.g. Tx1, Tx2). At least a second of the plurality of Tx channels may be identified based on positions of the remaining Doppler bins associated with the same range bin relative to the bin positions of the first identified pair of Doppler bins. The remaining Doppler bins can have higher amplitudes than the remaining Doppler bins. Those skilled in the art, taking advantage of the present disclosure, will recognize that the Doppler bin window and the positions of the bins within the window may vary depending on the phase modulation vectors employed.

Under a (different) assumption that the peaks SNAP _{RX(i) P1} and SNAP _{RX(i) P3} correspond to the Tx channel Tx3, SNAP _{RX(i) P2} corresponds to the Tx channel Tx1 and SNAP _{RX(i) P4} to that Tx channel corresponds to Tx2, the Tx channel identification for all Rx channels RX(1) to RX(4) can be implemented according to the following pseudocode:

After all Tx channels Tx1, Tx2, Tx3 have been identified, the DDM synthesis 914 may include a step 914-3 to determine distance and velocity for all Tx channels Tx1, Tx2, Tx3 from the tip(s). , which are assigned to the non-modulated Tx channel (here: Tx1).

In step 916, virtual array synthesis may be performed by combining the results of the previous steps for all Rx channels. In this way, range/Doppler map data can be generated for each Tx-Rx channel pair.

With the results obtained so far, the angle calculation 918 can be performed by DoA (direction of arrival) processing over the virtual array including each Tx-Rx channel pair. DoA processing can be done by performing a 3rd FFT (angular FFT) across all antennas of a virtual array. Here phase information of the detected peaks can be used in the range/Doppler maps.

As in 11 (right), embodiments can achieve improved range resolution, enabling identification of different targets where traditional solutions could not distinguish targets. The range resolution of the embodiments may be R _res = c / (2*BW*Δf* ^k (N _R -1)), while the conventional range resolution is R _res = c / (2*BW).

The aspects and features mentioned and described together with one or more of the previously detailed examples and figures may also be combined with one or more of the other examples to replace a like feature of the other example or to incorporate the feature into the other Introduce an additional example.

Examples may further include or relate to a computer program that includes program code for executing one or more of the foregoing methods when the computer program is executed on a computer or processor. Steps, operations, or processes of various methods described above may be performed by programmed computers or processors. Examples may also cover program storage devices, such as digital data storage media, that are machine, processor, or computer readable and encode machine-executable, processor-executable, or computer-executable programs of instructions. The instructions perform or cause to be performed some or all of the steps in the procedures described above. The program storage devices can e.g. B. include or be digital storage, magnetic storage media such as magnetic disks and magnetic tapes, hard disk drives or optically readable digital data storage media. Further examples may also include computers, processors or control units that are programmed to carry out the steps of the methods described above, or (field) programmable logic arrays ((F)PLAs; (field) programmable logic arrays). Cover gate arrays ((F)PGA; (Field) Programmable Gate Arrays) programmed to carry out the steps of the methods described above.

The description and drawings represent only the principles of the disclosure. Furthermore, all examples listed here are expressly intended to serve illustrative purposes only to assist the reader in understanding the principles of the disclosure and the concepts contributed by the inventor(s) to the further development of the technology. All statements herein regarding principles, aspects, and examples of the revelation, as well as specific examples thereof, are intended to include their equivalents.

A functional block that performs a specific function, referred to as a "means for...", may refer to a circuit that is designed to perform a specific function. Thus, a “means for something” can be implemented as a “means designed for or suitable for something”, e.g. B. a device or a circuit that is designed for or suitable for the respective task.

Functions of various elements shown in the figures, including any functional blocks referred to as "means", "means for providing a signal", "means for generating a signal", etc., may be in the form of dedicated hardware, e.g. B. "a signal provider", "a signal processing unit", "a processor", "a controller" etc. as well as implemented as hardware capable of executing software in conjunction with associated software. When provided by a processor, the functions may be provided by a single dedicated processor, by a single shared processor, or by a plurality of individual processors, some or all of which may be shared. However, the term "processor" or "controller" is far from limited to hardware capable of executing software only, but can include digital signal processor (DSP) hardware, network processor, application specific integrated circuit (ASIC). ), field programmable gate array (FPGA), read only memory (ROM) for storing software, random access memory (RAM), and nonvolatile storage. Other hardware, conventional and/or custom, may also be included.

For example, a block diagram may represent a detailed circuit diagram that implements the principles of the disclosure. Similarly, a flowchart, a sequence diagram, a state transition diagram, pseudocode, and the like may represent various processes, operations, or steps, for example, represented substantially in a computer-readable medium and so executed by a computer or processor, regardless of whether such Computer or processor is explicitly shown. Methods disclosed in the specification or claims may be implemented by an apparatus having means for carrying out each of the respective steps of these methods.

It is to be understood that the disclosure of several steps, processes, operations or functions disclosed in the specification or claims should not be construed as being in the particular order unless explicitly or implicitly stated otherwise, e.g. B. for technical reasons. Therefore, the disclosure of multiple steps or functions does not limit them to a particular order, unless these steps or functions are not interchangeable for technical reasons. Further, in some examples, a single act, function, process, operation, or step may each include multiple sub-acts, functions, processes, operations, or -comprise steps and/or be broken down into the same. Such sub-steps may be included and be part of the disclosure of this individual step, provided they are not explicitly excluded.

Furthermore, the following claims are hereby incorporated into the Detailed Description, where each claim may stand on its own as a separate example. Although each claim can stand on its own as a separate example, it should be noted that - although a dependent claim may refer to a particular combination with one or more other claims in the claims - other examples also include a combination of the dependent claim with the subject matter of each may include other dependent or independent claims. Such combinations are explicitly suggested here, unless it is stated that a particular combination is not intended. Furthermore, features of a claim should also be included for any other independent claim, even if that claim is not made directly dependent on the independent claim.

Claims (21)

A MIMO radar device (700) comprising: a transmitter circuit (710) comprising a plurality of transmission channels (712); wherein the transmitter circuit (710) is formed to send a first sequence of FMCW radar chirps via a first transmission channel (712-1) such that the start and / or stop frequencies of each of at least a portion of the FMCW radar chirps of the first sequence are different from one another, to transmit a second sequence of FMCW radar chirps via a second transmission channel (712-2; 712-3), wherein the start and / or stop frequencies of each of at least a portion of the FMCW radar chirps of the second sequence are different from one another , wherein the first and second sequences of FMCW radar chirps are transmitted simultaneously; and a control circuit (720) configured to control the first and second transmission channels (712-1; 712-2) to phase the FMCW radar chirps of the first sequence and the second sequence according to a predefined Doppler division Multiplex, DDM, scheme to set. The MIMO radar device (700) according to Claim 1 , wherein the control circuit (720) is designed to control the first and second transmission channels (712-1; 712-2) such that the respective FMCW radar chirps of the first and second sequences include at least one of: the same respective duration, or an equal respective bandwidth between the respective start and stop frequencies. The MIMO radar device (700) according to one of the preceding claims, wherein the control circuit (720) is adapted to control the first and second transmission channels such that the starting frequencies of each subsequent pair of subsequent FMCW radar chirps of the first sequence differ a predefined frequency offset and the start frequencies of each subsequent pair of subsequent FMCW radar chirps of the second sequence differ by the predefined frequency offset. The MIMO radar device (700) according to Claim 3 , wherein the control circuit (720) is configured to control the first and second transmission channels such that the starting frequencies of each subsequent pair of subsequent FMCW radar chirps of the first and second sequences increase or decrease by the predefined frequency offset. The MIMO radar device (700) according to one of the preceding claims, wherein the control circuit (720) is adapted to control the first and second transmission channels such that the respective simultaneous FMCW radar chirps of the first and second sequences have the same start - and stop frequency. The MIMO radar device (700) according to any one of the preceding claims, wherein the control circuit (720) is adapted to assign each transmit channel a unique sequence of phases which is applied to the respective sequence of FMCW chirps of the respective transmit channel. The MIMO radar device (700) according to one of the preceding claims, wherein the control circuit (720) is configured to select the phases for the predefined Doppler division multiplexing scheme from an M-ary phase modulation alphabet, where M ≥ 2 is an integer. The MIMO radar device (700) according to one of the preceding claims, wherein the control circuit (720) is configured to assign a first phase to the FMCW chirps of the first sequence, to assign the first and a second phase to the FMCW chirps of the second sequence, wherein Simultaneously transmitted FMCW chirps of the first and second sequences have predetermined phase differences, the predetermined phase differences comprising a first and a second predetermined phase difference. The MIMO radar device (700) according to Claim 8 , wherein the first predetermined phase difference is 0° and the second predetermined phase difference is 180°. The MIMO radar device (700) according to Claim 8 or 9 , wherein the control circuit (720) is formed while two successive FMCW chirps of the first sequence are transmitted with the first phase, the first phase being a first FMCW chirp of the second sequence and the second phase being a subsequent second FMCW chirp of the second sequence to assign. The MIMO radar device (700) according to one of the preceding claims, wherein the control circuit (720) is designed, assign a sequence of first phases to the first sequence of FMCW chirps, the first phases comprising either a first phase value or a second phase value, the first phase value and the second phase value differing by 180°, and assign a sequence of second phases to the second sequence of FMCW chirps, the second phases comprising either a third phase value or a fourth phase value, the third phase value and the fourth phase value differing by 180°. The MIMO radar device (700) according to Claim 11 , wherein at least one of the first phase value and the second phase value is different from each of the third and fourth phase values or at least one of the first and second phase values is equal to the third phase value. The MIMO radar device (700) according to one of the preceding claims, wherein the transmitter circuit (710) comprises a third transmission channel (712-3) and is designed to transmit a third sequence of FMCW radar chirps via the third transmission channel such that the start and/or stop frequencies of each of at least a portion of the FMCW radar chirps of the third sequence are different from each other, and wherein the control circuit (720) is designed, assign a first phase to the first transmit channel (712-1), which is applied to each FMCW radar chirp of the first sequence, to assign to the second transmission channel (712-2) a sequence of second phases that is applied to the FMCW radar chirps of the second sequence, the sequence of second phases comprising the first phase and a second phase in a time specific to the second transmission channel Order includes, and to assign to the third transmission channel (712-3) a third phase sequence that is applied to the FMCW radar chirps of the third sequence, the third phase sequence comprising the first phase and the second phase in a time order specific to the third transmission channel . The MIMO radar device (700) according to Claim 13 , wherein the control circuit (720) is formed while two successive FMCW chirps of the first sequence are transmitted with the first phase, the first phase being a first FMCW chirp of the second sequence and the second phase being a subsequent second FMCW chirp of the second sequence and while four consecutive FMCW chirps of the first sequence are transmitted with the first phase, the first phase assigns two consecutive FMCW radar chirps of the third sequence, followed by the second phase, the two more consecutive FMCW radar chirps of the assigned to the third sequence. The MIMO radar device (700) according to any one of the preceding claims, further comprising a receiver circuit (730) comprising at least one receiver channel (732); wherein the at least one receiver channel (732) is designed to receive a received signal Reflections of the first sequence of FMCW radar chirps and the second sequence of FMCW radar chirps correspond. The MIMO radar device (700) according to Claim 15 , wherein the receiver circuit (730) is designed to perform a first FFT of the received signal to generate a plurality of first bins; perform a second FFT using the first bins to generate a plurality of second bins, and combine the first and second bins to obtain a combined range/Doppler map for the plurality of transmit channels. The MIMO radar device (700) according to Claim 15 , wherein the receiver circuit (730) is configured to identify each of the transmit channels based on evaluating mutual distances and/or phase differences between a plurality of second bins associated with the same first bin of the combined range/Doppler map. The MIMO radar device (700) according to Claim 16 , wherein the receiver circuit (730) is configured to identify a first of the plurality of transmit channels based on a pair of second bins with a predetermined mutual bin distance and a predetermined phase difference associated with the predefined Doppler division multiplexing scheme, and identify at least a second one of the plurality of transmission channels based on amplitudes of the remaining second bins associated with the same first bin. The MIMO radar device (700) according to Claim 18 , where the predetermined phase difference is 90°. A MIMO radar method, the method comprising: Sending a first sequence of FMCW radar chirps (752) over a first transmission channel (712-1) such that the start and / or stop frequencies of each of at least a portion of the FMCW radar chirps of the first sequence are different from one another, Transmitting a second sequence of FMCW radar chirps (752) over a second transmission channel (712-2), wherein the start and / or stop frequencies of each of at least a portion of the FMCW radar chirps of the second sequence are different from one another, wherein the first and second sequences of FMCW radar chirps are transmitted simultaneously; and Controlling the first and second transmit channels (712-1; 712-2) to adjust phases of the FMCW radar chirps of the first sequence and the second sequence according to a predefined Doppler Division Multiplex, DDM, scheme. The MIMO process according to Claim 20 , wherein the start and stop frequencies of respective subsequent FMCW radar chirps (752-n, 752-(n-1)) of the first and second sequences are increased or decreased by a predefined frequency offset.

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