DE102015012090A1 - Method for low-interference and ISO26262 compliant operation of an inductive rotation angle sensor with m outputs - Google Patents

Abstract

The invention relates to a method for operating and evaluating an inductive rotation angle sensor head (303) with m inductive reception loops and with a field-inducing excitation loop and with a rotor with a rotor loop that distorts the field of the exciter loop. A signal generator (320) generates an exciter signal (322). An electrical energy source (301) modulated by the exciter signal (322) excites the exciter loop. A signal transformation block (307) performs a signal coordinate transformation on the output signal vector (304) of the rotational angle sensor head (303), consisting of the output signals (304) of the m receive loops, to form n transformed signals (308) of greater significance. Synchronous demodulation and / or scalar product formation per transformed sub-signal (308, i) of the transformed signals (308) on the one hand and the exciter signal (322) or a correlated control signal for the synchronous demodulation (321) on the other hand each result transformed signal (308, i), a level signal (310, j) of k level signals (310).

Description

introduction

Various inductive rotation angle sensors are known in the prior art. An example is here DE 19738839 A1 , Such sensors are also used, for example, in safety-relevant applications. Various standards are known for the requirements of such safety-relevant sensors. Exemplary here would be the Standard ISO26262 for the automotive sector or the Standard ISO62508 , which generally defines various Safety Integrity levels, ie safety classes.

Thus, a rotation angle sensor, in particular a rotation angle sensor, which on an inductive measuring principle according to DE 19738836 A1 In order to meet the requirements of such standards, the evaluating electronics must meet certain requirements resulting from the difficulties of such rotational angle sensor systems.

For a better introduction to the in the DE 19738836 A1 technology used, the basic principle of such an inductive rotation angle sensor is briefly described again at this point to work out in this context, the requirements of the evaluation of such an inductive rotation angle sensor and represent the problems.

One of the DE 19738836 A1 corresponding inductive rotation angle sensor is rotationally symmetric in the first approximation with an n-fold rotational symmetry (see 1 ). It typically consists of a carrier, for example of a conventional board material, on which meandering, typically three loops, the receiver loops (S1, S2, S3) are applied. The meander structures of the receiver loops (S1, S2, S3) have an angular periodicity which is the same and constant with regard to the circulation angle. One can here speak of an angular wavelength (Δφ) of the meander structures of the receiver loops (S1, S2, S3). The typically three receiver loops (S1, S2, S3) are rotated relative to one another by typically one third in the said angular wavelength (Δφ). If instead of three receiver loops (S1, S2, S3) m receiver loops (S1, S2, .... Sm) are used, then the twist is typically only 1 / m of the angular wavelength (Δφ) of the meander structures. In the following, three receiver loops are treated without limiting the number to this disclosure in terms of this disclosure. In addition, at least a fourth loop, the exciter loop ( 2 ), rotationally symmetrical to the symmetry axis (Sym) applied, which preferably has no meandering structure. This whole construction is typically used stationary and will be referred to hereinafter as a stator.

The axis of rotation, which is to be measured in its angle of rotation relative to the stator, pierces perpendicular to the image plane of the 1 the stator at its point of symmetry, which is the puncture point of the symmetry axis (Sym) of the system. On this axis of rotation is fixedly connected to this runner mounted, which itself is not electrically conductive as a carrier for a short-circuit loop, the rotor loop. This rotor loop preferably has the same structure and angular wavelength (Δφ) as the receiving loops (S1, S2, S3), which are preferably identical to each other, except for the rotation.

The exciter loop ( 2 ) now sends a primarily magnetic transmit signal that induces a current into the rotor. Because the exciter loop ( 2 is rotationally symmetrical with respect to the axis of symmetry (Sym) and thus rotationally symmetrical with respect to the axis of rotation, this induction in the rotor loop of the rotor is independent of its rotational angular position and thus of the twist angle of the rotor relative to the stator, which is to be measured. With this axis of rotation, which runs along the axis of symmetry (Sym), the rotor and thus the rotor loop is typically firmly connected. Therefore, induction depends on the angle of rotation of the axis of rotation with respect to the stator.

Likewise, the exciter loop ( 2 ) in the three receiver loops (S1, S2, S3) each have a current which should be the same for all receiver loops (S1, S2, S3), since they should have an identical geometry and only one rotation angle (Δφ / m) the axis of rotation, ie the symmetry axis (Sym), different. However, this plays because of the rotational symmetry of the exciter loop ( 2 ) not matter.

Depending on the rotational position of the rotor with its rotor loop to the three receiver loops (S1, S2, S3), however, the rotor induces a different current into the respective receiver loops (S1, S2, S3). The reverse induction can be compared to the induction in an asynchronous motor in which the squirrel cage induces fields in the three stator coils. (See, for example, column 5, lines 45 to 46 of DE 19738836 A1 )

As always, such a system is in reality not undisturbed and error-free to operate. Since such inductive rotation angle sensors are also to be used in safety-relevant applications, in particular in automobiles, it is necessary to classify and intercept these errors, insofar as they can not already be intercepted in the construction of the sensor element, in electronic post-processing.

This is in the DE 19738836 A1 (See, for example, column 5, lines 45 to 49 of the DE 19738836 A1 ) proposed to operate the three receiver loops (S1, S2, S3) in a star connection and to use the neutral point for the correction of disturbing influences, since this should have a value of 0 V due to the design of the system, without proposing a concrete method for this purpose. The excitation loop of the inductive rotation angle sensor head ( 303 ) is in the 2 with an alternating current through a field current source ( 301 ) is energized by means of an excitation current, which is an alternating current. As a result, when the inductive rotation angle sensor head ( 303 ) whose receiver loops (S1, S2, S3) each individually an AC signal. The sum of these alternating voltage signals ( 304 ) with correct function of said inductive rotation angle sensor head ( 303 ) in the case of star connection zero.

However, the compensation of the disturbances in this star connection is not sufficient. In particular, the star connection is not capable of common-mode signals which are transmitted, for example, via the excitation loop (FIG. 2 ) into the sensor system ( 300 ) are coupled with an EMC irradiation, reliably suppress. As a result, the gain of the subsequent amplifier is limited, thus reducing the robustness of the system.

That in the DE 19738836 A1 For this reason, disclosed methods for the correction of disturbing influences are in particular unable to reliably enable the detection of certain common-mode errors. For example, a lack of excitement of the excitation loop ( 2 ) not reliably recognizable and classifiable.

In addition, it was recognized in the invention that the in the DE 19738836 A1 proposed star wiring for interference suppression is also unfavorable because it leads to a favoring of common-mode signals and thus to increased sensitivity, for example, to magnetic alternating fields, which flood the receiver loops (S1, S2, S3). This additional common-mode signal, which should by definition have the same amplitude in all three receiver loops (S1, S2, S3), respectively results in a loss of amplitude bandwidth in the subsequent three input amplifier stages for the output signals of the receiving loops of the inductive rotation angle sensor head ( 303 ). If not enough amplitude reserve is provided at their inputs, the input amplifiers may overdrive. Since this override typically does not occur simultaneously in all three input amplifiers, the erroneous common mode signal may ultimately be distorted into a push-pull signal, resulting in a faulty rotation angle measurement result. However, such a faulty measurement as a result of a common-mode signal is to be regarded as a critical fault in safety-critical applications and therefore must be reliably prevented or at least classified and identified. This is not possible in the prior art.

A similar device is from the DE 19738839 A1 known.

All of these aforementioned devices are typically operated with a monofrequency signal. As a result, these devices are sensitive to narrow-band interferers.

In addition, all of these devices are unable to detect, classify, and safely distinguish more complex perturbation modes as such.

Object of the invention

• a) Gleichtaktfehler wie eingestreute Felder, Bruch der Erregerschleife, Bruch des Läufers etc. (im Folgenden symmetrische Fehlerzustände genannt)
• b) Gegentaktfehler wie Bruch einer Empfängerschleife oder Verbindung zu derselben, Schäden an den Verstärkereingängen, Kurzschlüsse auf dem Stator oder Läufer, EMV etc. (im Folgenden asymmetrische Fehlerzustände genannt)
• c) Weitere Fehler, die durch ein Fehlermodell beschrieben werden können.

It is the object of the invention to specify a method in which the signals provided by the sensor can be sufficiently different from the interfering background and can at the same time be evaluated so that the following errors can be detected, classified and distinguished:

• a) Common mode errors such as interspersed fields, breakage of the excitation loop, breakage of the rotor, etc. (hereinafter referred to as symmetrical error states)
• b) Push-pull errors such as breakage of a receiver loop or connection to it, damage to the amplifier inputs, short circuits on the stator or rotor, EMC, etc. (referred to below as asymmetrical fault states)
• c) Other errors that can be described by an error model.

To solve this task, the significance of the recognition must be increased.

This object is achieved by a method according to claim 1.

The faults a to c are in the prior art in such sensor systems, as described, not sure recognized, classified and are therefore not distinguishable.

Description of the invention

The sensor system ( 300 ) of the device according to the invention, which carries out the method according to the invention, is first with the aid of 2 described. In the following, all figures are simplified for clarity. Decisive for the claimed scope are only the claims. Essential parts of the invention relate to the disclosures DE 19738836 A1 and DE 19738836 A1 , Their contents are therefore part of this disclosure without claiming their disclosure content.

This disclosure claims the priority of German patent applications with file numbers DE 102014013948.4 and DE 102014013949.2 that are a full part of this revelation.

The exciter loop ( 2 ) of the inductive rotation angle sensor head ( 303 ) will be as before in the DE 19738836 A1 again with a field current through the excitation current source ( 301 ). Again, if the inductive rotation angle sensor ( 303 ) whose receiver loops (S1, S2, S3) each have an alternating voltage signal in the form of the output signals ( 304 ) of the inductive rotation angle sensor head, wherein each of the output signals ( 304 ) of the inductive rotation angle sensor head ( 303 ) is associated with a receiver loop (S1, S2, S3). The sum of these output signals ( 304 ) of the inductive rotation angle sensor head ( 303 ) with correct function of said inductive rotation angle sensor head ( 303 ) in the case of star connection zero. Typically, but not necessarily, the inductive rotation angle sensor head ( 303 ) already interconnected in a star shape. In the following, the invention will be described using the example of a star-connected rotation angle sensor head, without limiting the invention thereto.

The output signals ( 304 ) of the rotation angle sensor head ( 303 ) are stored in an EMC filter block ( 305 ) to filtered output signals ( 306 ) of the rotation angle sensor head ( 303 ) prefiltered, whereby typically only such signal components through the EMC filter block ( 305 ) are passed, the excitation signal ( 322 ), with which the excitation current source ( 301 ) controlling the excitation current for the exciter loop ( 2 ) or within the bandwidth of the excitation signal ( 322 ) lies. In this case, preferably one filter per output signal ( 304 ) and / or per receiving loop (S1, S2, S3) of the inductive rotation angle sensor head ( 303 ) intended.

According to the invention, however, it has now been recognized that it makes more sense, instead of the exemplary neutral point voltages, more precisely the m output signals ( 304 ) of the rotation angle sensor head ( 303 ) to evaluate other, more significant signals after a signal coordinate transformation for the purpose of error detection. For this purpose, it was recognized that the signals of the m neutral point voltages are m-dimensional vectors with m signals as vector elements, which on the one hand have a temporal and on the other hand an m-dimensional extent. It has been recognized according to the invention that an affine mapping of this m-dimensional output signal vector ( 304 ) to a more significant signal vector unlike the prior art allows the recognition of more complex error conditions. According to the invention, it was further recognized that after this vector signal coordinate transformation, pattern recognition can be performed by scalar product formation with prototypical error model signal vectors that can be used to detect more complex erroneous states. Only the combination of this signal coordinate transformation in the form of an affine mapping and the subsequent scalar product formation yields the task-related result of a increase in significance and robustness as well as the ability of the sensor system to classify and differentiate the system states.

In the exemplary case of the affine mapping, the exemplary output voltages at the outputs ( 304 ) of the inductive rotation angle sensor ( 303 ) with exemplary m = 3 receive loops (S1, S2, S3) evaluated. Since the induction of a common-mode disturbance in all m, three exemplary receiver loops (S1, S2, S3) should be the same, n, here by way of example three transformed signals of m, here by way of example three star voltages, should no longer have common-mode signal components. In the following, such preferred transformation is described without limiting this disclosure. It is important to realize that one of the output signals ( 304 ) of the rotation angle sensor head ( 303 ) must first undergo a transformation to the subsequent circuits to be adapted. This transformation should be done in a signal transformation block ( 307 ) which receives its input signals either directly from the rotation angle sensor head ( 303 ) as its output signals ( 304 ) or only preprocessed, for example filtered as a filtered output signal ( 306 ) of the rotation angle sensor head ( 303 ) via a filter block ( 305 ).

Such exemplary signal transformation, as an example of the affine signal vector transformation for significance enhancement, is performed in the said signal transformation block (FIG. 307 ) carried out in 4 again shown in more detail. From the filtered output signals ( 306 ) of the rotation angle sensor head ( 303 ) form, for example, three differential amplifiers ( 401 to 403 ) each have a signal voltage of a transformed signal ( 308 ) by respective difference formation.

This corresponds to the technical realization of the vector equation V → ₃₀₈ = A → ₃₀₇ V → ₃₀₄ + V → _off

Here, V _{304 is} the m-dimensional signal vector of the m signal voltages of the m output signal ( 304 ) of the rotation angle sensor head ( 303 ). The filtering of the output signals ( 304 ) of the rotation angle sensor head ( 303 ) to the filtered output signals ( 306 ), we have not always included in this description, as it is obvious to the person skilled in the art that the filtering can also mean a direct looping through of the signals, whereby the output signals ( 304 ) of the rotation angle sensor head ( 303 ) also directly ( 304 ) or filtered ( 306 ) at the subsequent signal transformation block ( 307 ) can be connected. V ₃₀₈ stands for the n-dimensional signal vector of the signal voltages of the n transformed signals ( 308 ). A ₃₀₇ represents a linking matrix between the m-dimensional signal vector of the m signal voltages of the m sub-signals ( 304 l ) of the output signals ( 304 ) of the rotation angle sensor head ( 303 ) or the m filtered output signals ( 306 ) on the input side of the signal transformation block ( 307 ) and the n-dimensional signal vector of the n signal voltages of the n partial signals ( 308 i ) of the transformed signals ( 308 ) of the signal transformation block ( 307 Full is an n-dimensional offset vector which is represented by the signal transformation block ( 307 ) is added to the result of the spin-stretch transform by means of the link matrix (A ₃₀₇ ). In the case of an arbitrary transformation, chosen only by way of example for the presentation of the invention, the linking matrix (A ₃₀₇ ) could, for example, be designed as follows:

Such an exemplary link matrix (A ₃₀₇ ), which is populated with ones, negative ones and zeros, can be realized by simple difference formation, for example in differential amplifiers. (please refer 4 )

wenn das Sensorsystem (300) ansonsten, wie im fehlerfreien Fall vorgesehen, symmetriert ist. Hierbei bedeutet symmetriert, dass alle Empfängerschleifen (S1, S2, S3) bis auf die besagte Verdrehung um die Drehachse, d. h. die Symmetrieachse (Sym), zueinander die gleiche Geometrie mit einer n-zähligen Symmetrie aufweisen und exakt zentrisch angeordnet sind und dass dies auch für die Läuferschleife gilt. Auch die Erregerschleife (2) ist in diesem idealen Fall bezüglich der Symmetrieachse (Sym) vollkommen symmetrisch angebracht.The transformed signals according to the invention ( 308 ) when applying the above link matrix (A ₃₀₇ ) and thus after signal vector transformation in the signal transformation block ( 307 ) preferably no more direct component. But it is also possible, for example, herauszuprojizieren the DC component as a separate signal. The exemplary link matrix (A ₃₀₇ ) is then no longer, in the exemplary case of m = 3, a 3 × 3 matrix as a m × n matrix, but a 3 × 4 matrix,

when the sensor system ( 300 ) otherwise, as provided in the error-free case, is symmetrized. This means symmetrized that all receiver loops (S1, S2, S3) except for the said rotation about the axis of rotation, ie the symmetry axis (Sym), have the same geometry with an n-fold symmetry to each other and are arranged exactly centric and this also for the runner loop applies. Also the exciter loop ( 2 ) is perfectly symmetrical in this ideal case with respect to the symmetry axis (Sym).

Due to the signal transformation, the voltage amplitude of the transformed signals ( 308 ) Furthermore, with respect to their useful signal component by one of the selected transformation (A ₃₀₇ ) dependent factor, here for example √ 3 , greater than the amplitudes of the output signals ( 304 ) of the inductive rotation angle sensor head ( 303 ) be. As a result, the level difference between the useful signal of the receiving loops (S1, S2, S3) after transformation and a possibly radiated interference signal, such as a radiated common mode noise, increased and on the other, and this is essential, the Aussteuerbereich the subsequent amplifier stages with respect to the useful signals better utilized.

If the mode of action of a possible interfering signal V _{sturgeon is} known, this can be fed into the signal path in an exemplary linear relationship, for example by means of a weighting vector G, which can be measured or calculated from construction data: V → ₃₀₈ = A → ₃₀₇ V → ₃₀₄ + V → _off + G → V _disturb

In the case of an exemplary common mode signal, then

The disturbance voltage that enters the rotation angle sensor head ( 303 ) with three output signals ( 304 ) is fed.

In this exemplary case, the matrix multiplication in the signal transformation block ( 307 ):

Here we have assumed G as G = (1, 1, 1). In the first three channels, the common-mode signal is reliably suppressed, while in the fourth channel, the interference signal is amplified. Thus, the error pattern becomes more significant and as such can be better recognized as it is separated from the wanted signals in the first two channels.

This previous example is therefore an excellent example to increase the significance of an affine vector signal coordinate transformation in the signal transformation block ( 307 ).

Typically, the device according to the invention is supplied in a vehicle with a simple DC voltage which is not symmetrical about the zero point. Therefore, in a preferred version of the device according to the invention, to each of the n voltages of the transformed signals ( 308 ) a DC voltage vector signal (V _off ), the offset, for example by differential amplifier ( 401 to 403 ) of the 4 added. In the exemplary case, this offset (V _off ) per sub-signal ( 308 i Preferably, half of the operating voltage (V _BD ) and / or is placed in the middle in the middle of the control range of the input amplifiers. Typically, the offset voltages are the same for all channels. This adder circuit according to the prior art is in 4 not shown for the sake of simplicity.

While the above-described significance increase by signal coordinate transformation in the exemplary form of a matrix multiplication of the output vector signal ( 304 ) of the rotary angle measuring head ( 303 This addition of the offset now represents a shift of the resulting signal vector. Such a multiplication of a vector signal with a matrix, here a predetermined link matrix (A ₃₀₇ ) for a hardwired signal coordinate transformation, in combination with an addition, here the addition of a fixed one Vector signal offsets (V _off ) by adder circuits, that is to say a quasi vectorial adder, represent an affine mapping of the output signal vector onto a new signal vector, here the transformed signals ( 308 ). In addition to these affine mappings, more complex mappings are possible. For example, a complete and / or partial feedback of the output vector of the Signal transformation block ( 307 ) consisting of the transformed signals ( 308 ) as part of the input signals of the signal transformation block ( 307 ) into the signal transformation block ( 307 ), allowing for nonlinear transformations that are part of this disclosure (see also 12 ). This feedback can be weighted and mixed, for example, by a weighting matrix (G ₂ ).

beschrieben werden.Following the increase in significance in the signal transformation block ( 307 ), the output signals of the signal transformation block ( 307 ), the transformed signals ( 308 ), checked for the presence of various characteristics. The most important feature is the presence of the measurement signal itself. The increase in significance in the signal transformation block ( 307 ) is therefore preferably carried out so that the measurement signal on other sub-signals of the transformed signals ( 308 ) is to be found as the level of the interference signals for error detection. The detection of measurement signals and error signals now takes place by comparison with prototypical signals. This is done by scalar product formation between the respective prototype signal of a prototypical signal vector, which can also be one-dimensional, and the respective output signal ( 308 i ) of the signal transformation block ( 307 ), the respective transformed signal ( 308 i ). The individual measurement signal level (V _mess ), which here is a measurement signal level for the respective prototype and thus can also be an error signal level, can be determined, for example, by the formula

to be discribed.

Preferably, as a prototypical signal, one of typically several, e.g. B. prototypic signals the excitation signal ( 322 ) used. Likewise, it is useful to have an exciter signal (which is shifted by a predetermined phase (eg + 90 °) and / or a predetermined time ΔT (FIG. 322 ) here as another prototypical signal to be able to determine, for example, the angle directly as arccos of the ratio of the resulting amplitudes.

The sum of the amplitudes of the signal voltages of the sub-signals ( 308 i ) of the transformed signals ( 308 ) should, for example, in the presented link matrix (A ₃₀₇ ) for the signal transformation to increase the significance in the signal transformation block ( 307 ) Be zero volts. If this is not the case, z. B. a sum error can be detected.

It has been recognized according to the invention that this sum of the amplitudes of the transformed signals ( 308 ), ie not the sum of the neutral point voltages of the output signals ( 304 ) of the rotary angle measuring head ( 303 ), as in the prior art itself, can be evaluated to evaluate the sensor state. According to the invention, it has been recognized that in this way in particular task-related fault conditions can be detected, classified and distinguished, which relate to the sensor and / or the evaluation in an asymmetric or symmetrical manner. Common mode errors affect the sum of the amplitudes of the signal voltages of those signals of the transformed signals ( 308 ) representing the measured values, after the increase in significance with the above-mentioned linking matrices (A ₃₀₇ ) in the signal transformation block ( 307 ) just not anymore. It is thus possible according to the invention to detect push-pull errors and their differentiation from common mode errors and at the same time distinguish them from other errors. This is not known in the prior art. In the apparatus according to the invention, therefore, the operating voltage is subtracted once or the said offset twice from the voltage sum, as a result of which the sum signal ( 326 ) is again in the middle of the control range subsequent amplifier.

This formation of the sum signal ( 326 ) by summation of the level signals ( 310 ) representing the output signals of the synchronous demodulator block ( 307 ), in which the scalar product formation takes place, takes place by way of example in a summing device ( 324 ) instead of. Incidentally, the formation of the sum signal again corresponds to a scalar product formation between the signal vector of the transformed signals ( 308 ) and a unit vector signal whose elements are occupied by constant one signals.

The summation signal ( 326 ) is, for example, by means of a first comparator ( 341 ) with a first threshold ( 327 ), for example in the form of a first reference _voltage (V _ref1 ) evaluated by comparison, wherein the reference _voltage (V _ref1 ) is preferably above said offset, and for example by means of a second comparator ( 342 ) with a second threshold ( 330 ), for example in the form of a second reference _voltage (V _ref2 ) is compared, which is preferably below the offset. This then typically results in a first result signal ( 334 ) and a second result signal ( 333 ). Is the voltage of the sum signal ( 326 ) outside of the first reference _voltage (V _ref1 ), ie the first threshold value ( 327 ), and the second reference _voltage (V _ref2 ), ie the second threshold value ( 330 ), defined, permissible sum voltage hose, so there is a corresponding to the object of the invention to be detected symmetry error, ie a push-pull error before. In addition, the device recognizes that this error is not a common mode error, so it is not a symmetric error state.

In the signal transformation block ( 307 ), the associated transformed signals ( 308 ) of the output signals ( 304 ) of the inductive rotation angle sensor head ( 303 ), as described above, transformed by means of the described affine mapping. As a result, depending on the chosen combination matrix (A ₃₀₇ ), the DC interference, as described, preferably removed and overdriving the subsequent amplifier inputs is thus less likely. The headroom in the subsequent signal path increases. Against this background, it often makes sense to carry out the signal transformation and scalar product formation multiple times and thus several stages in succession and to attach importance in the first transformation to preventing override, for example by a suitable choice of the link matrix (A ₃₀₇ ) , Here and below, only a single-stage transformation without feedback and filters is described for simplified representation. The three transformed signals ( 308 ) are then stored in a synchronous demodulator block ( 309 ) scalar multiplied and filtered to detect the excitation signal ( 322 ) of the excitation current source ( 301 ) to transform into a DC signal or low-frequency signal corresponding to the respective reception level.

A clock generator ( 320 ) supplies the excitation signal ( 322 ) for the excitation current source ( 301 ) and at the same time a control signal ( 321 ) for the synchronous demodulation by scalar product formation. Strictly speaking, the transformed signals ( 308 ) with the control signal ( 321 ) now in the form of the excitation signal ( 322 ) of the excitation current source ( 301 ) scalar multiplied and then low pass filtered. This is a procedure known from synchronous demodulators. Here, a scalar product between the excitation signal ( 322 ) of the excitation current source and the respective transformed signal ( 308 ) in the synchronous demodulator block ( 309 ) in the space-multiplex by n, formed here by way of example three separate synchronous demodulators. Of course, a time multiplex in a digital scan is also possible. Due to the low-pass filtering, signal components with higher frequencies are suppressed. Basically, instead of the use of scalar products, the use of any other linear forms is possible, provided that the orthogonality of two different signals under determinable conditions with respect to the difference of these signals in the form of the disappearance of the result (V _{mess, i} ) of the linkage of these signals achieve this linear shape. 3 shows a simple realization of a unit for scalar product formation. In this case, a first signal (SIG1) is multiplied by a second signal (SIG2) in a multiplier (M1) and filtered in the subsequent low-pass filter (F1) to the output signal (OUT). If one of the signals is a binary signal, for example the first signal (SIG1), then the multiplier can be realized very simply. For example, the multiplier may be an impedance converter or the like, which may be restructured by means of switches operated by the first signal (SIG1) so as to have a gain of 1, depending on the first signal (SIG1) and another time has a gain of -1. Typically, the said offset is previously subtracted and then added up again. If the second signal (Sig2) then also changes its sign in synchronism with the first signal (SIG1) after offset subtraction, the result is always a positive signal whose DC component is then separated by the subsequent low-pass filter. Other structures for scalar product formation are of course conceivable. The cut-off frequency of the low-pass filter (F1) is chosen so that it is below the minimum frequency (ω _min ) of the excitation signal (F1). 322 ) lies.

Incidentally, any two signals A and B are referred to as being orthogonal with respect to a synchronous demodulator in the sense of this disclosure if they are inserted in one of the synchronous demodulators of the synchronous demodulator block (SIG1) and the second signal (SIG2). 309 ) and the associated output signal, ie the associated level signal ( 310 ) then 0 would result. In reality, such a synchronous demodulator of the synchronous demodulator block ( 309 ) but a system noise and a control error show. This system noise and this control error can be achieved by feeding the exciter signal ( 322 ) in place of the first signal (SIG1) in the respective synchronous demodulator and simultaneously hold constant the second signal (SIG2) of the respective synchronous demodulator. The output of the synchronous demodulator would then ideally be zero, but in reality typically exhibits the said noise and control error.

In the example of the synchronous demodulator block ( 309 ), for example, each of the transformed signals ( 308 ) each having a first signal input (SIG1) each corresponding to a device for scalar product formation 3 connected. All second signals (SIG2) of these devices for scalar Product formation is typically done with the control signal for the synchronous demodulation ( 321 ) from the signal generator ( 320 ) connected. Typically, this control signal for the synchronous demodulation ( 321 ) with the excitation signal ( 322 ). All these three scalar product formation devices together form the synchronous demodulator block ( 309 ). Each of the three output signals (OUT) of the three scalar-product-forming devices is connected to one of the three level signals ( 310 ) connected.

Preferably, in a specific embodiment of the invention, the excitation signal ( 322 ) generated by the clock or signal generator ( 320 ) to form a bandwidth limited signal having a lower cutoff frequency ω _min and a different upper cutoff frequency ω _max , which should be different. In this case, the lower limit frequency ω _{min should be} greater than half the frequency bandwidth Δω = ω _max , -ω _min . The low-pass filter is now preferably designed so that it signal components with a frequency greater than the lower limit frequency ω _min and / or better greater than half the frequency bandwidth Δω / 2 reliably suppressed and frequency components below half the frequency bandwidth Δω / 2 as possible undamped. Here all statements refer to the frequency amounts. The result of this scalar product formation are three level signals ( 310 ) whose voltage value is typically a respective measure of the magnitude of the induction of the excitation signal ( 322 ) in the receiving loops (S1, S2, S3) of the inductive rotation angle sensor head ( 303 ).

According to the invention, instead of simple mono-frequency exciter signals ( 322 ), as in the prior art, now spread code based excitation signals ( 322 ) by the signal generator ( 320 ) generated. For this purpose, the relevant spreading code is typically sent repeatedly. This creates a band-limited signal. Particularly preferred is the use of digital spreading codes consisting of v ordered bits. Here, it is particularly preferable that the v bits of the spreading code have an equal number of 1-bits and 0-bits. Only this ensures that the generated excitation signal ( 322 ) does not contain the angular frequency (ω = 0 s ^-1 ). Rather, it is desirable that the generated excitation signal ( 322 ) has a minimum frequency (ω _min ) below which the amplitude of the exciter signal ( 322 ) disappears or is insignificant. This lower, minimum frequency (ω _min ) is determined by the sweep of the clock period T ₀ at which the bits of the spreading code are sent. The upper frequency (ω _max ) is determined by the sweep of the transmission duration of a spreading code (T ₁ = n · T ₀ ).

These spreading codes may be based on random signals and / or pseudo-random signals. Also, from a plurality of spreading codes by a random signal at the end of the transmission of a preceding spreading code, the spreading code to be sent next can be selected.

The use of spreading codes is particularly useful in particularly severely disturbed environments or for special reliability requirements.

In working out the invention, it has been found that it is beneficial to have the synchronous demodulation or the scalar product formation after the increase in significance in the signal transformation block (FIG. 307 ) in the signal path, because then the symmetrical noise levels are already removed and an over- or understeer of the measurement signal path is less likely. It has thus been recognized according to the invention that it makes sense to use the signal coordinate system from the m, here three output signals ( 304 ) of the inductive rotation angle sensor head ( 303 ) by said matrix coordinate transformation with the linking matrix (A ₃₀₇ ) and to change its dimension from m to n such that the significance of the resulting n-dimensional vector signal, here the transformed signals ( 308 ), maximum becomes. If the system were always affected by an asymmetric error in the same way, instead of the link matrix (A ₃₀₇ ), as exemplified above, another equalizing matrix operation with an alternative link matrix (A ₃₀₇ ') by the signal transformation block (FIG. 307 ). The link matrix (A ₃₀₇ ) given above is just one possible example of such an operation. In this disclosure, if a significance increase is mentioned, an affine mapping of the output signal vector from the output signals ( 304 . l ) of the output signal vector ( 304 ) of the inductive rotation angle sensor head ( 303 ) to an optimized transformed signal vector ( 308 ), in this disclosure this signal always transformed by the term ( 308 ) for easier understanding.

The invention thus consists of the two processing stages: significance maximization in stage 1 by means of the signal transformation block (US Pat. 307 ) and correlation with as few as possible interference-correlated exciter signals and error-model signals by means of the synchronous demodulator block ( 309 ) and the summer ( 324 ) in stage 2.

Such multi-stage, in particular two-stage processing is not known for inductive angle sensors from the prior art.

However, the use of spreading codes prolongs the dead time of the sensor system. It is therefore useful in some cases, the type of excitation signal ( 322 ) to switch dynamically in operation and, for example, to switch the length of the spreading code as a function of the allowable dead time and / or the necessary precision. A longer spreading code is used when greater precision or the suppression of significant disturbances is required, a shorter one, or none at all, when a short dead time is required.

Up to this time in the course of the signal processing in the device according to the invention, the processing was preferably carried out in space division. Ie. Up to this point in the signal path of the device according to the invention, a signal path is preferably present for each of the exemplary three receiver loops (S1, S2, S3), or preferably, three signal paths are always paralleled up to this point. It has been shown that it is advisable from this point on to carry out the processing in the signal path, preferably in time-division multiplex, from now on. For this purpose, for example, an analog multiplexer ( 311 ) cyclically between the k generated up to here, here three level signals ( 310, j ) around. The multiplexer output signal ( 312 ) is then preferably a controllable amplifier ( 313 ).

The variable gain amplifier ( 313 ) amplifies the multiplexer output signal ( 312 ) to the amplifier output signal ( 314 ). Its gain is amplified by an amplifier control signal ( 318 ) for all k, here three level signals ( 310, j ) are set once for the same amount for each measuring cycle. For the duration of the measurement, in which all k, here three level signals ( 310, j ) of the level signals ( 310 ), the gain of the amplifier ( 313 ) in each case and becomes only after all k, here three level signals ( 310, j ) once by a subsequent control logic ( 317 ) were evaluated by this control logic ( 317 ) readjusted at the end of the measurement run. This adjustment takes place in such a way that at least one of the k, here three level signals ( 310, j ) to a full scale of the following analog-to-digital converter (ADC, 315 ) without an override of the same leads. For this, the amplifier output signal ( 314 ) the analog-to-digital converter (ADC, 315 ), which receives the ADC output signal ( 316 ) forms from this. A the ADC ( 315 ) subordinate control logic ( 317 ) stores the conversion results of the ADC ( 315 ) in the form of the ADC output signal ( 316 ) during the measurement and evaluates these conversion results of the ADC at the end of the measurement cycle ( 315 ) in the form of the ADC output signal ( 316 ) out. The control logic ( 317 ) sets the gain of the amplifier ( 313 ) by means of the amplifier control signal ( 318 ) for each measuring cycle of all k, here three level signals ( 310, j ) up or down at the end of the measurement cycle, which causes the gain control loop ( 313 ) is closed. It should be noted that the information to be measured, namely the angle of rotation, in the height of the amplitudes of the level signals ( 310 ) is contained relative to each other and by the similar gain for all level signals ( 310 ) is not changed during a measurement run. The control logic ( 317 ) now controls via a multiplexer control signal ( 319 ) the multiplexer ( 311 ). The control logic ( 317 ) converts the k, here three states of the ADC output signal ( 316 ) during the measurement of the k, here three different level output signals ( 310, j ) now in the searched measured value ( 345 ) of the rotation angle.

According to the invention, it has now likewise been recognized that the value of the amplifier control signal ( 318 ) the gain of the amplifier ( 313 ) itself is a parameter which is suitable for determining a symmetrical error state, that is to say a common mode error, of the rotational angle measuring system according to the invention.

Is the value of the amplifier control signal ( 318 ) above a third threshold ( 335 ), the allowable maximum gain value, this may indicate, for example, a lack of excitation by a missing excitation current and / or an interrupted exciter loop or a missing runner.

Is the value of the amplifier control signal ( 318 ) below a fourth threshold ( 336 ), the allowable minimum gain value, this may be due to asymmetric irradiation or an interrupted receiver loop ( 2 ).

For this purpose, the amplifier control signal ( 318 ), for example by a third comparator ( 343 ) with said third threshold ( 335 ) and a third evaluation signal ( 337 ) generated.

The amplifier control signal ( 318 ) is, for example, by a fourth comparator ( 344 ) with said fourth threshold ( 336 ) and a fourth evaluation signal ( 338 ) generated.

In addition, it can be checked by comparison with a seventh threshold and an eighth threshold, whether in fact always a level signal ( 310, j ) of the level signals ( 310 ) is in the upper control range, a second level signal ( 310, j ) of the level signals ( 310 ) is in the middle control range and a third level signal ( 310, j ) of the level signals ( 310 ) is located in the lower control area. This comparison is preferably made by the control logic ( 317 ) at the end of a measurement run. Until then, the level values of the individual level signals ( 310, j ) after amplification in the controllable amplifier ( 313 ) and digital conversion in the ADC ( 315 ), as described, from this control logic ( 317 ) are cached.

In its most general form, the invention thus relates to a device for operating and evaluating an inductive rotation angle sensor head (US Pat. 303 ), which typically has three, more generally inductive, reception loops (S1, S2, S3). Furthermore, in the most general form, the angle of rotation sensor head ( 303 ) via one or more field-inducing exciter loops ( 2 ) and a runner with one or more of the field of the exciter loop ( 2 ) distorting rotor loops. The device according to the invention has at least one signal generator ( 320 ), the at least one excitation signal ( 322 ) generated. A modulated source of electrical or voltage or energy source ( 301 ) excites the exciter loop ( 2 ) of the inductive rotation angle sensor head ( 303 ) by means of said exciter signal ( 322 ). In contrast to the prior art, the device according to the invention has at least one subassembly for increasing the significance of the output signal vector of the angle of rotation sensor head ( 303 ) on. This is done in the example discussed above in a signal transformation block ( 307 ). This sub-device leads here by way of example to an affine mapping of the output signal vector ( 304 ) of the rotation angle sensor head ( 303 ) with m output signals on n new signal lines of a further transformed signal vector ( 308 ) transformed signals ( 308 i ) out. In the example above these are the n, here three transformed signals ( 308 ). Preferably, the dimension of this new signal vector is equal to the dimension of the output signal vector, ie m = n. However, this is not necessarily so. Also, several pictures can be switched one behind the other. After this transformation mapping, the significance of the transformed signal vector ( 308 ) increased. Incidentally, it is conceivable - but usually not opportune - that outputs of this block ( 307 ), which are part of its output signal vector ( 308 ), again as part of its input signal vector ( 304 . 306 ), resulting in non-linear maps.

After this signal coordinate transformation in the signal transformation block ( 307 ) forms in each case a synchronous demodulator within a synchronous module module block ( 309 ) and a subsequent summation by means of a summer ( 324 ) a scalar product between a reference signal vector and the transformed signals ( 308 ) as the input signal vector of this synchronous demodulator block ( 309 ).

In the simplest case scalar product formation can be described as follows:

P_304,i:

Pegel des i-ten Ausgangssignals von m Ausgangsignalen des induktiven Drehwinkelsensorkopfes (303). Diese Pegel bilden den Ausgangssignalvektor des Drehwinkelsensorkopfes.

A_307,ik:

Transformationsmatrix zur Signalkoordinatentransformation des Ausgangssignalvektors des Drehwinkelsensorkopfes (303) im Signaltransformationsblock (307). Diese Transformationsmatrix ist vorzugsweise fest verdrahtet und führt zu einer Signifikanzsteigerung des sich ergebenden Signalvektors gegenüber dem Ausgangssignalvektor des Drehwinkelsensorkopfes (303).

O_307,ik:

Offset-Signalvektor zur Anpassung des transformierten Ausgangssignalvektors an die nachfolgende Signalverarbeitung.

P₃₂₂:

Pegel des Erregersignals (322). Hierbei wird der Pegel dieses Erregersignals (322) mit jeder Vektorsignalkomponente zu einem Zwischensignalvektor multipliziert.

F[]:

Tiefpassfilterfunktion des Filters (F1). Ohne dies näher auszudifferenzieren, handelt es sich hier typischerweise um eine integrierende lineare Filterung mit Tiefpasscharakter. Typischerweise wird die obere Grenzfrequenz dieses Tiefpassfilters so gewählt, dass sie unterhalb der Minimalfrequenz (ω_min) des Erregersignals (322) liegt. Das Ergebnis dieser Filterung sind die besagten Pegelsignale (310). Das j-te Pegelsignal (310,j) gibt insbesondere bei geeigneter, quadratischer Dimensionierung der Verknüpfungsmatrix A_i,j die Amplitude des Anteils im Pegel des j-ten Ausgangssignals (340,j) wieder, der mit dem Erregersignal (322) korreliert.

P₃₂₆:

Pegel des Summensignals (326) der Pegelsignale (310). Erst durch diese Summierung wird der gesamte euklidische Abstand zwischen dem Erregersignal (322) und dem Ausgangsignalvektor (304) des Drehwinkelsensorkopfes (303) bestimmt. Diese Summierung macht somit erst die Skalar-Produkt-Bildung komplett. Das Ziehen der Wurzel zur korrekten Bestimmung des euklidischen Abstands ist in der Praxis typischerweise nicht notwendig und zu aufwendig.

The symbols have the following meaning:

P _{304, i} :

Level of the i-th output signal of m output signals of the inductive rotation angle sensor head ( 303 ). These levels form the output signal vector of the rotation angle sensor head.

A _{307, ik} :

Transformation matrix for signal coordinate transformation of the output signal vector of the rotation angle sensor head ( 303 ) in the signal transformation block ( 307 ). This transformation matrix is preferably hardwired and leads to an increase in significance of the resulting signal vector compared to the output signal vector of the rotation angle sensor head (US Pat. 303 ).

O _{307, ik} :

Offset signal vector for adapting the transformed output signal vector to the subsequent signal processing.

P ₃₂₂ :

Level of the exciter signal ( 322 ). In this case, the level of this excitation signal ( 322 ) multiplied by each vector signal component to an intermediate signal vector.

F []:

Low pass filter function of the filter (F1). Without further differentiation, this is typically an integrating linear filtering with a low-pass character. Typically, the upper limit frequency of this low-pass filter is selected so that it is below the minimum frequency (ω _min ) of the excitation signal ( 322 ) lies. The result of this filtering are the said level signals ( 310 ). The jth level signal ( 310, j ) gives, in particular with suitable, square dimensioning of the linking matrix A _{i, j} the amplitude of the Proportion in the level of the jth output signal ( 340, j ), which with the exciter signal ( 322 ) correlates.

P ₃₂₆ :

Level of the sum signal ( 326 ) of the level signals ( 310 ). Only by this summation the total Euclidean distance between the excitation signal ( 322 ) and the output signal vector ( 304 ) of the rotation angle sensor head ( 303 ) certainly. This summation thus completes the scalar product formation. Dragging the root to correctly determine the Euclidean distance is typically not necessary and too expensive in practice.

The device according to the invention thus has each transformed signal ( 308 i ) of the transformed signals ( 308 ) at least one sub-device with two inputs (SIG1, SIG2), from the excitation signal ( 322 ) or a control signal for the synchronous demodulation ( 321 ) as the first input signal (SIG1) of the sub-device and the respective transformed signal (SIG1) ( 308 ) as the second input signal (SIG2) of the sub-device per transformed signal ( 308 i ) one of the sub-device and the respective transformed signal ( 308 i ) associated level signal of a total of k, but at least three, level signals ( 310 ) is generated by synchronous demodulation and / or scalar product formation. In this case, this sub-device can be in particular a synchronous demodulator.

In a particular embodiment of the invention, this structure can now be used to detect other errors and to distinguish them from simple failure modes such as common mode and differential mode. For this purpose, this expression has an error model signal generator ( 1100 ) containing an error model signal ( 1101 ) generated. For example, this can be thought of as a distorted excitation signal. For example, a foreign body in the sensor can lead to a phase shift of the signal response of the sensor in common mode and / or in push-pull.

This error model signal vector ( 1101 ), which typically consists of several, for example μ error model signals ( 1101 u ) is now used with the output signals ( 304 ) of the rotation angle sensor head ( 303 ) in a similar way.

In the simplest case, scalar product formation for the error model vector signal can be described as follows:

P_304,i:

Pegel des i-ten Ausgangssignals von m Ausgangsignalen (304) des induktiven Drehwinkelsensorkopfes (303). Diese Pegel bilden den Ausgangssignalvektor des Drehwinkelsensorkopfes.

A_307,ik:

Transformationsmatrix zur Signalkoordinatentransformation des Ausgangssignalvektors (304) des Drehwinkelsensorkopfes (303) im Signaltransformationsblock (307). Diese Transformationsmatrix ist vorzugsweise fest im Signaltransformationsblock (307) verdrahtet und führt zu einer Signifikanzsteigerung des sich ergebenden Signalvektors der transformierten Signale (308) gegenüber dem Ausgangssignalvektor der Ausgangssignale (304) des Drehwinkelsensorkopfes (303).

O_307,ik:

Offset-Signalvektor zur Anpassung des transformierten Ausgangssignalvektors an die nachfolgende Signalverarbeitung innerhalb des Signaltransformationsblocks (307).

F_1101,j:

Pegel des j-ten Fehlermodellsignals (1101). Hierbei wird der Pegel dieses Fehlermodellsignals (1101) mit jeder Vektorsignalkomponente zu einem Zwischensignalvektor multipliziert.

F[]:

Tiefpassfilterfunktion des Filters (F1). Ohne dies näher auszudifferenzieren, handelt es sich hier um eine integrierende lineare Filterung mit Tiefpasscharakter. Typischerweise wird die obere Grenzfrequenz dieses Tiefpassfilters so gewählt, dass sie unterhalb der Minimalfrequenz des Erregersignals (322) und des Fehlermodellsignals (1101) liegt. Das Ergebnis dieser Filterung sind die besagten Fehlerpegelsignale (1110). Diese geben die Amplitude des Anteils im Pegel des i-ten Ausgangssignals (304) wieder, der mit dem j-ten Fehlermodellsignal (1101) korreliert.

G₁₁₀₆

Pegel des Gewichtungsfaktorsignals. Dieser Pegel kann auch konstant eingestellt sein, also fest verdrahtet sein. Mit diesem Gewichtungsfaktorsignal erfolgt die Summierung der Skalar-Produkt-Ergebnisse in Form Fehlerpegelsignale (1110) aus dem Fehlersynchron-Demodulator-Block (1109) in der Fehler-Summier-Einrichtung (1124).

F₁₁₀₄:

Pegel des Fehlersummensignals (1104) der Fehlerpegelsignale (1110). Erst durch diese gewichtete Summierung in der Fehler-Summier-Einrichtung (1124) wird der gesamte euklidische Abstand zwischen dem jeweiligen Fehlermodellsignal (1101) und dem Ausgangsignalvektor (304) des Drehwinkelsensorkopfes (303) bestimmt. Diese Summierung macht somit erst die Skalar-Produkt-Bildung komplett. Diese Summierung findet typischerweise in der Fehler-Summier-Einrichtung (1124) entsprechend dem Pegel des Gewichtungsfaktorsignal (1106) statt. Das Ziehen der Wurzel zur korrekten Bestimmung des euklidischen Abstands ist in der Praxis typischerweise nicht notwendig und zu aufwendig.

The symbols have the following meaning:

P _{304, i} :

Level of the i-th output signal of m output signals ( 304 ) of the inductive rotation angle sensor head ( 303 ). These levels form the output signal vector of the rotation angle sensor head.

A _{307, ik} :

Transformation matrix for signal coordinate transformation of the output signal vector ( 304 ) of the rotation angle sensor head ( 303 ) in the signal transformation block ( 307 ). This transformation matrix is preferably fixed in the signal transformation block ( 307 ) and leads to an increase in significance of the resulting signal vector of the transformed signals ( 308 ) with respect to the output signal vector of the output signals ( 304 ) of the rotation angle sensor head ( 303 ).

O _{307, ik} :

Offset signal vector for adapting the transformed output signal vector to the subsequent signal processing within the signal transformation block ( 307 ).

F _{1101, j} :

Level of the jth error model signal ( 1101 ). Here, the level of this error model signal ( 1101 ) multiplied by each vector signal component to an intermediate signal vector.

F []:

Low pass filter function of the filter (F1). Without further differentiation, this is an integrating linear filtering with low-pass character. Typically, the upper limit frequency of this low-pass filter is chosen so that it is below the minimum frequency of the excitation signal ( 322 ) and the error model signal ( 1101 ) lies. The result of this filtering are the said error level signals ( 1110 ). These give the amplitude of the component in the level of the i-th output signal ( 304 ), which with the j-th error model signal ( 1101 ) correlates.

G ₁₁₀₆

Level of the weighting factor signal. This level can also be set constant, so be hardwired. With this weighting factor signal, the summation of the scalar Product results in the form of error level signals ( 1110 ) from the error synchronous demodulator block ( 1109 ) in the error totalizer ( 1124 ).

F ₁₁₀₄ :

Level of the error sum signal ( 1104 ) the error level signals ( 1110 ). Only by this weighted summation in the error totalizer ( 1124 ), the total Euclidean distance between the respective error model signal ( 1101 ) and the output signal vector ( 304 ) of the rotation angle sensor head ( 303 ) certainly. This summation thus completes the scalar product formation. This summation is typically found in the error totalizer ( 1124 ) according to the level of the weighting factor signal ( 1106 ) instead of. Dragging the root to correctly determine the Euclidean distance is typically not necessary and too expensive in practice.

This embodiment of the invention thus has per transformed signal ( 308 i ) at least one sub-device with two inputs (SIG1, SIG2), which from the error model signal ( 1101 ) as the first input signal (SIG1) of the sub-device and the respective transformed signal (SIG1) ( 308 i ) as a second input signal (SIG2) of the sub-device for each transformed signal ( 308 i ) one of the sub-device and the respective transformed signal ( 308 i ) associated error level signal ( 1102 ) is generated by synchronous demodulation and / or scalar product formation, which subdevice may be, in particular, a synchronous demodulator.

To complete the scalar product formation, the embodiment according to the invention has an error summing device ( 1124 ), which is a fault signal ( 1104 ) with the weighting factor signal ( 1106 ) Weighted summation of multiple error level signals ( 1110 ).

Finally, it has an error determination ( 1142 . 1141 ) containing the error sum signal ( 1104 ) or a signal derived therefrom with one or more error thresholds ( 1127 . 1130 ) compares whether the error sum signal ( 1104 ) is above or below the one threshold or between two thresholds. It then gives an error detection signal ( 1134 . 1133 ) depending on the comparison result.

From the preceding text, the following list of thresholds results so far First threshold ( 327 ) for detecting a too small sum signal ( 526 ) of the level signals ( 310 ) Second threshold ( 330 ) for the detection of a too large sum signal ( 326 ) of the level signals ( 310 ) Third threshold ( 335 ) for detecting an excessive amplifier control signal ( 326 ) Fourth threshold ( 336 ) for detecting a too small amplifier control signal ( 326 ) Fifth threshold ( 527 ) for detecting a too small second sum signal ( 526 ) of the second level signals ( 510 ) (This threshold will be later explained.) Sixth threshold ( 530 ) for detecting a too large second sum signal ( 526 ) of the second level signals ( 510 ) (This threshold will be later explained.) Seventh threshold for detection by the control logic ( 317 ), whether one of the Level signals ( 310 ) is located in the upper control area, whose Lower edge is defined by this seventh threshold. Eighth threshold for detection by the control logic ( 317 ), whether one of the Level signals ( 310 ) is located in the lower control range, whose Top edge is defined by this eighth threshold. Ninth threshold for detection by the control logic ( 317 ), if at least one the level signals ( 310 ) in the upper tax area but not yet in Override is whose allowable upper edge through this ninth threshold is defined. Further thresholds for evaluation of the measurement result ( 345 ) (explained below)

5 FIG. 10 shows the exemplary sequence of an exemplary measuring pass for measurement for exemplary three receiver loops (S1, S2, S3) in time-division multiplex.

In a first step ( 600 ) is passed through the multiplexer ( 311 ) a first of the k, here by way of example three level signals ( 310 ) and a first measured value ( 316 ) determined by the control logic ( 317 ) is stored.

In a second step ( 601 ) is passed through the multiplexer ( 311 ) a second of the k, here by way of example three level signals ( 310 ) and a second measured value ( 316 ) determined by the control logic ( 317 ) is also saved.

In a third step ( 602 ) is passed through the multiplexer ( 311 ) a third of the k, here by way of example three level signals ( 310 ) and a third reading ( 316 ) determined by the control logic ( 317 ) typically also stored, etc.

In the fourth step ( 603 ) becomes from these k, here three values the measured value ( 345 ) by the control logic ( 317 ), but typically not yet spent.

In the fifth step ( 604 ) the measured value ( 345 ) typically rated.

If the measured value, in particular in comparison with other measured values that were measured in parallel or before, and / or with other thresholds than in order by the control logic ( 317 ), in a sixth step ( 606 ) this is typically output and the system is typically not reparameterized.

If the measured value, in particular in comparison with other measured values, which were measured in parallel or before, and / or with other thresholds as not in order by the control logic ( 317 ), in a fifth step ( 605 ) For example, the last as i. O evaluated measured value is output and the system is typically reparameterized. In addition, an error message is typically sent to a higher-level system. However, it is also conceivable to record this error message in a counter and to have an error message issued only when a certain counter reading is reached by the control logic. The behavior of this counter can be designed so that a correct measurement resets the counter immediately and, if necessary, a corresponding all-clear is sent to a higher-level system. However, it can also be counted down to zero at the same or a different speed, as counted up.

• a. die die Frequenz des Erregersignals (322, 522) und/oder
• b. einen für die Erzeugung des Erregersignals (322, 522) verwendeten Spreiz-Code und/oder
• c. ein für die Erzeugung des Erregersignals verwendetes Pseudozufallssignal, insbesondere beispielsweise das Rückkoppelpolynom eines rückgekoppelten Schieberegisters zur Erzeugung eines Pseudozufallssignals, das ausgetauscht oder geändert werden kann.

The said reparameterization of the system typically affects

• a. the frequency of the excitation signal ( 322 . 522 ) and or
• b. one for the generation of the exciter signal ( 322 . 522 ) used spreading code and / or
• c. a pseudo-random signal used for the generation of the excitation signal, in particular, for example, the feedback polynomial of a feedback shift register for generating a pseudo-random signal which can be exchanged or changed.

After the sixth or fifth step, the measurement cycle in this example starts again with the first step.

In 6 an extension of the device according to the invention is shown. In addition to the first excitation current source ( 301 ) of the 2 is now a second excitation current source ( 501 ), which are superimposed on the first exciter current of the first excitation current source ( 301 ) a second excitation current into the exciter loop ( 2 ) of the inductive rotation angle sensor head ( 303 ) feeds. This second excitation current source ( 501 ) is replaced by a second excitation signal ( 522 ) controlled. The signal path is behind the inductive rotation angle sensor head ( 303 ) initially unchanged until the synchronous demodulator block ( 309 ), in addition, in parallel now a typically identical second synchronous demodulator block ( 509 ) is connected in parallel. Its second control signal for the synchronous demodulation ( 521 ) is now based on the second excitation signal ( 322 ). It may, for example, be identical to this. It is now preferred that the two exciter signals are selected in relation to one another such that the signal components of the output signals ( 304 ) of the inductive rotation angle sensor head ( 303 ), which respond to the first excitation signal ( 322 ), no signals at the second level signals ( 510 ), these are the outputs of the second synchronous demodulator ( 509 ), cause. Conversely, in this case, it is preferable that the signal components of the output signals ( 304 ) of the inductive rotation angle sensor head ( 303 ), which respond to the second excitation signal ( 522 ), no signals at the first level signals ( 310 ), these are the outputs of the first synchronous demodulator ( 309 ), cause. In that case, the excitation signals are referred to as orthogonal to each other.

The multiplexer ( 311 ) will now be expanded accordingly. Preferably, a measurement pass refers to the outputs of a synchronous demodulator block ( 309 . 509 ) It now makes sense to use the correct function of the sensor system with the additional second level signals ( 510 ) connected to the second synchronous demodulator block ( 509 ), also check. The sum of the amplitudes of the transformed voltages of the transformed signals ( 308 ) after the demodulation by the synchronous demodulator to the second level signals ( 509 ) must be zero volts here again. Also this sum of the amplitudes of the transformed voltages of the transformed signals ( 308 ) can thus be evaluated again to evaluate the sensor state. This also makes it possible to detect fault conditions (push-pull fault) that affect the sensor and / or the evaluation in an asymmetrical manner. By a first excitation signal ( 322 ) different second exciter signal ( 522 ) this can be done, for example, at a different frequency. In the apparatus according to the invention, therefore, the operating voltage is subtracted once or the said offset twice from the voltage sum, as a result of which the second summation signal (FIG. 526 ) is again in the middle of the control range subsequent amplifier.

This formation of the second sum signal ( 526 ) by summation of the second level signals ( 510 ) takes place in a second summing device ( 524 ) instead of.

The second sum signal ( 526 ) is, for example, by means of a fifth comparator ( 541 ) with a fifth threshold ( 527 ), for example also in the form of a first reference _voltage (V _ref1 ), which lies above said offset, and for example by means of a sixth comparator ( 542 ) with a sixth threshold ( 530 ), for example, also in the form of a second reference _voltage (V _ref2 ) compared, which lies below the offset. It then typically results in a fifth result signal ( 534 ) and a sixth result signal ( 533 ). Is the voltage of the second sum signal ( 526 ) outside of the first reference _voltage (V _ref1 ), ie the fifth threshold value ( 527 ), and the second reference _voltage (V _ref2 ), ie the sixth threshold value ( 530 ), defined, permissible sum voltage hose, so there is again according to the object of the invention to be detected symmetry error.

Of course it is also conceivable to pass the measurement on all synchronous demodulator blocks ( 309 . 509 ) to acquire. But this is disadvantageous in that the resolution of a block is then reduced.

Possibly. For example, the above-mentioned thresholds relating to the third and fourth and the seventh to ninth thresholds can be doubled accordingly.

It is particularly advantageous if at least one of the signal generators ( 320 . 520 ) a random signal as an excitation signal ( 322 . 522 ) used. Also, the use of a pseudo-random signal is possible.

A further advantageous embodiment is the use of a further spreading code for the generation of a further exciter signal ( 522 ) by the signal generator ( 320 ) or another signal generator ( 520 ). With a very long further spreading code, however, the reaction time of the sensor system increases. Therefore, it makes sense if in this case the other signal generator uses a shorter spreading code or no spreading code. The spreading codes or the basic clock of the signal generators ( 520 . 320 ) for generating the excitation signals ( 322 . 522 ) should preferably be chosen so that they are mutually orthogonal exciter signals ( 322 . 522 ) to lead. This can be ensured, for example, if the base clock (T ₀ ) of the first signal generator ( 322 ) the 2 × w-fold of the base clock (T ₁ ) of the second signal generator ( 522 ) and the spreading code length for the first spreading code of the first signal generator, assuming a digital spreading code, is 2 * w clocks and the first spreading code contains w 1 values and w 0 values. Also, the second spreading code for the signal generation by the second signal generator should have a length of an even number of clocks and have the same number of 1's and 0's. This ensures that the minimum frequency of the further excitation signal ( 522 ) above the maximum frequency of the first exciter signal ( 322 ) lies. Their spectra do not overlap with each other. As a result, the two frequency channels do not interfere with each other.

The lower limit frequency of the synchronous demodulator block ( 309 . 509 ) is below the smallest minimum frequency of the excitation signals ( 322 . 522 ). Through the output filters (F1) of the sub-devices of the synchronous demodulator block ( 309 . 509 ), the synchronous demodulation of the transformed signals ( 308 ) with the respective excitation signal ( 322 . 522 ), the frequency bands of the excitation signals ( 322 . 522 ) due to their low-pass characteristic from the output signal of the respective synchronous demodulator block ( 309 . 509 ), the level signals ( 310 . 510 ), filtered away.

It is also conceivable that a signal generator ( 520 . 320 ) or both signal generators ( 522 . 322 ) a monofrequent exciter signal ( 522 . 322 ), which is not in the frequency range of the other signal generator. In the case of two monofrequency signals, the frequency of the one excitation signal ( 322 ) an integer multiple of the frequency of the other exciter signal ( 522 ) or the other way around. Excitation signals ( 322 . 522 ) of different frequencies are incidentally the simplest examples of orthogonal exciter signals.

It makes sense if the control logic ( 317 ) in this case for the additional excitation signal ( 522 ) a second measured value ( 545 ).

• d. die Frequenz des Erregersignals (322, 522) betreffen, was einem Frequency-Hopping-Verfahren entspricht, und/oder
• e. einen für die Erzeugung des Erregersignals (322, 522) verwendeten Spreiz-Code betreffen und/oder
• f. ein für die Erzeugung des Erregersignals verwendetes Pseudozufallssignal, insbesondere das Rückkoppelpolynom eines rückgekoppelten Schieberegisters, betreffen.

Likewise, it is of course possible that one or more excitation signals ( 322 . 522 ), as described, after each measurement sequence, ie the measurement of all level signals ( 310 . 510 ). This can

• d. the frequency of the exciter signal ( 322 . 522 ), which corresponds to a frequency hopping method, and / or
• e. one for the generation of the exciter signal ( 322 . 522 ) apply spreading code and / or
• f. a pseudo-random signal used for the generation of the excitation signal, in particular the feedback polynomial of a feedback shift register concern.

In this way, by spatial multiplexing and / or time division multiplexing and / or frequency and / or spreading code multiplex multiple measured values ( 345 . 545 ) can be obtained, which can be compared with each other and with other thresholds in particular by mean and difference in an evaluation unit. As a result, it is possible, for example, to generate three measured values in the case of three measured values measured in parallel with three exciter current sources and three respectively associated exciter currents, which are controlled by three exciter signals, and thus generate a majority voting for the probably correct measured value. Here, for example, three level signals are generated by three synchronous demodulator blocks, which are then processed as already described. Each of the three branches is fed by a respective excitation signal, which originates from a respective one of three signal generators. Of course, it is conceivable that a signal generator generates several of these different excitation signals simultaneously.

Depending on the evaluation result, the excitation signals can then be changed by the control logic or another control unit. This makes it possible to detect a fault in a strong EMC radiation and to minimize by optimizing the excitation signals, as long as the disturbance does not meet all the excitation signals in the same way, which applies to typically narrow-band interferers.

As in many other inductive sensor systems, so it is in a sensor system according to the DE 19738836 A1 useful, the exciter loop ( 2 ) within an LC resonant circuit, for example, together with a parallel-connected resonant circuit capacitance (C) to operate. The advantage of the high resonance amplitude, however, has the disadvantage of a very narrow-band resonance frequency. Parallel to the modulation of the AC source is therefore also the modulation of the resonant frequency of the LC Resonant circuit itself by means of a modulatable capacitance ( 701 ) and / or a modulatable inductance ( 801 ) as a pathogen loop ( 2 ) possible and particularly preferred. This will be done with the help of 7 to 10 explained.

7 shows an exemplary system with a modulatable capacity ( 701 ).

This modulability can be achieved in the case of a modulatable capacity ( 701 ) happen, for example, by connecting and disconnecting partial capacities (C ₁ , C ₂ ). 8th shows by way of example a very simple modulatable capacity ( 701 ). The exemplary modulatable capacity ( 701 ) consists of a first partial capacitance (C ₁ ), which via the switch (SW ₁ ) in response to the second further excitation signal (C ₁ ) 722 ) of the second sub-capacitance (C ₁ ) is connected in parallel, or is separated from this. As a result, the resonant frequency of the LC resonant circuit in 7 be modulated. As a switch (SW ₁ ) are mechanical switches such as relays and micromechanical relays in particular, but also MOS and bipolar transistors and similar semiconductor switches that can fulfill the function of such a switch. These may also be more complex circuits that perform this function of a switch. The first excitation signal ( 322 ) of the first signal generator ( 320 ) is in this case also typically synchronously switched so that the first excitation current source ( 301 ) an exciter signal ( 322 ) receives that at the current resonant frequency of the LC resonant circuit of exciter loop ( 2 ) and modulatable capacity ( 701 ) corresponding to the value of the second further excitation signal ( 722 ) fits. Particularly preferred is a variant in which the second further excitation signal ( 722 ) has a period length which is a multiple of the period length of the first excitation signal ( 322 ).

The processing is analogous to that of the description of 6 in a second further synchronous demodulator block ( 709 ), which depends on a second further control signal ( 721 ) for the second synchronous demodulation second further level signals ( 710 ). The second additional control signal ( 721 ) for the synchronous demodulation is typically equal to the second further excitation signal ( 722 ). A second additional clock or signal generator ( 720 ) generates said second further exciter signal ( 722 ). Typically, this happens synchronously with the generation of the first exciter signal ( 322 ) in the signal generator ( 320 ).

It is obvious that the second further level signals ( 710 ) should be symmetrical and should have the same level. Deviations can be detected as previously as errors.

10 shows an exemplary system with a modulatable inductance ( 801 ) and a resonant circuit capacitance (C).

This modulability can be achieved in the case of a modulatable inductance ( 801 ) happen, for example, by connecting and disconnecting Teilinduktivitäten (L _i , L ₂ ). 10 shows by way of example a very simple modulatable inductance ( 801 ). The exemplary modulatable inductance ( 801 ) consists of a first Teilinduktivität (L ₁ ), which via the switch (SW ₁ ) in response to the second further exciter signal ( 822 ) of the second partial inductance (L ₁ ) is short-circuited and thus removed from the resonant circuit. As a result, the resonant frequency of the LC resonant circuit of exciter loop ( 2 ) and resonant circuit capacitance (C) are modulated. The excitation loop itself can consist of several such switchable Teilerregerschleifen, which has the same effect. As a switch (SW ₁ ) come here mechanical switches such as relays and micromechanical relays in particular, but also MOS and bipolar transistors and similar semiconductor switches, etc., which can fulfill the function of such a switch. The first excitation signal ( 322 ) of the first signal generator ( 320 ) is in this case typically also synchronously switched so that the first excitation current source ( 301 ) an exciter signal ( 322 ) receives that at the current resonant frequency of the LC resonant circuit of exciter loop ( 2 ) and modulatable inductance ( 801 ) and resonant circuit capacitance (C) corresponding to the value of the second further exciter signal ( 822 ) fits. Particularly preferred is a variant in which the second further excitation signal ( 822 ) has a period length which is a multiple of the period length of the first excitation signal ( 322 ).

The processing is again analogous to that of the description of 6 in a second further synchronous demodulator block ( 809 ), which depends on a second further control signal ( 821 ) for the second synchronous demodulation second further level signals ( 810 ). The second additional control signal ( 821 ) for the synchronous demodulation is typically equal to the second further excitation signal ( 822 ). A second additional clock or signal generator ( 820 ) generates said second further exciter signal ( 822 ). Typically this happens again synchronously with the generation of the first exciter signal ( 322 ) in the signal generator ( 320 ).

It is obvious that the second further level signals ( 810 ) should be symmetrical again and should have the same level. Deviations can be detected as previously as errors.

Of course, this modulation technique can also be used in combination by the simultaneous use of a modulatable capacitance and a modulatable inductance.

11 shows an embodiment of the invention, in which the detection of an error according to an error model signal ( 1101 ) he follows. This is generated in a fault model signal generator.

An error sync demodulator block ( 1109 ) generates the error level signals in the manner described above ( 1110 ) according to the correlating proportions of the error model signals ( 1101 ) in the transformed signals ( 308 ).

An error totalizer ( 1124 ) sums these error model signals ( 1101 ) weights with factors corresponding to the weighting factor signals ( 1106 ) to one or more error sum signals ( 1104 ).

One or more fault detection devices ( 1142 . 1141 ) compare the levels of these error sum signals ( 1104 ) with one or more error thresholds ( 1130 . 1127 ) and generate error detection signals ( 1133 . 1134 ) depending on the result of this comparison.

12 shows the system of 2 where the transformed signals ( 308 ) as inputs of the signal transformation block ( 307 ) are exemplarily fed back. This creates a non-linear transformation that is no longer an affine transformation.

Advantages of the invention

In contrast to the prior art, the device according to the invention and the method according to the invention can thus detect the errors of the task with high EMC robustness. It should be emphasized that the device is able to classify and differentiate common mode and differential mode errors and other errors, and in particular in the combination of claims 1 and 2 according to the invention, to detect them all safely.

LIST OF REFERENCE NUMBERS

2

Exciter loop;

300

Sensor system;

301

Excitation current source;

303

inductive rotation angle sensor head;

304

Output signals of the inductive rotation angle sensor head ( 303 ) with m partial signals ( 304 l );

304 l

l-th partial signal of m partial signals of the output signals ( 304 ) of the inductive rotation angle sensor head ( 303 );

305

EMC filter block;

306

filtered output signals ( 306 ) of the rotation angle sensor head ( 303 );

307

Signal transformation block;

308

transformed signals from n individual sub-signals ( 308 i );

308 i

i-th partial signal of n partial signals of the transformed signals ( 308 );

309

Synchrondemodulatorblock;

310

Level signals from k partial signals ( 310, j );

310, j

j-tes sub-signal of k sub-signals of the level signals ( 310 );

311

Analog multiplexer;

312

multiplexer output;

313

adjustable amplifier;

314

Amplifier output signal;

315

Analog-to-digital converter (ADC);

316

ADC output signal;

317

Control logic;

318

Gain control signal;

319

multiplexer;

320

Clock or signal generator;

321

Control signal for the synchronous demodulation;

322

Excitation signal;

324

Summing device for forming the sum signal ( 326 );

326

Sum signal of the level signals ( 310 );

327

first threshold;

330

second threshold;

333

second result signal;

334

first result signal;

335

third threshold, the maximum permissible gain value;

336

fourth threshold, the allowable minimum gain value;

337

third result signal;

338

fourth result signal;

341

first comparator for forming the first result signal ( 334 ) by comparing the first threshold ( 327 ) with the sum signal ( 326 );

342

second comparator for forming the second result signal ( 333 ) by comparing the second threshold ( 330 ) with the sum signal ( 326 );

343

third comparator for forming the third result signal ( 337 ) by comparing the third threshold ( 335 ) with the amplifier control signal ( 318 );

344

fourth comparator for forming the fourth result signal ( 338 ) by comparing the fourth threshold ( 336 ) with the amplifier control signal ( 318 );

345

searched measured value ( 345 ) of the rotation angle;

401

first differential amplifier for signal transformation of the filtered output signals ( 306 ) of the rotation angle sensor head ( 303 ) into a transformed signal ( 308 );

402

second differential amplifier for signal transformation of the filtered output signals ( 306 ) of the rotation angle sensor head ( 303 ) into a transformed signal ( 308 );

403

third differential amplifier for signal transformation of the filtered output signals ( 306 ) of the rotation angle sensor head ( 303 ) into a transformed signal ( 308 );

501

second excitation current source;

509

second synchronous demodulator block;

510

second level signals;

520

second clock or signal generator;

521

second control signal for the synchronous demodulation and / or scalar product formation;

522

second or further exciter signal;

524

second summing device for forming the second summation signal ( 526 );

526

second sum signal of the second level signals ( 510 );

527

fifth threshold;

530

sixth threshold;

533

sixth result signal;

534

fifth result signal;

541

fifth comparator for forming the fifth result signal ( 534 ) by comparing the fifth threshold ( 527 ) with the second sum signal ( 526 );

542

sixth comparator for forming the sixth result signal ( 533 ) by comparing the sixth threshold ( 530 ) with the second sum signal ( 526 );

545

second measured value of the angle of rotation;

600

first step of an exemplary measurement run (measurement of a first of the exemplary three level signals ( 310 ));

601

second step of an exemplary measurement passage (measurement of a second of the exemplary three level signals ( 310 ));

602

third step of an exemplary measurement run (measurement of a third of the exemplary three level signals ( 310 ));

603

fourth step of an exemplary measurement run (determination of the measurement result ( 345 ) before output by the control logic ( 317 ));

604

fifth step of an exemplary measurement run (evaluation of the measurement result ( 345 ));

605

sixth step of an exemplary measurement run (execution of the actions in the case of a correct measurement result, eg output of the measurement result ( 345 ));

606

Seventh step of an exemplary measurement run (execution of the actions in the event of an incorrect measurement result, eg output of a previous measurement result ( 345 ));

701

modulatable capacitance associated with the exciter loop ( 2 ) forms an LC resonant circuit;

709

second further synchronous demodulator block;

710

second further level signals;

720

second additional clock or signal generator;

721

second further control signal for the synchronous demodulation;

722

second further excitation signal;

801

modulatable inductance associated with the exciter loop ( 2 ) and another capacitance forms an LC resonant circuit;

809

second further synchronous demodulator block;

810

second further level signals;

820

second additional clock or signal generator;

821

second further control signal for the synchronous demodulation;

822

second further excitation signal;

1100

Fault model signal generator;

1101

Error model signals from μ error model sub-signals;

1101 u

u-th error model sub-signal of the u error model part signals of the error model signals ( 1101 );

1104

Error sum signal;

1105

Error detection signal;

1106

Weighting factor signal;

1109

Error synchronous demodulator block;

1110

Error level signal with q partial signals;

1110 r

rth partial signal of q partial signals of the error level signals ( 110 );

1110

Error level signal;

1124

Error summing means for forming the second error sum signal ( 1104 );

1127

error threshold;

1130

error threshold;

1133

Error detection signal;

1134

Error detection signal;

1141

Error detection;

1142

Error detection;

Δφ

Angle wavelength;

Δω

= ω _max - ω _min bandwidth of the exciter signal ( 322 ) (Amount);

ω _max

upper limit frequency of the exciter signal ( 322 ) (Amount);

ω _min

lower limit frequency of the exciter signal ( 322 ) (Amount);

A ₃₀₇

Link matrix for the signal transformation in the signal transformation block ( 307 );

A _{307, ij}

Element of the i-th column and j-th row of the link matrix (A ₃₀₇ ) for the signal transformation in the signal transformation block ( 307 );

C

Resonant circuit capacitance;

C ₁

first sub-capacity of the exemplary modulatable capacity ( 701 );

C ₂

second sub-capacity of the exemplary modulatable capacity ( 701 );

F1

Filter;

G

exemplary weighting vector for an exemplary error model. Here, the weighting vector indicates the factors with which an interfering signal with the disturbance signal _voltage V _{stub sets} in the three exemplary receiver _loops (S1, S2, S3);

k

Number of level signals ( 310 . 510 Etc.)

L ₁

first partial inductance of the exemplary modulatable inductance ( 801 );

L ₂

second partial inductance of the exemplary modulatable inductance ( 801 );

m

Number of receiver loops;

n

Number of transformed signals ( 308 );

Out

Output signal of the device for scalar product formation. This output signal is typically connected to one of the outputs of a synchronous demodulator block (e.g. 309 ) connected, so z. B. a level signal ( 310 l ) of the k level signals ( 310 );

S1

first receiver loop;

S2

second receiver loop;

S3

third receiver loop;

SIG1

first input signal of the device for scalar product formation

SIG2

second input signal of the device for scalar product formation

sm

mth receiver loop

SW ₁

Switch of the exemplary modulatable capacitance ( 701 ), which by the second excitation signal ( 522 ) can be operated.

sym

axis of symmetry

V _BD

operating voltage

V ₃₀₄

Signal voltage vector of the output signals ( 304 ) of the rotation angle sensor head

V _{304, i}

the ith signal of the signal voltage vector of the output signals ( 304 ) of the rotation angle sensor head

V ₃₀₇

Signal voltage vector of the output signals ( 308 ) of the signal transformation block ( 307 ), the transformed signals ( 308 )

V _{307, i}

the ith signal of the signal voltage vector of the output signals ( 308 ) of the signal transformation block ( 307 ), the transformed signals ( 308 )

V _mess

k-dimensional signal vector for coding the errors and measured values. Typically, some sub-signals of this signal vector preferably contain only measurement signals, while other error level signals correspond to error levels corresponding to one or more of the prototypical m-dimensional signal voltage vectors (V _proto ).

V _{mess, i}

i-th partial signal of the k-dimensional signal vector (V _mess ) for coding the errors and measured values. Typically, some sub-signals of this signal vector preferably contain only measurement signals, while other error level signals correspond to error levels corresponding to one or more of the prototypical m-dimensional signal voltage vectors (V _proto ).

V _off

exemplary offset voltage vector corresponding to the transformed signal voltage vector of the output signals ( 304 ) of the rotation angle sensor head ( 303 ) in the signal transformation block ( 307 ) is added.

V _proto

prototypical, m-dimensional signal voltage vector for the prototype of an error or a measured value

V _proto

the i-th signal of the prototypical, m-dimensional signal voltage vector (V _proto ) for the prototype of an error or a measured value

V _{sturgeon, i}

Noise signal (here, for example, one-dimensional). This may be, for example, an EMV irradiation.

QUOTES INCLUDE IN THE DESCRIPTION

This list of the documents listed by the applicant has been generated automatically and is included solely for the better information of the reader. The list is not part of the German patent or utility model application. The DPMA assumes no liability for any errors or omissions.

Cited patent literature

• DE 19738839 A1 [0001, 0014]
• DE 19738836 A1 [0002, 0003, 0004, 0008, 0010, 0010, 0012, 0013, 0021, 0021, 0023, 0104]
• DE 102014013948 [0022]
• DE 102014013949 [0022]

Cited non-patent literature

• Standard ISO26262 [0001]
• Standard ISO62508 [0001]

Claims (10)

Method for operating and evaluating an inductive rotation angle sensor head ( 303 ) by means of a device a. with m, but at least three inductive reception loops (S1, S2, S3) and b. with a field-inducing pathogen loop ( 2 ) and c. with a runner with a field of the exciter loop ( 2 ) distorting rotor loop, d. wherein in particular the star voltage of the receiving loops (S1, S2, S3) of the inductive rotation angle sensor head ( 303 ) in the ideal faultless case is zero volts, comprising the steps of: a. Generating at least one exciter signal ( 322 ) by at least one signal generator ( 320 ); b. Modulating an electrical power or voltage or energy source ( 301 ) by means of the exciter signal ( 322 ); c. Excitation of the exciter loop ( 2 ) of the inductive rotation angle sensor head ( 303 ) by the electrical power or voltage or energy source ( 301 ); d. Performing a signal coordinate transformation, in particular an affine signal coordinate transformation, for the output signal vector ( 304 ) of the rotation angle sensor head ( 303 ), consisting of the output signals ( 304 ) of the m, but at least three receiving loops (S1, S2, S3), for the formation of n, but at least three and preferably m, transformed partial signals ( 308 i ) as subsets of the amount of transformed signals ( 308 ) of greater significance, which may also entirely encompass this quantity, by means of a signal transformation block ( 307 ); e. Synchronous demodulation and / or scalar product formation, in particular by a synchronous demodulator, between the exciter signal ( 322 ) or a control signal for the synchronous demodulation ( 321 ) on the one side and a transformed partial signal ( 308 i ) of the transformed signals ( 308 ) on the other side to generate a level signal ( 310, j ) of a total of k, but at least three - preferably n or m, level signals ( 310 ), whereby preferably this each transformed partial signal ( 308 i ) of the m transformed signals ( 308 ) is preferably carried out once each; Method for operating and evaluating an inductive rotation angle sensor head ( 303 ) according to claim 1, comprising the additional steps: a. Generating a fault model signal ( 1101 ) by a fault model signal generator ( 1100 ); b. Synchronous demodulation and / or scalar product formation, in particular by a synchronous demodulator, between the error model signal ( 1101 ) on the one side and a transformed partial signal ( 308 i ) of the transformed signals ( 308 ) on the other side for generating partial signal ( 1110 r ) the error level signals ( 1110 ), whereby preferably this each transformed partial signal ( 308 i ) of the m transformed signals ( 308 ) is preferably carried out once each. Method for operating and evaluating an inductive rotation angle sensor head ( 303 ) according to claim 2, comprising the additional steps a. Formation of a Fault Signal ( 1104 ) by weighted or unweighted summation of several partial signals ( 1110 r ) the error level signals ( 1110 ) by means of an error summing device ( 1124 ). Method for operating and evaluating an inductive rotation angle sensor head ( 303 ) according to claim 3, comprising the additional steps, a. Comparing the error sum signal ( 1104 ) or a signal derived therefrom with one or more error thresholds ( 1127 . 1130 ) as to whether the error sum signal ( 1104 ) is above or below the respective error threshold or between two error thresholds, the comparison being determined by an error detection ( 1141 . 1142 ) he follows; b. Output of an error detection signal ( 1133 . 1134 ) depending on the result of the comparison by the error determination ( 1141 . 1142 ). Method for operating and evaluating an inductive rotation angle sensor head ( 303 ) according to one or more of claims 1 to 4 comprising the additional step, a. Generating at least one exciter signal ( 322 ) based on a spreading code by the signal generator ( 320 ). Method for operating and evaluating an inductive rotation angle sensor head ( 303 ) by means of a device b. with m, but at least three, inductive reception loops (S1, S2, S3) and c. with a field-inducing pathogen loop ( 2 ) and d. with a runner with a field of the exciter loop ( 2 ) distorting rotor loop, e. wherein in particular the star voltage of the receiving loops (S1, S2, S3) of the inductive rotation angle sensor head ( 303 ) in the ideal faultless case is zero volts, comprising the steps: f. Generating at least one exciter signal ( 322 ) by at least one signal generator ( 320 ) based on a spreading code; G. Modulating an electrical power or voltage or energy source ( 301 ) by means of the exciter signal ( 322 ); H. Excitation of the exciter loop ( 2 ) of the inductive rotation angle sensor head ( 303 ) by the electrical power or voltage or energy source ( 301 ); i. Performing a signal coordinate transformation, in particular an affine signal coordinate transformation, for the output signal vector ( 304 ) of the rotation angle sensor head ( 303 ), consisting of the output signals ( 304 ) of the m, but at least three receiving loops (S1, S2, S3), for the formation of n, but at least three and preferably m, transformed partial signals ( 308 i ) as subsets of the amount of transformed signals ( 308 ) of greater significance, which may also entirely encompass this quantity, by means of a signal transformation block ( 307 ); j. Synchronous demodulation and / or scalar product formation, in particular by a synchronous demodulator, between the exciter signal ( 322 ) or a control signal for the synchronous demodulation ( 321 ) on the one side and a transformed partial signal ( 308 i ) of the transformed signals ( 308 ) on the other side to generate a level signal ( 310, j ) of a total of k, but at least three - preferably n or m, level signals ( 310 ), whereby preferably this each transformed partial signal ( 308 i ) of the m transformed signals ( 308 ) is preferably carried out once each; Method according to one or more of claims 1 to 6 comprising the step, a. Generating an exciter signal ( 322 ) that a band limited signal having a nonzero upper cutoff frequency ω _max and a nonzero lower cutoff frequency ω _min having a frequency bandwidth Δω = ω _max - ω _min with respect to the magnitude of the frequency is, by the signal generator ( 320 ), wherein the frequency band thus defined does not include 0 Hz and wherein the lower limit frequency ω _{min of} the exciter signal ( 322 ) greater than half the frequency bandwidth Δω / 2 of the excitation signal ( 322 ). Method according to one or more of claims 5 to 7; characterized by, a. that the spreading code for the generation of the excitation signal ( 322 ) is a random signal or a pseudo-random signal Method according to one or more of the preceding claims 5 to 8, comprising the additional step; a. Generating at least one further exciter signal ( 522 ) based on a further spreading code by at least one signal generator ( 320 ) or by another signal generator ( 520 ); b. Modulating a further excitation current source ( 501 ) by means of the further exciter signal ( 522 ); c. Excitation of the exciter loop ( 2 ) of the inductive rotation angle sensor head ( 303 ) with at least one additional superimposed further exciting current through the further modulated excitation current source ( 501 ). d. Synchronous demodulation and / or scalar product formation, in particular by a synchronous demodulator, between the further exciter signal ( 522 ) or a control signal for the synchronous demodulation ( 521 ) on the one side and a transformed partial signal ( 308 i ) of the transformed signals ( 308 ) on the other side to generate a level signal ( 310, j ) of a total of k, but at least three - preferably n or m, level signals ( 310 ), whereby preferably this each transformed partial signal ( 308 i ) of the m transformed signals ( 308 ) is preferably carried out once, e. wherein the first and the further exciter signals ( 322 . 522 ) orthogonal to the synchronous demodulation and / or the scalar product formation in the synchronous demodulation block ( 309 . 509 ), where orthogonal means that when the respective two excitation signals ( 322 . 522 ) by a synchronous demodulator in the synchronous demodulation block ( 309 . 509 ) as its first input signal (SIG1) and its second input signal (SIG2), the output of this synchronous demodulator would then be zero except for control errors and system noise. Method according to claim 9, characterized by, a. that the excitation signal ( 322 ) and the further exciter signal ( 522 ) are based on two different spreading codes, the spreading code and the further spreading code, which have a different spreading code length.

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