DE102007003771B4 - Adaptive control device for an actuating device, in particular a clutch or a transmission - Google Patents

Abstract

Device for the adaptive control of an actuating device, in particular a clutch or a transmission for motor vehicles, which has a direct-current electric motor (14) which is supplied with energy via an H-bridge circuit (16), at least one transmitter which records the position of a of an adjustable member, a circuit (32) for controlling the position of this member and a current loop control circuit (34) connected in cascade with the position control circuit (32), the current loop control circuit (34) including a comparator (54) having a the position control circuit (32) receives a signal (52) for the current setpoint and a signal (24) for the current flowing in the motor and which generates a current error signal (58), characterized in that the current error signal (58) is fed to a PI correction circuit (60 ) is applied with antisaturation, which is applied to the H-bridge circuit (16) Tas t ratio signal (62) to determine the supply voltage of the electric motor (14), the correction circuit (60) of the current loop comprising a parallel connection of an amplifier (64) with gain Kp and an integrator (66) whose output is connected to an amplifier (68) with amplification factor 1/Ti, the outputs of these two amplifiers (64, 68) being connected through an adder (70) to a saturation circuit (72) which uses the duty cycle signal (62 ) and wherein a comparator (74) receives the output of the adder (70) and the output (62) of the saturation circuit (72) and generates a signal (76) which is applied via an amplifier (78) to the input of another comparator (80) connected to the input of the integrator (66) to reduce the integration when the output of the adder (70) is greater than the limit set by the saturation circuit (72). e is.

Description

The invention relates to a device for the adaptive control of an actuating device, in particular for a clutch or a transmission for motor vehicles, with this device having a DC electric motor whose energy supply takes place via an H-bridge circuit controlled by a microprocessor, at least one sensor which records the position of a of the member moved by the actuator, a circuit for controlling the position of this member and a current loop control circuit connected in cascade or series with the position control circuit.

From WO 2004/102 022 A1 a device for controlling an actuating device is known, which is based on the identification of the parameters of the electromechanical equations of the electric motor of the actuating device.

From the DE 100 62 025 C1 a drive control is known in which the oscillation of the speed controller at standstill is reduced. In the standstill control method, the control circuit for the operation of the electric drive is switched to a position control mode when the desired standstill position of the drive is reached. The position control mode has a speed control circuit with a proportional component and a superimposed position control circuit with a proportional component and an integral component, with the proportional gain of the speed control circuit being optimized for the moment of inertia of the electric motor.

The object of the present invention is to determine the characteristics of this device when the electric motor is powered through an H-bridge circuit, with a view to simplifying the function of the corrector of the position control circuit, taking into account the influence of the supply voltage and electrical resistance of the motor of the actuating device should be minimized.

Conventionally, the electromechanical devices for controlling a clutch or a gearbox generally comprise at least one electric motor, often of the DC motor type, a power module for the power supply of this electric motor, a position sensor for an output member of the actuator and a digital calculation circuit for the position control intended for this purpose is to calculate the ideal voltage to be applied to the motor in order to reach a given position in a given time and with a given precision.

The DC electric motors used in these devices usually have a very low electrical resistance to ensure the dynamic performance of the actuators, especially when the operating temperature is high. These electric motors include permanent magnets that can become demagnetized if the current fed to the motor is too high. In addition, excessive current can also cause a risk of interference for other systems connected to the vehicle's electrical power network. This in turn has the consequence that these motors cannot be started by applying a maximum electrical voltage and that the starting current is limited in particular by using a current loop control circuit connected in series with the position control circuit.

The position control circuit then calculates a current setpoint which can be saturated at values without risk to the actuator's electric motor.

However, it is quite difficult to set the operating parameters of these control circuits, due to the fact that certain parameters of the actuator's electric motor, such as its electrical resistance and the constant of the back electromotive force (which is proportional to the torque constant), depending on the operating conditions and, above all can vary significantly with temperature. This prevents nominal or arbitrary values from being assigned to the operating parameters of the control circuits.

The present invention is based on a continuous determination, in real time, of the parameters of the electric motor of the actuator, in a way that allows the correct adjustment of the operating parameters of the control circuits and eliminates the influences that could affect the performance of the control device and which are caused by the modifications of the operating conditions as well as the variations that result from the manufacture of the components of the control device.

To this end, it proposes a device for the adaptive control of an actuating device, in particular a clutch or a transmission for motor vehicles, which has a direct-current electric motor fed via an H-bridge circuit, at least one sensor which transmits the position of an element adjusted by the actuating device detected, a circuit for position control ment of this member and a current loop control circuit connected in series with the position control circuit, characterized in that the current loop control circuit comprises a comparator which receives a current reference signal provided by the position control circuit and a signal for the current flowing in the motor and which generates an error signal applied to a PI correction circuit with antisaturation providing a duty cycle signal applied to the H-bridge circuit to determine the supply voltage of the electric motor, the current loop correction circuit comprising an integrator connected in series to an amplifier with a gain of 1/ Ti connected in parallel to an amplifier of gain Kp and connected through an adder to a saturation circuit providing a duty cycle signal applied to the H-bridge circuit and a comparator providing the output ssignal of the adder and the output of the saturation circuit and generates a signal which is applied to the input of another comparator which is connected to the input of the integrator to reduce the integration when the output of the adder is greater than that through the saturation circuit set limits is..

This device allows the automatic start-up control of the electric motor and its power supply with a maximum electric voltage in order to obtain the maximum speed of the motor, while protecting the motor from excessive current and ensuring the stability of the current corrected by the corrector of the current loop control circuit systems.

The task of the current loop is to ensure that the electromagnetic torque to be supplied by the motor is supplied by applying the ideal voltage to the terminals of the motor with predetermined dynamic performances according to response time, overshoot and static deviation, and the effects of fluctuations in the electrical Compensate for the resistance of the motor and the influences of temperature and supply voltage.

Limiting the integration when the adder output is greater than the limits set by the saturation circuit makes it possible to avoid integrator runaway, which would make the current loop unstable.

According to one embodiment of the invention, the position control circuit comprises a filter which receives a position setpoint signal as an input and generates a filtered and gain-adjusted position setpoint signal which is applied to a comparator which also receives a position signal for the actuator which is applied via a state reconstruction filter to the position measurement is applied, and a position correction circuit with an anti-saturation mechanism which receives the output of the comparator and generates a limited current command signal which is applied to the input of the comparator of the current loop control circuit.

The filtering and the adjustment of the gain of the position reference signal make it possible to limit the mechanical stresses on the actuating device by reducing the abrupt changes in the reference value. The filtered and gain adjusted set point is compared to a position reading to produce an error signal, the position reading being reconstructed and gain adjusted from the output of a position transducer associated with the actuator.

This deviation signal is corrected by the corrector, which creates a current reference signal for a current limited between two positive and negative values to protect the motor. This corrector comprises at least one integrator state allowing the cancellation of the static error between the pre-filtered and gain-adjusted position command and the measured position value, and anti-saturation means for this integrator state when the desired current command is beyond the constraint limits for the applied current command.

This current reference represents, except for one parameter, an image of the electromagnetic torque that the motor has to deliver and it makes it possible to control the acceleration of the electric motor.

The position control circuit makes it possible to compensate in real time the variations in the torque constant of the electric motor as well as the effects of temperature on this torque constant, in order to obtain the desired dynamic performances in terms of overshoot, static drift and response time.

This device can be provided with means for calculating and adjusting the gains Ti and Kp of the amplifiers of the current loop correction circuit so that the corrected system is insensitive to low-frequency and high-frequency operation the changes in the resistance of the motor and the supply voltage of the system.

The means for calculating these gains include means for calculating the speed of the motor from the measured position of the actuator, for calculating and filtering the mean supply voltage of the motor by multiplying the duty cycle and the supply voltage of the H-bridge circuit and for filtering the voltage flowing through the motor current, the filtering operations making it possible to obtain the same phase delay on the various processed signals, in order to ensure the causality of the signals with one another, and means for calculating the electrical resistances and the constant of the back electromotive force of the motor at a measured ambient temperature from the filtered ones Values of motor current, motor speed and voltage applied to the motor.

The electrical resistance of the motor and its back electromotive force constant are calculated by the least squares method and the calculated values are applied to means for calculating mean values, with initial values previously stored and/or determined by learning when the device is first commissioned have been.

In addition, the device includes means for calculating the values of the gain factors Ti and Kp of the correction circuit of the current loop, starting from the average values of the resistance and the constant of the back electromotive force of the motor, from the measured value of the supply voltage of the H-bridge circuit and from the nominal values of this resistance and these Tension.

In addition, the device advantageously includes means for enabling the operation of these means for calculating the values of the gains only when the H-bridge circuit is at rest and when the actuator is not being controlled.

According to an embodiment of the invention, this device also comprises means for reduced operation when the measured value of the current in the motor is not available, for example in the event of a failure of the corresponding sensor, these means including means for disabling the correction circuit of the current loop and means for activating means for calculating the duty cycle for the control of the H-bridge circuit based on the current command signal, the measured value of the supply voltage of the H-bridge circuit and the calculated value of the electrical resistance of the motor.

In order to maintain the dynamic performance of the actuator when the measured value of the current in the motor is not available, the device can, as a variant, include means for calculating information on the motor current from the measured value of the supply voltage of the H-bridge circuit, the calculated values of the resistance and the constant of the back electromotive force of the motor, from the measured value for the position of the actuator and from the value of the duty cycle of the control of the H-bridge circuit.

If the actuator is a clutch actuator with stress compensation and adjustable preload, the device advantageously includes means for adapting the compensation stress to the wear of the clutch, so that the resulting stress on the motor of the actuator when opening and closing the clutch is approximately the same fails.

These means for adapting the compensating stress include means for estimating the instantaneous value of the engine load from the measured position of the actuator, the constant of the back electromotive force and the measured value of the motor current and the dynamic force of the motor calculated from its acceleration and its inertia, Means for determining the load trend as a function of the position of the actuator for each of the two directions of movement of the actuator, means for calculating the disengagement and engagement energies from these tendencies and means for deciding to change the preload of the balancer as a function of a deviation threshold between these energies, means for calculating the number of adjustment operations to be carried out as a function of the deviation between the disengagement and engagement energies, means for determining the imbalance between the stress compensation and the preload correction to be performed, and means for determining the command position signal for the actuator and for performing the predetermined number of adjustment correction operations upon deactivation of the actuator control system. The processes for changing the compensating preload take place in a partial area of the adjustment path in which the movement of the actuating device of the clutch does not result in any movement of the clutch.

This method, based on the calculation of tendencies and the resulting energies, has the advantage of giving excellent results even in the presence of noisy signals, as can be the case when deriving the speed of the electric motor.

Furthermore, estimating the constant of the back electromotive force and the resistance of the motor in real time allows access to information on the temperature of the winding and the temperature of the motor magnets. This information is useful for detecting abnormal use of the actuator motor and, for example, providing for less frequent gear ratio changes in the gearbox in order to slow down the overheating of the actuator motor.

To do this, the device includes means for calculating the temperatures of the winding and magnets of the motor, starting from the initial and current values of the resistance of the motor and the constant of the back electromotive force, the trends of which are known in each case as a function of the temperature, means for monitoring the variations in the calculated winding and magnet temperatures and means for reducing the actuation times and/or the number of actuations of the motor of the actuator when predetermined limits of these temperatures or their degrees of variation are reached.

In addition, the invention provides for the case in which the information on the position of the actuator is not available, for example due to a failure of the corresponding sensor, and it proposes a reduced operating mode of the actuator by replacing the measured position in the position control circuit with a starting from the current flowing in the motor is replaced by the recalculated position.

To do this, the device includes means for calculating a position of the actuator in the absence of a corresponding signal by calculating the electromotive force and the load on the motor from the measured value of the motor current, the calculated value of the counter electromotive force constant and the slope and ordinate at the zero point of the aforementioned Tendency of the load, then by calculation of the magnitude of the movement and by two successive integrations providing calculated values of the speed of the engine and the position of the actuator, which are again applied to the input of the means for calculating the load on the engine.

Means for limiting engine stress may also be provided. Although the position control circuit calculates a current command that saturates at values sized so that the motor cannot demagnetize, the electromagnetic torque resulting from the current limit multiplied by the torque constant can be greater than the value which the actuating device or the system controlled by the actuating device can accept.

Since the counter-electromotive force constant is known and the torque constant can be derived from it in real time, real-time modified current limits included in the current limits for protecting the motor against demagnetization can be determined based on information about the maximum electromagnetic torque to be applied, which is divided by the estimated motor torque constant.

The control device can be used with a DC motor with electronic switching, with or without magnets, with a synchronous or asynchronous motor.

The control device can be a control device for a clutch, a transmission or a window regulator, a wiper system, a sunroof, a seat adjustment or a device for automatically switching off or restarting the internal combustion engine of the stationary vehicle of the type known as "stop and go". type.

• - 1 eine schematische Gesamtansicht einer erfindungsgemäßen Vorrichtung;
• - 2a eine schematische Darstellung eines ersten Typs des Steuersignals der H-Brückenschaltung;
• - 2b eine schematische Darstellung der Zustände der Schalter der H-Brückenschaltung;
• - 3a eine schematische Darstellung eines anderen Typs von Steuersignalen der H-Brückenschaltung;
• - 3b die entsprechenden Zustände der Schalter der H-Brückenschaltung;
• - 4 eine schematische Darstellung der Positionsregelungs- und Stromschleifen-Schaltungen der erfindungsgemäßen Vorrichtung;
• - 5 eine schematische Darstellung einer Ausführungsart des Korrektors der Stromschleife;
• - 6 den Verlauf der Steuersignale der H-Brückenschaltung und ihre Auswirkungen auf die Geschwindigkeit der Betätigungseinrichtung;
• - die 7 und 8 Diagramme des Verstärkungsfaktors in Abhängigkeit von der Phase;
• - 9 eine schematische Darstellung der Mittel zur Berechnung der Parameter des Korrektors der Stromschleife;
• - die 10 und 11 schematische Darstellungen von zwei Varianten der Steuerungsvorrichtungen zu einer reduzierten Betriebsart bei Nichtvorliegen einer Informationen über den im Motor der Betätigungseinrichtung fließenden Strom;
• - 12 ein Prinzipschaltbild einer Betätigungseinrichtung einer Kupplung mit Beanspruchungsausgleich;
• - die 13 und 14 zwei Grafiken zur Veranschaulichung des Prinzips des Beanspruchungsausgleichs, wobei 13 die Veränderung der Beanspruchung am Kolben in Abhängigkeit vom Verstellweg der Betätigungseinrichtung und 14 die Veränderung des Widerstandsmoments am Motor der Betätigungseinrichtung in Abhängigkeit vom Verstellweg darstellt;
• - die 15 und 16 Diagramme entsprechend denen der 13 und 14 zur Veranschaulichung der Auswirkung des Verschleißes der Kupplung und seiner Folgen für den Motor der Betätigungseinrichtung;
• - 17 eine Grafik zur Veranschaulichung des Prinzips der Berechnung des Ungleichgewichts des Drehmoments am Motor beim Ein- und Auskuppeln;
• - 18 eine schematische Darstellung der Mittel zur Berechnung der Kraft des Motors und der Einstellung der Vorspannung der Betätigungseinrichtung der Kupplung;
• - 19 eine schematische Darstellung der Mittel zur Schätzung der Temperaturen der Wicklung und der Magnete des Motors;
• - 20 eine schematische Darstellung der Mittel zur Bestimmung der Position der Betätigungseinrichtung;
• - 21 eine Verdeutlichung des Prinzips der Berechnung der Motorlast.

The understanding of the invention as well as other features, details and advantages of the invention will be facilitated by the following description given by way of example with reference to the accompanying drawings. It shows in detail:

• - 1 a schematic overall view of a device according to the invention;
• - 2a a schematic representation of a first type of the control signal of the H-bridge circuit;
• - 2 B a schematic representation of the states of the switches of the H-bridge circuit;
• - 3a a schematic representation of another type of control signals of the H-bridge circuit;
• - 3b the corresponding states of the switches of the H-bridge circuit;
• - 4 a schematic representation of the position control and current loop circuits of the device according to the invention;
• - 5 a schematic representation of an embodiment of the corrector of the current loop;
• - 6 the behavior of the control signals of the H-bridge circuit and their effect on the speed of the actuator;
• - the 7 and 8th graphs of gain versus phase;
• - 9 a schematic representation of the means for calculating the parameters of the corrector of the current loop;
• - the 10 and 11 schematic representations of two variants of the control devices for a reduced operating mode in the absence of information about the current flowing in the motor of the actuating device;
• - 12 a schematic diagram of an actuating device of a clutch with stress compensation;
• - the 13 and 14 two graphs to illustrate the principle of stress balancing, where 13 the change in the stress on the piston as a function of the adjustment path of the actuating device and 14 represents the change in the resistance torque on the motor of the actuating device as a function of the adjustment path;
• - the 15 and 16 Diagrams corresponding to those of 13 and 14 to illustrate the effect of wear on the clutch and its consequences on the actuator motor;
• - 17 a graph illustrating the principle of calculating the torque imbalance on the engine during clutch engagement and disengagement;
• - 18 a schematic representation of the means for calculating the force of the engine and the adjustment of the preload of the actuating device of the clutch;
• - 19 a schematic representation of the means for estimating the temperatures of the winding and the magnets of the motor;
• - 20 a schematic representation of the means for determining the position of the actuating device;
• - 21 a clarification of the principle of calculating the engine load.

First up 1 Referring to reference, which represents a control device according to the invention, with a digital computer 10, which consists essentially of one or more microprocessors, memories and associated interfaces and which generates a signal 12 for controlling an electrical power supply of an electric motor 14, the power supply here being an H -bridge circuit 16 connected to control circuits 18 to which the control signals 12 are applied, the H-bridge circuit being connected to the + and - terminals of a source of electrical energy such as a motor vehicle battery.

The H-bridge circuit 16 includes four electronic switches, such as transistors 20a, 20b, 20c and 20d implemented as MOSFET transistors, connected as a bridge and controlled by circuitry 18, such as a level adapter, charge pump and wired Include logic circuits.

Via its output shaft, the engine 14 drives a control element for a clutch or a transmission or a system for automatically switching off and restarting the internal combustion engine of the vehicle when stationary (for example in a “stop and go” design).

The control signals 12 combine control information on the running direction of the motor, for example a Boolean logic signal, and information on the application of an electrical voltage to the terminals of the motor 14, for example as a PWM signal whose duty cycle is controlled. The voltage applied to the terminals of the motor 14 is then directly proportional to the duty cycle of the PWM signal.

The control signals 12 are generated by the system 10 on the basis of signals provided by sensors connected to the motor 14 and the H-bridge circuit 16, with a means 22 for measuring the electric current 24 flowing in the electric motor 14, a means 26 for measuring the electrical voltage 28 feeding the H-bridge circuit and a means for measuring the (absolute or relative) position 30 of a moving member moved by the motor 14 for actuating the associated system (clutch, gearbox, etc.).

The switching off of the supply voltage of the motor 14 by the transistors 20 of the H-bridge circuit can be done in two different ways be carried out in the 2a , 2 B on the one hand and 3a, 3b on the other hand are illustrated.

The in the 2a and 2 B The procedure shown consists of switching the switches 20a and 20d in phase with one another and in phase with the PWM control signal and, on the other hand, switching the switches 20b and 20c in phase with one another, but in phase opposition to the PWM control signal 12 and thus in phase opposition to all switches 20a and 20d.

As long as the duty cycle of this control signal is 50%, the voltage applied to the motor 14 is zero on average. To control the motor running in one direction the duty cycle is decreased below 50% and to control the motor running in the other direction the duty cycle is increased above 50%. A single control signal 12 is then sufficient to control the mean voltage level applied to the motor and the direction of rotation of this motor.

This type of control has the advantage of being simple, and the disadvantage of having a constant ripple current flowing in the motor, the amplitude of which is at its maximum, by alternately applying the supply voltage and its opposite to the motor. This leads to unnecessary heating of the motor and constant drawing of a ripple current from the vehicle battery.

Preferably, therefore, in the 3a and 3b illustrated second type of control is used, which uses two control signals 12a and 12b whose changes in 3a are shown schematically.

The first signal 12a is a PWM signal which defines the average voltage level applied to the motor via the duty cycle, while the second control signal 12b is a boolean type logic signal which defines the direction of rotation of the motor.

For a given running direction of the motor determined by the logic signal 12b, a switch in one of the phases of the bridge circuit is closed and a switch is kept open, while the switches located in the other phase of the circuit are switched in phase opposition. As in 3b 1, switch 6b is kept open and switch 6d is kept closed, whereupon switches 6a and 6b are closed alternately. When the logic signal 12b changes state to control the motor running in the other direction, the switch 6a is opened and the switch 6c is kept closed while the switches 6b and 6d are alternately switched.

This results in a phase in which the switches 6a and 6b are open and the switches 6c and 6d are closed, as a result of which the motor 14 is short-circuited. The current ripple in the motor is reduced because the supply voltage is alternately applied only once. In the quiescent state, the control duty cycle has a value equal to zero and the motor is shorted on itself, with no current being drawn or injected at the battery to power the H-bridge circuit.

In all cases: where Vp is the supply voltage at the terminals of the bridge circuit and α is the control duty cycle 12, which varies between +1 and -1 and whose sign indicates the sign of the torque to be applied to the motor in relation to the desired direction of rotation , then the voltage Um applied to the motor 14, neglecting the transient voltage drops of the transistors 20, is equal to Vp.α.

The DC motor 14 used in this actuator usually has a very low electrical resistance in order to be able to be powered by a rated current in the worst cases, especially when hot, when the resistance of the motor decreases due to the change in the resistivity of the copper of the windings on the largest, which can reach 40% with a temperature increase of 100°C. It follows that at low temperatures and with the same power supply voltage, the current drawn by the motor can be very high and can affect the proper functioning of the actuator, for example causing demagnetization of the motor magnets.

It is therefore advisable to limit the voltage applied to the motor at start-up or when the actuator encounters an obstacle in its path, such as a stop or hard point. Then the duty cycle of the PWM signal for controlling the switching bridge is to be controlled as a function of the current in the motor.

For this reason, the in 4 In the control device 10 according to the invention shown in more detail, a position control circuit 32 and a current loop control circuit 34, these two circuits being connected in cascade.

The position control circuit 32 includes a pre-filter circuit 36 which receives an actuator position setpoint signal 38 and generates a new setpoint signal 40 which is filtered and gain adjusted to limit the mechanical stresses on the actuator by reducing the abrupt setpoint changes, and which is applied to a input of a comparator 42 to be compared with an actuator position signal 44 applied to another input of the comparator 42 which provides a deviation or error signal 46 as an output. The position signal 44 is reconstructed and gain-adjusted by a state reconstruction filter 48 based on the position measurement based on the position signal taken at the motor 14 .

The deviation signal 46 is applied to the input of a correction circuit 50 which generates a signal 52 for a setpoint current limited between two positive and negative values in order to protect the electric motor 14 and limit the current drawn from the battery.

The correction circuit 50 includes at least one integrator state that allows for the cancellation of the static error between the filtered and gain-matched command signal 40 and the position signal 44, and anti-saturation or anti-drift means for that integrator state when the desired current command signal is outside the constraint limits of the current command signal 52.

This current command signal 52 provides a representation (save for one parameter which is the torque constant of the motor 14) of the electromagnetic torque that the motor is required to deliver and therefore allows the acceleration of the electric motor 14 to be controlled. The position control circuit 32 allows this to compensate in real time the variations in the torque constant of the electric motor 14 as well as the influences of the temperature on this constant in order to obtain the best possible dynamic performances in terms of overshoot, static drift and response time.

The current command signal 52 is applied to the input of the current loop control circuit 34 which includes a comparator 54 which receives the current command signal 52 and the motor current signal provided by the H-bridge circuit 16 connecting the terminals voltage 56 applied to motor 14 is generated. The comparator 54 supplies a current deviation signal 58 which is applied to the input of a correction circuit 60 which supplies a signal 62 for the desired value of the duty cycle limited between -100% and +100%, for example. With the exception of the voltage drops in the circuit breakers 20, this pulse duty factor setpoint signal makes it possible to achieve an average motor voltage Um that can be regulated by the duty cycle and is proportional to the supply voltage (Um=a.Vp).

The main task of the current loop control circuit 34 is to ensure the delivery of an electromagnetic torque required by the motor by applying an ideal voltage to the terminals of the motor in order to achieve given dynamic performances according to response time, overshoot and static deviation and the effects of the fluctuations in the electrical Compensate for the resistance of the motor and the influences of temperature and supply voltage.

The correction circuit 60 of the current loop control circuit is in 5 shown in more detail.

This correction circuit is a proportional-integral circuit with anti-saturation according to Astrom and Wittenmark. It comprises a parallel connection of an amplifier 64 with gain Kp and an integrator 66, followed by an amplifier 64 with gain 1/Ti whose output is connected to the output of amplifier 64 at the inputs of an adder 70 whose output is an unsaturated control signal for provides the duty cycle αu(t) to a saturation circuit 72 which provides the command duty cycle signal a(t) which is applied to the H-bridge circuit 16 . This saturation circuit comprises a comparator 74 which receives the signals au(t) and a(t) at its input and which supplies at its output a signal 76 which is applied by an amplifier 78 to one input of a comparator 80 from which another input receives the current error signal 58 receives and whose output provides a signal applied to the input of the integrator 66.

In this corrector, the non-saturated control signal αu(t) is the sum of the proportional response caused by the amplifier 64 and the integral response caused by the integrator 66. This signal αu(t) is saturated by circuit 42 to provide the signal for the duty cycle command value a(t). As long as the signal αu(t) remains within the limits set by the saturation circuits 42, which are generally equal to -1 and +1 (-100% and +100% with respect to the duty cycle), the difference from the reference signal α(t) is is equal to zero, so that the output signal 76 of the comparator 74 is also equal to zero. The integrator 66 then directly integrates the deviation signal 58.

When the unsaturated signal αu(t) exceeds the limits set by the saturation circuit 72, the non-zero difference signal 76 is subtracted from the error signal 58 by the comparator 80, reducing the integration until the signal αu(t) equals the setpoint signal a (t) is. The gain of amplifier 72 is used to adjust the antisaturation convergence speed. This circuit makes it possible to prevent the integrator 66 from running away, which would cause the current loop to become unstable.

6 illustrates the effect of the integral response in the correction circuit 60.

In this figure, curve A represents the variation of the current command signal as a function of time and curve B represents the variation in the current reading for the motor voltage coming from the proportional-integral correction circuit, while curve C represents the corresponding current reading for a dem voltage corresponding to the reference current, curve D for the corresponding motor voltage for the reference current, curve E for the motor voltage coming from the correction circuit, curve F for the speed of the actuator for the motor voltage corresponding to the reference current and curve G for the speed of the actuator for the motor voltage provided by the correction circuit.

To create these curves, the electric motor 14 was started with a stepped current setpoint signal A, the influence of the corrector 60 being compared with a constant voltage setpoint that is dimensioned such that the desired setpoint current results.

In phase I, which corresponds to the increase in current in the motor, the speed F or G of the actuator is equal to zero. The corrector applies a voltage E that is directly proportional to the output error signal, which is the difference between A and B, through gain Kp. This voltage gradually increases, due to the integration effect and the presence of residual error, until the current reaches and slightly exceeds the current command signal A.

When the electrical voltage is applied in an open-loop fashion as shown by curve D, it takes longer for the corresponding current shown by curve C to be established.

During phase II, which is an adjustment phase, curve B representing the measured current exceeds curve A of the current reference, which inverts the sign of the error signal and causes the curve E of the motor voltage coming from the corrector to decrease .

During phase III, which is a saturation phase, the measured current curve B coincides with the current command curve A and the motor starts.

During phase IV, the speed of the motor (curve G) is sufficient for the back electromotive force to cause a voltage drop, so that the motor can no longer draw the setpoint current A. The curve B for the measured current begins to decrease towards the value which corresponds to the mechanical load on the motor, while the curve G for the actuator speed continues to increase towards a maximum value. As the measured current curve B decreases, the error from the setpoint increases, resulting in a response from the integrator which causes the voltage curve E to increase to its maximum value.

Since the speed of the motor is proportional to its supply voltage, during the following phase V the curve G for the final speed is significantly higher than the curve F for the final speed, which results from curve D with the constant voltage.

It can be seen that the advantage of the integrator is to automatically regulate the motor start and the application of the maximum voltage in order to reach the maximum speed of the motor while at the same time protecting the motor from excessive current.

The invention provides for determining the gain parameters Kp and Ti of the amplifiers 64 and 68 of the current loop correction circuit 60 in order to obtain the desired dynamic performances of the system in terms of response time and overshoot, while ensuring the stability of the system .

To do this, the overall gain of the system, which here consists of the correction circuit, the H-bridge circuit and the actuator motor, must be strictly less than -1 over the entire frequency band.

In 7 , which is a graph of gain versus phase, or Black Nichols plot, versus frequency, the abscissa is phase in degrees and the ordinate is gain factor in dB of the transfer function of the open-loop corrected system has been entered, curve 90 resulting. The -1 gain point corresponds to the phase equals -180° and zero gain point shown at 92 in this diagram. The stability of the closed-loop corrected system corresponds to the phase domain 94 and the gain domain 96. These two domains in the direction of the arrows must be strictly positive for the corrected closed-loop system to remain stable. The intersection 98 of the curve 90 with the phase axis indicates the cut-off frequency of the looped corrected system, i.e. the velocity. As point 92 is progressively approached, the overshoot increases with a step response such as that shown in 6 is shown. Getting close enough to this point results in less and less damped oscillations with increasing stability degradation.

The system under study, consisting of the correction circuit, the H-bridge circuit and the electric motor, is subject to the effects of manufacturing variations and changes in physical quantities, such as the electrical supply voltage and temperature, which affect the electrical resistance of the circuit. These changes can, as in 8th shown, lead to a change in the transfer function of the corrected system with a shift of the curve 90 to 100, which in turn can lead to a reduction in the phase and/or gain ranges with a corresponding degradation in stability and dynamic performances.

9 represents the means for calculating the parameters of the correction circuit 60, making it possible to ensure the stability of the system and the desired dynamic performances.

wobei R für den elektrischen Widerstand des Motors, Im für den Motorstrom, Ke für die Konstante der gegenelektromotorischen Kraft und ω für seine Drehzahl steht.These means comprise a pre-filter circuit 102 which performs a phase-shifting matched to the signals inputted to it, with the signal 24 for the motor current, the signal 28 for the supply voltage of the H-bridge circuit, the signal 30 for the position of the actuator and a signal 104 corresponding to the delayed duty cycle signal 62 of a sufficient number of computation steps predetermined by a delay operator 106 . The function of this pre-filter circuit 102 is to calculate the engine speed information 108 from the actuator position signal 30 by means of a derivative filter when the signal 30 is an analog position measurement, or by measuring the time deviation between signals derived from a digital encoder are supplied, followed by low-pass filtering, this circuit 102 also having the task of calculating an average voltage signal 110 of the motor by taking the product of the signal relating to the duty cycle with the signal 28 relating to the supply voltage of the H-bridge circuit and that Filter the motor current signal 24 to obtain a filtered current signal 112 . The filtering resulting in signals 108, 110 and 112 is performed using a first order low pass filter with a time constant sized to give an equal phase delay in each signal relative to real magnitude. Thanks to the pre-filtering performed by circuit 102, the effects of motor inductance become negligible, so the electrical equation of motor 14 can be simplified to the following expression: $$\text{R}.\text{in the}+\text{Ke}.\omega =\text{Around},$$

where R is the electrical resistance of the motor, Im the motor current, Ke the back electromotive force constant, and ω its speed.

The values of R and Ke may be calculated by a conventional least squares method performed at 114 .

A median filter 116 calculates an average 118 of R and 120 of Ke from values 122, 124 initially stored for a reference temperature in two physical memories 126 when the device is first started up and corrected as a function of the ambient temperature. If no value of R and Ke is available in memory, for example in the case of a first start-up, then the filter uses default values for its initialization and once sufficient new values of R and Ke have been calculated, the averaging filter then calculates initial values for a reference temperature based on the measured ambient temperature and as a function of the respective development curves of R and of Ke as a function of temperature and stores them.

The calculated mean values 118 of R are subjected to a processing 128 which calculates the values of the parameters Kp and Ti of the current loop, starting from the mean values 118 for the resistance of the motor, from the power supply voltage Vp (28) of the H-bridge circuit and from calibratable nominal values for the Motor resistance and this supply voltage determined.

A signal 126 is also provided to the processing module to allow the current control parameters to be changed only when the H-bridge circuit is in the quiescent state and the motor 14 is not being driven so as not to affect the stability of the controlled system.

An averaging filter calculates an average of R and Ke from values for a reference temperature initially stored in two physical memories when the device is first started up and corrected as a function of the ambient temperature. If no value of R and Ke is available in memory, for example in the case of a first start-up, then the filter uses default values for its initialization, and once sufficient new values of R and Ke have been calculated, the averaging filter then calculates initial values for a reference temperature starting from the ambient temperature measured and as a function of the respective evolutions of R and of Ke as a function of temperature.

The calculated values of Kp and Ti are applied to the current loop correction circuit 60 which, from the current command and current error signals 52 and 58, generates the duty cycle command signal 62 which is applied to the H-bridge circuit 16.

In addition, the invention envisages reduced operating modes of this device in cases where certain information is missing, such as the measured value of the current flowing in the motor

10 , the 4 corresponds to, illustrates this reduced operating mode.

If the information on the current flowing in the motor is not available, the device switches to a reduced operating state in which the current regulation performed by the correction circuit 60 is deactivated and by means 140 for directly calculating the signal 62 for the duty cycle setpoint from the signal 52 for the current setpoint, by the measured value 28 for the supply voltage of the H-bridge circuit and by the calculated value 118 of the motor resistance.

In the steady state, the current in the motor and the duty cycle α for driving the H-bridge circuit are related by the following equation: $$\text{in the}=\left(\text{p}/\text{R}\right).a$$

wobei Ic das Stromsollwertsignal ist.The motor current Im is to be specified via the signal 52 for the desired current value in such a way that there is equality between Im and this desired value signal. This allows the duty cycle α to be calculated from the previous equation: $$a=\left(\text{R}/\text{p}\right).\text{IC},$$

where Ic is the current command signal.

This results in the case in 6 represented by curves C for motor current, D for motor voltage and F for motor speed, resulting in limited performances in terms of actuator speed since the benefit of the integration effect of the current loop correction circuit 60 is lost.

As a variant, the in 11 means shown are used to maintain the dynamic performance of the actuator, these means allowing the maintenance of the correction circuit 60 of the current loop in the absence of information on the measured value of the motor current, which is replaced by information on the motor current coming from the means 142 from the measured value 28 of the supply voltage of the H-bridge circuit, from the calculated value 118 of the motor resistance, from the calculated value 120 of the constant of the back electromotive force of the motor, from the measured position 30 of the actuator and from the value 62r delayed by an operator 144 by the correction circuit 60 supplied signal 62 for the target value of the duty cycle is reconstructed. The reconstructed information 24a about the motor current supplied by means 142 is used as in FIG 4 is applied to one input of comparator 54, the other input of which receives the current command signal 52 provided by the position control circuit.

• - das Produkt der Speisespannung der H-Brückenschaltung mit dem vorzeichenbehafteten Steuerungstastverhältnis,
• - minus das Produkt des elektrischen Widerstands, geschätzt durch die vorstehend beschriebene Methode, mit dem Wert des Stroms, der im vorangehenden Berechnungsschritt geschätzt wurde,
• - minus das Produkt der Konstante der gegenelektromotorischen Kraft, geschätzt durch die vorstehend beschriebene Methode, mit dem Wert der Drehzahl des Motors,
• - das Ganze dividiert durch den Wert der Induktivität des Motors.

In a first embodiment, the current is estimated from the integration of the quantity given by:

• - the product of the supply voltage of the H-bridge circuit with the signed control duty cycle,
• - minus the product of the electrical resistance estimated by the method described above with the value of the current estimated in the previous calculation step,
• - minus the product of the back electromotive force constant, estimated by the method described above, by the value of the speed of the motor,
• - the whole divided by the value of the inductance of the motor.

• - das Produkt der Speisespannung der H-Brückenschaltung mit dem vorzeichenbehafteten Steuerungstastverhältnis,
• - minus das Produkt der Konstante der gegenelektromotorischen Kraft, geschätzt durch die vorstehend beschriebene Methode, mit dem Wert der Drehzahl des Motors,
• - das Ganze dividiert durch den Wert des elektrischen Widerstands des Motors, geschätzt durch die vorstehend beschriebene Methode.

In a second embodiment, neglecting the influence of the motor inductance, the current is determined by calculating the quantity given by:

• - the product of the supply voltage of the H-bridge circuit with the signed control duty cycle,
• - minus the product of the back electromotive force constant, estimated by the method described above, by the value of the speed of the motor,
• - the whole divided by the value of the electrical resistance of the motor, estimated by the method described above.

The control device according to the invention can be used in particular in an actuating device for a clutch with stress compensation of the in 12 of the type shown schematically can be applied. The principle of such an actuator is known from the prior art and is for example in FR-A-2790806 described. The clutch E comprises a Belleville washer actuated by a hydraulically actuated release bearing 150 via a feed line 152 connected to the power cylinder 154 fixed to the actuator housing. The piston of the power cylinder is actuated by a rack 156 driven by a geared motor 158 . A balancing or biasing spring 160 acts through a roller 162 on a cam 164 mounted on the rack 156 .

In such an actuator, the device according to the invention controls the operation of the geared motor 158.

The principle of operation of this actuator is through the 13 and 14 illustrated.

13 15 represents the variation in the longitudinal stress acting on the rack 156 in the axis of the piston of the cylinder 154 as a function of the travel of the actuator measured on the axis of the piston of the cylinder 154. Curve 166 corresponds to the stress exerted by the clutch on rack 156 on the axis of the piston of cylinder 154 in the direction of clutch disengagement (i.e. disengagement), while curve 168 represents this stress in the direction of clutch E engagement.

Curve 170 represents the stress in the axis of the piston of cylinder 154 exerted by balance spring 160 and carried by cam 164 in the direction of clutch disengagement, while curve 172 represents this stress in the direction of clutch engagement.

In 14 Fig. 12 shows the variation in the resistive torque with respect to the engine 158 as a function of the displacement of the actuator in the axis of the piston of the cylinder 154, while curve 174 represents the engine load in the direction of clutch disengagement and corresponds to the difference between curves 166 and 172, whereas curve 176 represents engine load in the direction of clutch engagement and corresponds to the difference between curves 168 and 170 .

It should be noted that thanks to the presence of the compensating spring 170 in this device, the motor of the actuator supplies only a very small part of the release force.

the 15 and 16 show the same curves as that 13 and 14 , but after the wear of the clutch. The stress exerted on the rack by the clutch increases as the clutch wears, and curves 166 and 168 of FIG 13 shift upwards as at 166a and 168a in 15 shown versus curves 170 and 172 which represent the stress exerted by the balance spring. It follows that the moment of resistance against the motor increases in the direction of disengagement, as indicated by curve 174 in FIG 16 and decreasing in the direction of engagement as illustrated by curve 176a.

In order for the performances of the actuator to remain optimal, the balancing stress provided by the spring 160 must be adapted to the wear of the clutch so that the load on the engine in the direction of disengagement is reduced from the point of view of the force or stress of the engine load in the direction of engagement is equivalent to.

The compensating spring 160 is pretensioned and relaxes when the clutch is disengaged (shifting of the toothed rack 156 to the left in 12 ). A corresponding variation in the preload of the spring 160 varies the levels of the compensating stresses 170 and 172. The spring preload is modified by a mechanical system with a bistable rocker with ratchet that allows the initial length of the spring to be modified, with any displacement of the Rocker in one direction or the other causes an elementary change in the length of the spring in one direction or the other.

The change in the preload of the spring takes place in a part of the adjustment path of the actuating device which is outside of the useful adjustment path of the clutch. This adjustment is made by a certain number of movements of the rocker, this number being determined on the basis of a strategy for identifying the load and calculating the imbalance, the principle of which is given in 17 is illustrated.

In this figure, the 14 and represents the curves 174 and 176 of variation in the resisting torque with respect to the engine as a function of the travel of the actuator, two position thresholds Pmin at 180 and Pmax at 182 on the useful travel of the actuator, which is only affected by clutch wear, and in one Flow direction of the adjustment determined. For these position thresholds, a determination is made of the trend 184 of the engine load in the engaged state Emb and the trend 186 of this engine load in the disengaged state Deb, these trends being defined by the least squares best straight lines which give torque as a function of position.

Subsequently, the areas are calculated, which are between these trend lines shown here with dashed lines and the abscissa axis between the position thresholds, these areas in 17 are shown hatched and correspond to the engagement or disengagement energy. By comparing these energies, one obtains the imbalance of the engine load, after which it is possible to determine in which direction the preload of the balancing spring 160 must be modified and how many adjustment operations are necessary, that is, how many times the bistable rocker to adjust the length of the compensating spring is to be actuated.

The means of calculating the engine load, identifying the trend and deciding to change the preload of the spring are in 18 shown schematically.

wobei Kc die Drehmomentkonstante des Motors ist, die sich aus der Konstante der gegenelektromotorischen Kraft des Motors ergibt, während Cr für das Moment der Motorlast und Jm für die Trägheit des Motors steht.They comprise means 190 for calculating the motor load moment from the signal 44 for the position of the actuator, from the signal 24 for the current flowing in the motor and from the calculated value 120 for the constant of the back electromotive force of the motor using the mechanical equation of the electric motor : $$\text{jm}.\text{i.e}\omega /\text{German}=\text{Kc}.\text{in the}-\text{Cr},$$

where Kc is the torque constant of the motor resulting from the motor back electromotive force constant, while Cr stands for the moment of the motor load and Jm stands for the inertia of the motor.

The values 192 of the moment of the engine load and 194 of the speed of the engine are applied with the signal 44 for the position of the actuator to means 196 for determining the tendency of the load as a function of the position of the actuator for a given direction of travel defined by the direction of the Engine speed is defined, this trend being the best straight line determined by the least squares method.

For example, the disengaged direction of travel is defined as the positive direction resulting from positive values of engine speed resulting in increasing values of the actuator position signal 44, and the engaged direction of travel as the negative direction resulting from negative values of engine speed resulting in decreasing values of the control position signal. The values of the engine load moment in the disengaged direction and the values of this moment in the engaged direction are then measured, the slope and the ordinate at the zero point of these two straight lines corresponding to the tendency of the load in each of the two directions of travel for positions between the in 17 specified thresholds 180 and 182 of the useful adjustment path are determined.

The parameters 198, which are calculated for the load tendency in the disengaged running direction, and the parameters 200, which are calculated for the load tendency in the other direction, are applied to means 202 for calculating the disengaging and engaging energies. A control signal 204 supplied by the arithmetic means 196 indicates that the two parameter sets 198 and 200 have been calculated and that the disengaging and engaging energies can be determined with the aid of the arithmetic means 202 .

Once thus calculated, the declutching energy 206 and the engaging energy 208 are applied to means for determining the deviation between these energies, whereupon this deviation is applied to means 212 for determining the number of adjustment operations to be carried out for the adjustment of the balancing spring and to means 214. which determine the imbalance of the load compensation and which as a function of the deviations of the engagement and disengagement pelenergie decide to increase the preload of the spring, decrease this preload or leave it unchanged.

The results provided by means 212 and 214 are applied to means 216 for determining the position reference signal 38 which in turn is applied to the input of the position control circuit 32 of FIG 4 is applied, the means 216 also carrying out a certain number of adjustment operations provided by the means 212 following the decisions provided by the means 214 when they receive a signal 218 signaling the shutdown of the system for controlling the Actuating device brings about.

The real-time determination of the values of the back electromotive force constant and the electrical resistance of the motor allows access to temperature values of the winding and magnets of the motor. In particular, this information makes it possible to detect abnormal use of the engine 14 and to slow down its overheating, for example by reducing the frequency of gear changes in the gearbox.

19 12 schematically illustrates the method of estimating temperature in an actuator strategy using the initial recorded value 122 for the motor resistance at a reference temperature and a recalculated value 118 for that resistance at unknown winding temperatures. The evolution of the electrical resistance of the motor as a function of temperature is known and is applied at 222 to derive the winding and/or magnet temperature value 224 by evaluating the winding and/or magnet temperature with respect to the reference temperature. This deviation is then added to the reference temperature to obtain the final temperature of the windings and/or the magnets.

The same operation takes place at 226 starting from the initial value 124 of the motor back emf and the filtered average 120 of this back emf constant to obtain a value 228 for the temperature of the magnets, the curve relating this temperature to the aforesaid values connecting the constant of the electromotive force is known. The values 224 and 228 are applied together with the value of the ambient temperature T to means 230 for monitoring the temperature of the windings and the temperature of the magnets and for outputting a temperature error signal 232 to a monitoring system 234. If the means 230 determine that the temperature of the windings (or magnets) is above a calibratable threshold and/or that the difference between the temperature of the windings (or magnets) and the ambient temperature is greater than a calibratable threshold and/or or that the change in temperature of the windings (or magnets) at different points in time is greater than a predetermined degree of change, the temperature error signal 232 is applied to the monitoring system 234, which reduces the frequency and/or number of actuations of the electric motor via control signals 236 and which can output a warning signal 238 if necessary.

The device according to the invention is also capable of operating in reduced mode in the absence of the position signal for the actuator, using in the position control circuit position information calculated on the basis of the current flowing in the motor. This calculated position allows the control circuit to generate current command signals for the current loop control circuit and then to apply the desired voltage across the H-bridge circuit to the actuator's electric motor to effect the desired actuator displacement.

20 shows a schematic representation of the means used for this purpose.

The motor magnetomotive force 242 is calculated by means 244 from the filtered average value 120 of the motor back electromotive force and the measured value 24 for the current flowing in the motor. The calculation of the load imposed on the engine is performed by means 246 from the trend in the declutching load 198 and the trend in the declutching load 200 (indicating the values of the slope and the ordinate at the zero point of the load) as a function of the estimated values for the position of the control device and the speed of the engine.

The calculated load 248 is applied along with the magnetomotive force 242 to means 250 for calculating the amount of motion 252, which is defined as the difference between the magnetomotive force 242 and the motor load 248 divided by the motor's rotational inertia Jm. The motion quantity 252 is applied to an integrator 250 which provides a calculated value 256 for the speed of the motor, which in turn is applied to an integrator 258 which provides a calculated position 260 for the actuator. This calculated position and the The calculated engine speed is applied to the inputs of the computing means 246 as discussed above.

The calculated actuator position value is applied to the position control circuit 32 along with the position setpoint signal 38 .

21 explains the calculation of the engine load by means 246.

If the calculated value 256 for the speed of the engine is positive, ie greater than a calibratable threshold for the minimum speed, corresponding to the disengagement direction, and if the calculated position 260 of the actuator is between the minimum 180 and maximum 182 values of 17 then the load on the motor is calculated using the straight line 186, whose slope and ordinate at the zero point are known. If the position of the actuating device is less than the minimum value 180, the value that corresponds to the ordinate of the straight line 186 for the minimum position 180 is used as the value of the load. If the position of the actuating device is greater than the maximum position 182, the value that corresponds to the ordinate of the straight line 186 at the maximum position 182 is taken as the basis for the load value.

If the calculated value of the engine speed is negative, i.e. lower than a calibratable threshold with a negative value, corresponding to the engagement direction, and if the position of the actuator is between the minimum values 180 and 182, then the load is calculated using the straight line 184 whose slope and ordinate are known at the zero point.

If the position of the actuating device is less than the minimum value 180, the ordinate of the straight line 184 at the point of the minimum position 180 is used as the value for the load, while if the position of the actuating device is greater than the maximum position 182, the ordinate of the Straight line 184 at the point of maximum position 182 is taken as a basis.

If the calculated engine speed 256 stays between the two calibratable thresholds mentioned above, the load is considered to be zero, regardless of the actual position of the actuator.

Thanks to the identification and memorization of the torque constant and the load on the motor, as well as thanks to the knowledge of the rotational inertia of the motor, a position predictor is obtained that allows the actuator to be controlled when the position measurement signal is no longer available, with an accuracy significantly higher than that precision that could be achieved using the nominal values of the above parameters.

The device according to the invention also makes it possible to limit the stresses on the engine, avoiding any risk of damaging the actuator or the system it controls. This is because the position control circuit calculates a current command that saturates at values designed to prevent the motor from demagnetizing. However, the electromagnetic moment, as the product of the current limit multiplied by the torque constant, may be greater than the actuator or system being driven would allow. Since the constant of the back electromotive force of the motor is known and since from it the torque constant can be derived in real time, the current limits, which vary in real time and which are contained within the current limits for protecting the motor against demagnetization, can be calculated from information about determine the applied maximum electromagnetic moment, which is divided by the calculated value of the torque constant.

• - Schutz des Motors der Betätigungseinrichtung gegen zu starke Ströme sowie Schutz der Betätigungseinrichtung und des angesteuerten Systems gegen zu hohe Drehmomente,
• - größere Schnelligkeit der Betätigungseinrichtung,
• - Robustheit der dynamischen Leistungen gegenüber den Auswirkungen der Fertigungstoleranzen, der Temperatur und der Speisespannung,
• - Wegfall der Vorgaben im Zusammenhang mit einem besonderen Lernprozess,
• - Echtzeit-Identifizierung und Aufrechterhaltung der Leistungen einer Betätigungseinrichtung mit einstellbarem Ausgleich,
• - Möglichkeit eines reduzierten Betriebs in einer für den Fahrer transparenten Weise bei Nichtverfügbarkeit der Informationen, die durch die Stromschleifen-Regelungsschaltung oder durch die Positionsregelungsschaltung verwendet werden.

The invention basically offers the following advantages:

• - Protection of the motor of the actuating device against excessive currents and protection of the actuating device and the controlled system against excessive torques,
• - greater speed of the control device,
• - robustness of dynamic performances to the effects of manufacturing tolerances, temperature and power supply voltage,
• - Elimination of the requirements in connection with a special learning process,
• - real-time identification and maintenance of the performances of an actuator with adjustable compensation,
• - Possibility of reduced operation in a way transparent to the driver in the unavailability of the information used by the current loop control circuit or by the position control circuit.

Claims (6)

Device for the adaptive control of an actuating device, in particular a clutch or a transmission for motor vehicles, which has a direct-current electric motor (14) which is supplied with energy via an H-bridge circuit (16), at least one transmitter which records the position of a of an adjustable member, a circuit (32) for controlling the position of this member and a current loop control circuit (34) connected in cascade with the position control circuit (32), the current loop control circuit (34) including a comparator (54) having a the position control circuit (32) receives a signal (52) for the desired current value and a signal (24) for the current flowing in the motor and which generates a current error signal (58), characterized in that the current error signal (58) is fed to a PI correction circuit ( 60) with antisaturation applied to the H-bridge circuit (16). duty cycle signal (62) to determine the supply voltage of the electric motor (14), the correction circuit (60) of the current loop comprising a parallel connection of an amplifier (64) with gain Kp and an integrator (66) whose output is connected to an amplifier (68) with amplification factor 1/Ti, the outputs of these two amplifiers (64, 68) being connected through an adder (70) to a saturation circuit (72) which converts the duty cycle signal (62 ) and wherein a comparator (74) receives the output of the adder (70) and the output (62) of the saturation circuit (72) and produces a signal (76) which is fed through an amplifier (78) to the input of another comparator (80) connected to the input of the integrator (66) to reduce the integration when the output of the adder (70) is greater than the magnitude set by the saturation circuit (72). is. device after claim 1 , characterized in that the limits set by the saturation circuit (72) are equal to -1 and +1, or equal to -100% and +100% in terms of duty cycle. device after claim 1 or 2 , characterized in that the position control circuit (32) comprises a filter (36) which receives a position setpoint signal (38) on the input side and generates a filtered and gain-adapted position setpoint signal (40) which is applied to a comparator (42), which also receives a position signal (44) for the actuator which is applied to the position measurement via a state reconstruction filter (48), and in that the position control circuit (32) comprises a position correction circuit (50) with an anti-saturation mechanism which uses the output signal (46) of the comparator receives and generates a limited current setpoint signal (52) which is applied to the input of the comparator (54) of the current loop control circuit (34). Device according to any one of the preceding claims, characterized in that it comprises means for reduced operation when the measured value (24) of the current in the motor is not available, these means comprising means for disabling the current loop correction circuit (60) and means for activating means (140) for calculating the duty cycle for the control of the H-bridge circuit from the current setpoint signal (52), the measured value (28) of the supply voltage of the H-bridge circuit and the calculated value of the electrical resistance of the motor. Device according to one of the preceding claims, characterized in that the motor is a DC motor with electronic switching, with or without magnets, or a synchronous or asynchronous motor. Device according to one of the preceding claims, characterized in that it comprises an actuating device for a clutch, a transmission, a window winder, a wiper system, a sunroof, a seat adjustment or a system for automatically switching off or restarting the internal combustion engine when the motor vehicle is stationary.

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