DE10120746C1 - Method for evaluating laser-Doppler signals has pair of phase shifted signals mixed and used to generate a side band frequency - Google Patents

Abstract

The method for evaluating laser-Doppler signals involves generating two 90 degree phase shifted Doppler signals in two signal groups (31,32) and mixing (41) the Doppler signals with phase compensated signals of the mix frequency (fm) of the voltage controlled oscillators. The mixed signals is used to generate a side band frequency additional overlaid. The side band frequency is fed back to an Art PLL-regulating loop.

Description

The invention relates to a method and an arrangement for evaluating laser Doppler Signals, especially for homodyne laser Doppler anemometers (LDA) and laser Doppler Velocimeter (LDV) for contactless evaluation of the speed of particles or Bodies where mainly slow speeds, standstill or reversal of direction Are interested.

Laser Doppler devices detect the speed of a moving object by the Frequency change of the light scattered by the moving object is evaluated. The Doppler effect describes a linear relationship between the Doppler Frequency shift (Doppler frequency) and the object speed.

The cross-beam arrangement is a very frequently used configuration of laser Doppler devices (see, for example, KFpp: Lser Doppler anemometer for non-contact wind measurement over large distances [in: Laser and Optoelectronics 20 (3), 1988, p. 73 -83], in which the collimated light from a laser light source is divided by a beam splitter into two transmission beams, whose waists are superimposed on an object to be measured at angles of ± γ. The frequency-shifted scattered light is then collimated again by the transmission lens and with a Because of the different angles of incidence of the transmitted beams, the scattered light contains frequency shifts around the frequencies + f _D1 and -f _D2, if λ denotes the wavelength of the laser light, v the speed of the object and γ the angle of incidence of the transmitted beams on the object then a beat signal with the Doppler frequency is generated on the receiver z

f _D = (v / λ) sin (2γ), (1)

from which the object speed or after further calculations also the derived quantities, such as length and acceleration, can be determined.

One type of evaluation of laser Doppler signals is through the so-called heterodyne LDA given, in which there is a frequency shift in the optical beam path (Shift frequency) is generated and in which the received signal is the sum of Doppler frequency and the shift frequency. Heterodyne LDA allow that Forward / backward detection of object movement and measurement at small ones Speeds, including object downtime. The generation of the frequency shift is in  very complex in any case and therefore does not allow inexpensive laser Doppler devices. The evaluation of the detector signal can generally be done very differently, e.g. B. with Fast Fourier Transformation (FFT), counting method, etc., as used for example in C. Tropea [in: Laser Magazin 1/92, pp. 11-12] or B. Rupp [in: Laser und Optoelektronik No. 4, 1985, Pp. 362-375].

In the case of homodyne laser Doppler arrangements, the reception frequency is usually in the Range between 0 (material downtime) and a few MHz (maximum speed). Around however, to increase the signal-to-noise ratio, narrow-band filtering of the Received signal. This usually involves tracking systems and the principle of Frequency tracking demodulation used [F. Thirst and a .: Theory and practice of laser Doppler anemometry, G. Braun Verlag, Karlsruhe, 1987] because this is a real-time Allow demodulation of the received signal.

TH Wilmshurst and JE Rizzo [in: J. Phys. E: Sci. Instrum., Vol. 7 (1974), pp. 924-930] describe an Autodyne tracker arrangement for homodyne LDA. The received signal cos (ω _D t) is divided into two signal branches and mixed separately with two components of the output signal of an oscillator, which are mutually phase-shifted by 90 °. In a second mixing process, the cosine signal is differentiated from one signal path and mixed with the sine signal from the other signal path. Analogously, the sine signal differentiated in the second signal branch is mixed with the cosine signal from the first signal path. With a suitable sign of the two mixed signals, a summation amplifier then delivers an output signal which is proportional to the frequency difference between the oscillator and Doppler frequencies. This Autodyne tracker system minimizes the frequency difference between often very small Doppler frequencies and the oscillator frequency. To process the mixed signals with low frequencies, sideband suppression with narrow-band filters must always be carried out. Undesired suppression of the frequency range from 0 to a few kHz is unavoidable because the steepness of the band edge of a filter is limited. This means that low speeds cannot be measured.

Patent specification DE 197 42 608 C2 proposes a method in which two parts of the Doppler signal which are phase-shifted by 90 ° in two signal arms with two likewise orthogonally shifted signals, sin (ω _VCO t) and cos (ω _VCO t), of a voltage-controlled oscillator (VCO) are mixed and subsequently summed, so that one of the two sideband signals (cos (ω _VCO ± ω _D ) t) is suppressed. This means that signal processing is possible even at low Doppler frequencies by a comparator comparing the frequency-stable signal of an oscillator with the mixed received signal and controlling the VCO so that the frequency ω _VCO - ω _D = ω _{OS to} be evaluated always remains constant. The disadvantage of this method is that the generation of the VCO signals, which are also shifted by 90 °, has to be carried out in a relatively large frequency range (e.g. of ± 3 MHz). This phase shift is technically very complex and does not provide an exact 90 ° phase shift over the entire frequency range, so that the interfering sideband cannot be completely suppressed.

The invention has for its object a new way of evaluating Find laser Doppler signals by simple means at any size of the Doppler frequency a complete suppression of one by frequency mixing emerging sideband allowed.

According to the invention, the object is achieved in a method for evaluating laser Doppler signals, using a homodyne laser Doppler arrangement from the detector signal by phase shift two phase-shifted by 90 ° Doppler signals generated and in separate signal branches with the signal one voltage controlled oscillator are mixed, by summing the Signal branches a sideband of the mixed signal from Doppler frequency and oscillator frequency suppressed and the remaining sideband in a control loop for voltage controlled oscillator is returned so that a constant Output signal is available for evaluation, solved in that after the Generation of the two 90 ° phase-shifted Doppler signals in two Signal branches a mixture of the Doppler signals with in-phase signals of the Mixing frequency of the voltage controlled oscillator takes place after this mixing generated a relative phase shift of 90 ° between the two signal branches becomes additive before both signal branches to extract a sideband frequency are superimposed on the sideband frequency in a kind of PLL control loop Mixer is returned, the sideband frequency with a reference frequency  compared and in the event of deviations that supplied by the voltage-controlled oscillator Mixing frequency is adjusted so that the sideband frequency is kept constant and that by forming the difference between that of the voltage controlled oscillator generated mixing frequency and the reference frequency, the Doppler frequency is determined.

In a first advantageous variant, the relative phase shift of π / 2 between the two signal arms by means of a 90 ° phase shifter in one signal branch will be realized.

Another preferable way to compare the relative phase shift of π / 2 between The two signal branches are generated by means of a phase shifter in each of the Signal arms achieved by a certain proportion of the in each signal branch Phase shift is set. A shift around is best for this ± π / 4, since this is technically easiest to implement by using one Signal branch a high pass and a low pass is used in the second signal branch. Other options for generating a relative phase shift by π / 2 are through various measures of a delay in transit time by delay elements given.

Furthermore, the object according to the invention is achieved in an arrangement for evaluating Containing laser Doppler signals with a homodyne signal processing circuit a signal detector whose output signal contains the Doppler frequency, the one Signal detector downstream phase shifter for generating two signal branches of the delivered Doppler signal, the signal branches being mutually phase-shifted by 90 ° are and each have an electronic mixer, and a voltage controlled Oscillator for generating a mixing frequency which is supplied to the in the mixers, a summation element for summing the signals of both signal branches and Suppress one of two sidebands and a control loop to keep it constant the frequency of the sideband by means of the voltage controlled oscillator, thereby solved that the electronic mixer in the two signal branches with in phase Signals of the voltage-controlled oscillator are connected in at least one of the two signal branches a phase shifter downstream of the mixer for generation a relative phase shift of π / 2 between the two signal branches is arranged, the output signal of the summation element only one in two  sideband frequencies generated the mixers that the output of the Summation element in a control loop, consisting of a phase detector, a Reference oscillator as a basis for comparison for the phase detector, and the said voltage controlled oscillator, is integrated, the voltage controlled Oscillator can be controlled in such a way that the mixing signal supplied to the mixers Doppler frequency change is compensated and the output signal of the Summation amplifier has a constant frequency and that an evaluation unit to determine the Doppler frequency as the difference from the current of voltage controlled oscillator supplied mixed frequency and the reference frequency of the Reference oscillator is present.

It is useful to achieve the 90 ° phase shift between the two A π / 2 phase shifter is present in one of the signal branches.

However, it turns out to be particularly advantageous for the mixer in each of the signal branches a π / 4 phase shifter with different signs of the shift nachzuordnen. For this purpose, a low-pass filter and in another arranged a high pass filter. This particularly inexpensive solution Phase shifter is kept constant by the output frequency according to the invention of the summation element possible because the two at their resonance frequency operated filter realize exactly the required π / 4-phase shift.

Further possibilities of the phase shift arise in at least one the signal branches an electronic runtime chain or a delayed cable is arranged.

For frequency comparison, a phase detector is advantageously used in the control loop Low pass follows, used.

The control loop can be between the output of the summation amplifier and the voltage-controlled oscillator also expediently a conventional PLL module be used, however, the output of the internal voltage controlled Oscillator led to the mixer of the signal branches and the free input of the inherent phase detector can be connected with an external reference oscillator got to.

For the frequency comparison in the control loop there is preferably also a so-called Quadrant comparator with a subordinate integrator suitable, the  Reference oscillator provides an orthogonal reference frequency system that converted to square wave voltages, a sensitive interval switching system of the Quantum comparator represents when the sideband frequency at the output of the Summation element changes.

The basic idea of the invention is based on the fact that all frequency tracking modulators used for homodyne laser Doppler devices have the problem that either the bandpass suppression of one of the sidebands (by frequency admixture to the Doppler frequency) at low Doppler frequencies also suppresses the sideband to be evaluated or that the known constant control of an output signal from superimposed phase-shifted mixed signals causes considerable difficulties in maintaining the exact phase shift over large frequency ranges (± 3 MHz). Building on the latter principle, according to the invention, the Doppler signal sin (ω _D t) is used to generate two Doppler signals, sin (ω _D t) and cos (ω _D t), which are shifted relative to one another by 90 °, and which are carried separately in the two signal branches become. In these signal branches there are mixers which each add a (in-phase) signal (sin (ω _VCO t)) supplied by a voltage-controlled oscillator (VCO) to the two (phase-shifted) Doppler signals.

The following signals are generated in the two signal branches:

a: sin (ω _D t) sin (ω _VCO t) = 0.5cos [(ω _VCO - ω _D ) t] - 0.5cos [(ω _VCO + ω _D ) t] (2)

b: cos (ω _D t) sin (ω _VCO t) = 0.5sin [(ω _VCO - ω _D ) t] + 0.5sin [(ω _VCO + ω _D ) t] (3)

The signals are produced by a subsequent relative phase shift of π / 2 between the signals of both signal arms

a: cos [(ω _VCO - ω _D ) t] - cos [(ω _VCO + ω _D ) t] (4)

b: cos [(ω _VCO - ω _D ) t] + cos [(ω _VCO + ω _D ) t], (5)

which are subsequently superimposed so that exactly one sideband (cos [(ω _VCO + ω _D ) t] is completely suppressed (canceled) if the summation sign is selected accordingly). The evaluable signal then only consists of a sideband (here: cos [(ω _VCO - ω _D ) t]).

By comparison with a reference signal cos (ω _OS t) supplied by a frequency-stable reference oscillator, the VCO is readjusted in the event of differences, so that the evaluable sideband frequency always remains constant ω _VCO ω _D = ω _OS , so that a very narrow-band bandpass filter that transmits the signal / Noise ratio improved, can be used. On the other hand, the readjustment of the VCO also keeps the frequency in the signal branches after the two mixers constant, so that the phase shifter (s) only have to work in a very narrow band. As a further advantage of the invention, the evaluation of the Doppler frequency (due to the frequency being kept constant by means of the control loop) is limited to the detection of changes in the output frequency of the VCO.

With the invention it is possible for the first time to overcome the difficulties of homodyne laser Doppler Devices, each consisting of the incomplete or simultaneous suppression of the at Frequency lag demodulation resulting sidebands of the Doppler frequency result in overcoming. The invention allows simple means with any Magnitude of the Doppler frequency a complete suppression of one of the Sidebands arising from frequency mixing. Furthermore, the result in a closed control loop frequency shift kept constant Single sideband suppression reduces the demands on the electronic Components.

The invention will be explained in more detail below on the basis of exemplary embodiments become. The drawings show:

Fig. 1 shows an inventive principle scheme,

Fig. 2 shows a variant with a π / 2 phase shifter in a signal path,

Fig. 3 shows an embodiment with ± π / 4 phase shifters in the form of high and low pass filters.

In its basic variant, the method according to the invention comprises - as can be seen in FIG. 1 - the following processing steps:

• - from the Doppler signal sin (ω _D t) of the detector 1 which is preferably produced by the homodyne cross-beam method are generated via a phase shifter 2, two 90 ° phase-shifted Doppler signals into two signal branches 31 and 32,
• in the signal branches 31 and 32 the phase-shifted Doppler signals sin (ω _D t) and cos (ω _D t) are mixed with in-phase signals of the mixed frequency f _M = sin (ω _VCO t) of a voltage-controlled oscillator (VCO) 84 ,
• - After this mixture, a relative phase shift of π / 2 between the two signal branches 31 and 32 is generated and additively superimposed in a summation element 6 for single-sideband suppression, so that only one sideband frequency f _E = cos [(ω _VCO - ω _D ) t] or f _E = cos [(ω _VCO + ω _D ) t] remains,
• - After the summation, the sideband frequency f _{E is} applied to the mixer 41 via a control loop 8 ; 42 , whereby the sideband frequency f _{E is} compared with a reference frequency f _{O of} a reference oscillator 82 and, in the event of deviations between these two frequencies, the mixed frequency f _M supplied by the VCO 84 is adjusted so that the sideband frequency f _{E is} kept constant, and
• - The difference between the mixed frequency f _M = sin (ω _VCO t) generated by the VCO 84 and the reference frequency f _O = sin (ω _OS t) is then used to determine the Doppler frequency f _D = f _O - f _M.

The mixed frequency f _M = sin (ω _VCO t) added to the phase-shifted Doppler signals sin (ω _D t) and cos (ω _D t) is subject to the condition ω _VCO <ω _D , so that two sidebands (ω _VCO + ω _D ), (ω _VCO - ω _D ) arise around the VCO frequency ω _VCO . A closed loop 8 keeps the frequency f _E of the side band selected for evaluation, in this example cos (ω _VCO - ω _D ), constant at the reference frequency f _O.

As a result of the readjustment of the mixing frequency f _M depending on the change in the Doppler signals sin (ω _D t) and cos (ω _D t), the frequency fluctuations after the mixers 41 and 42 are very small, the relative 90 ° phase shift between the two Signal branches 31 and 32 can only be realized in a relatively small frequency range around f _O = f _M ± f _D. This relative phase shift can be realized with simple technical means, ie with phase shifters of almost any frequency response. It can be carried out both by a time delay shift in only one signal branch 31 ( FIG. 2) and particularly advantageously by a different ± π / 4 shift (according to FIG. 3) for each signal branch 31 or 32 .

The phase shift of the Doppler signal sin (ω _D t) for generating the signal branches 31 and 32 can also be implemented more easily with a ± π / 4 shift.

The arrangement according to the invention consists in its basic structure - as can also be seen in the illustration in FIG. 1 - from a phase shifter 2 arranged downstream of the signal detector 1 , which generates two Doppler signals phase-shifted by 90 ° in separate signal branches 31 and 32 , one electronic mixer 41 and one respectively 42 in the signal branches 31 and 32 , each of which is supplied with an in-phase mixed frequency f _M from a voltage-controlled oscillator (VCO) 84 , a phase shift unit 5 , which generates a phase difference of π / 2 between the two signal branches 31 and 32 , a summation element 6 for additive superimposition of the signals from both signal branches 31 and 32 , a bandpass filter 7 and a control loop 8 , which keeps a sideband frequency f _E selected for evaluation at the output of the summation element 6 constant by changing the Doppler signal by readjusting the mixed frequency f provided by the VCO 84 _{M is} compensated. The current Doppler frequency is calculated in an evaluation unit that contains at least one difference generator 9 by linking it to the reference frequency f _{O of} a frequency-stable reference oscillator 82 . It is therefore equivalent to the change in the mixed frequency f _M.

The relative phase shift in the two signal branches 31 and 32 after the frequency mixers 41 and 42 is provided by a phase shift unit 5 in such a way that exactly a path difference of π / 2 is set between the signals. This path difference between the Doppler-modulated mixed signals at the end of the signal branches 31 and 32 leads to the extinction of a sideband with a corresponding sign of the summation in the summation element 6 . The remaining sideband, here f _E = cos [(ω _VCO - ω _D ) t], is filtered out via the bandpass filter 7 in a narrow band. In contrast to previous homodyne LDA evaluation systems, the bandpass filter 7 plays a very subordinate role here, since the single-sideband suppression takes place during the additive superposition in the summation element 6 and the bandpass filtering only improves the signal / noise ratio.

The sideband signal f _E = cos [(ω _VCO - ω _D ) t] selected for evaluation, which is present at the output of the summation element 6 connected to the bandpass filter 7 , is initially used in the following control loop 8 to readjust the mixed frequency f _M depending on changes in the Doppler signal.

The control loop 8 contains a phase detector 81 which compares the filtered out sideband frequency f _{E of} the summation element 6 with the reference frequency f _{O of} the reference oscillator 82 . In the event of (phase or frequency) deviations, the phase detector 81 emits a signal which adjusts the VCO 84 to such an extent that the sideband frequency f _E corresponds to the reference frequency f _{O of} the reference oscillator 82 . In this example, the phase detector 81 carries out an analog multiplication of the two input signals, which results in a signal with the sum and difference frequencies of the frequency deviation, from which the sum frequency is suppressed by a downstream low-pass filter 83 . The remaining differential frequency is then amplified and controls the VCO 84 such that the deviation caused by the Doppler signals sin (ω _D t) and cos (ω _D t) in the signal branches 31 and 32 is compensated and the sideband frequency f _E after the summation element 6 again corresponds to the reference frequency f _{O of} the reference oscillator 82 . As a result of this readjustment, both the output signal of the summation element 6 and the signals in the two signal branches 31 and 32 after the mixers 41 and 42 are kept constant, so that the bandpass filter 7 and the phase shifting unit 5 have only low requirements with regard to edge steepness or frequency response are, as the following examples clearly show.

In FIG. 2, an embodiment of the invention is specified, the relative phase shift between the signal paths 31 and 32 is made only in the signal branch 31 with the signal after the admixture in the mixers 41 and 42. The phase shifter 51 used here delays the signal by π / 2. Since, due to the constant readjustment of the mixing frequency f _M as a function of the change in the Doppler signals sin (ω _D t) and cos (ω _D t), the frequency fluctuations after the mixers 41 and 42 are very small, the phase shifter 51 can be a delay element with almost any desired number Frequency response. It can simply contain a delayed cable or an electronic delay chain.

The phase-shifted signals generated in this way are introduced into a summation amplifier 61 and lead to the cancellation of the sideband (ω _VCO + ω _D ). The remaining sideband frequency f _E = [cos (ω _VCO - ω _D ) t] is processed in the control loop 8 in the same way as already described for FIG. 1. Mixing according to the invention without prior shifting the frequency of the mixed signal can simplify the generation of the mixed signal itself, so that the control loop 8 , as shown in FIG. 2, can be built up with a PLL module 84 (phase locked loop), but with the output of it existing VCO is separated from the phase detector input and passed to the mixers 41 and 42 and the free phase detector input is connected to an additional external oscillator, the reference oscillator 82 .

Fig. 3 shows a particularly simple and inexpensive variant of the invention. In this embodiment, the phase shift after mixers 41 and 42 is carried out in both signal branches 31 and 32 . Since the frequency of the side band f _E to be detected is kept constant by means of the closed control loop 8 , the relative phase shift by 90 ° in the signal branches need only be realized in a very narrow band. Therefore, the mixer 41 in one signal branch 31 can be followed by a low-pass filter 52 and the mixer 42 in the other signal branch 32 can be followed by a high-pass filter 53 . Low-pass filter 52 and high-pass filter 53 are operated at their resonance frequency and deliver a shift by -π / 4 in the first signal branch 31 and a phase shift by + π / 4 in the other signal branch 32 . The low-pass and high-pass filters 52 and 53 can therefore be simple RC and CR elements. But there are also high-quality high and low passes, such. B. CL links can be used.

For the frequency comparison in the control loop 8 Fig. 3 shows a further advantageous embodiment by a Quadrantenkomparator 86 is used. This is based on the fact that it compares the sideband frequency f _E with an orthogonal frequency system, that is to say that the reference oscillator 82 provides signals which are phase-shifted by 90 ° and which, converted into rectangular functions and superimposed additively, result in state intervals in which only the change in the state interval occurs directional up or down signal (also: u / d = up / down) at the output of the quadrant comparator 86 . The u / d signals are then converted in an integrator into a control voltage for the VCO 84 (this frequency comparison system has already been described in detail in German patent DE 197 42 608, which is referred to here).

The technically very undemanding design of the phase shifting unit 5 in the form of delay chains, delay lines and finally simple CR and RC filters brings the advantages of the invention to bear, and nevertheless guarantees highly effective single-sideband suppression. This is only possible because the in-phase frequency admixture of the mixed frequency f _M always compensates for the changes in the Doppler signals and thus leads to constant frequency ratios in the two signal branches 31 and 32 for the (otherwise susceptible) relative 90 ° phase shift of the Doppler-modulated mixed signals, so that the additive single sideband suppression can be done completely by additive superposition.

List of the reference symbols used

1

signal detector

2

phase shifter

31

.

32

signal branches

41

.

42

mixer

5

Phase shifter

51

phase shifter

52

Low Pass Filter

53

High Pass Filter

6

Summation member

61

Summing amplifier

7

Bandpass filter

8th

control loop

81

phase detector

82

reference oscillator

83

lowpass

84

VCO

85

PLL block

86

quadrant detector

87

integrator

9

differentiator
f _D

Doppler frequency
f _M

mixing frequency
f _E

Sideband frequency
f _O

reference frequency

Claims (14)

1. Method for evaluating laser Doppler signals, in which, using a homodyne laser Doppler arrangement, two phase-shifted Doppler signals are generated from the detector signal by phase shifting and mixed in separate signal branches with the signal of a voltage-controlled oscillator, whereby subsequent summation of the signal branches suppressed sideband of the Doppler-modulated composite signal and the remaining sideband is fed back in a control loop to the voltage controlled oscillator so that the frequency of the remaining sideband is kept constant and equivalent to the Doppler frequency magnitude of the control loop is available for evaluation, characterized characterized that
after the generation of the two Doppler signals phase-shifted by 90 ° in two signal branches ( 31 ; 32 ), the Doppler signals are mixed with in-phase signals of the mixed frequency (f _M ) of the voltage-controlled oscillator ( 84 ),
after this mixing, a relative phase shift of 90 ° is generated between the two signal branches ( 31 ; 32 ) before both signal branches are additively superimposed to extract a sideband frequency (f _E ),
the sideband frequency (f _E ) is fed back to the mixer ( 41 ; 42 ) in a kind of PLL control loop, the sideband frequency being compared with a reference frequency of a reference oscillator ( 82 ) and, in the event of deviations, the voltage-controlled oscillator ( 84 ) supplied by the (f _O ) Mixing frequency (f _M ) is tuned so that the sideband frequency (f _E ) is kept constant, and
the Doppler frequency (f _D ) is determined by forming the difference between the mixed frequency (f _M ) generated by the voltage-controlled oscillator ( 84 ) and the reference frequency (f _O ). 2. The method according to claim 1, characterized in that the relative phase shift of 90 ° between the two signal branches ( 31 ; 32 ) is realized by 90 ° phase shift in a signal branch. 3. The method according to claim 1, characterized in that the relative phase shift of 90 ° between the two signal branches ( 31 ; 32 ) is realized by means of a phase shifter in each of the signal branches. 4. The method according to claim 3, characterized in that the relative phase shift of 90 ° between the two signal branches ( 31 ; 32 ) by means of -π / 4-phase shift in one signal branch ( 31 ) and + π / 4-phase shift in the second signal branch ( 32 ) is realized. 5. The method according to claim 2 or 3, characterized in that the relative phase shift between the two signal branches ( 31 ; 32 ) is generated by means of a transit time shift. 6.An arrangement for evaluating laser Doppler signals with a homodyne signal processing circuit, comprising a signal detector whose output signal contains the Doppler frequency, a phase shifter downstream of the signal detector for generating two signal branches of the supplied Doppler signal, the signal branches being phase-shifted from one another by 90 ° and each have an electronic mixer, and a voltage-controlled oscillator ( 84 ) for generating a mixing frequency which is supplied to the mixers, a summing element ( 6 ) for summing the signals of both signal branches and suppressing one of two sidebands and a control loop for keeping the frequency of the Sideband by means of the voltage controlled oscillator, characterized in that
the electronic mixers ( 41 ; 42 ) in the two signal branches ( 31 ; 32 ) are connected to in-phase signals from the voltage-controlled oscillator ( 84 ),
In at least one of the two signal branches ( 31 ; 32 ), a phase shifter ( 51 ) arranged downstream of the mixer ( 41 ; 42 ) for generating a relative phase shift of π / 2 is arranged between the two signal branches ( 31 ; 32 ), the output signal of the summation element ( 6 ) has only one of two sideband frequencies (f _E ) generated in the mixers ( 41 ; 42 ),
the output of the summation element ( 6 ) is integrated in a control loop ( 8 ), consisting of a phase detector ( 81 ), a reference oscillator ( 82 ) as a basis for comparison for the phase detector ( 81 ), and said voltage-controlled oscillator ( 84 ), the voltage-controlled oscillator ( 84 ) can be controlled in such a way that the Doppler frequency change is compensated for by means of the mixed signal (f _M ) supplied to the mixers ( 41 ; 42 ) and the output signal of the summation amplifier has a constant frequency and
there is an evaluation unit ( 9 ) for extracting the Doppler frequency (f _D ) as the difference between the mixed frequency (f _M ) currently supplied by the voltage-controlled oscillator ( 84 ) and the reference frequency (f _O ) of the reference oscillator ( 82 ). 7. Arrangement according to claim 6, characterized in that in one of the signal branches ( 31 ; 32 ) there is a π / 2 phase shifter ( 51 ). 8. Arrangement according to claim 6, characterized in that in each of the signal branches there is a π / 4 phase shifter ( 52 ; 53 ) with different signs of the shift. 9. Arrangement according to claim 8, characterized in that a low-pass filter ( 52 ) is arranged in a signal branch ( 31 ) and a high-pass filter ( 53 ) is arranged in the other signal branch ( 32 ). 10. The arrangement according to claim 6, characterized in that an electronic runtime chain is arranged in at least one of the signal branches ( 31 ; 32 ). 11. The arrangement according to claim 6, characterized in that for delaying the π / 2 phase in at least one of the signal branches ( 31 ; 32 ) a delayed cable is arranged. 12. The arrangement according to claim 6, characterized in that in the control loop ( 8 ) between the output of the summation amplifier ( 6 ) and the mixers ( 41 ; 42 ), a PLL module ( 85 ) is used, the output of the internal voltage-controlled oscillator ( 84 ), however, led to the mixer ( 41 ; 42 ) of the signal branches ( 31 ; 32 ) and the free input of the phase detector ( 81 ) is connected to the reference oscillator ( 82 ). 13. The arrangement according to claim 6, characterized in that a phase detector ( 81 ), which is followed by a low-pass filter ( 83 ), is used in the control loop ( 8 ) for frequency comparison. 14. Arrangement according to claim 6, characterized in that a quadrant detector ( 86 ), which is followed by an integrator ( 87 ), is used in the control loop ( 8 ) for frequency comparison, the reference oscillator ( 82 ) providing an orthogonal frequency system.