DD292794A5 - ARRANGEMENT FOR PROCESSING A WIDE-TELEVISION SIGNAL FOR EQUIVALENT IMAGE AND INTERSECTION IMAGING - Google Patents

Abstract

The invention relates to an arrangement for processing television signals, which are the representation of a widescreen consisting of edge part and main part serve. The original widescreen signal is decomposed into a plurality of main components which are subjected to intraframe processing. Thereafter, they are combined again in a single signal transmission channel. Fig. 1 {television signal processing; Widescreen; Edge part; Bulk; Bildreihenverhaeltnis; Picture quality; Signal time compression; Intraframe signal processing; Signal components; Signaluebertragungskanal}

Description

For this 27 pages drawings

Field of application of the invention

The present invention relates to an arrangement for processing a television signal to display a wide picture having edge portion picture information and body picture information and an aspect ratio larger than that of a standard television picture.

Characteristic of the known state of the art

A conventional television receiver, such as a receiver tuned to r <! D in the USA and other common NTSC broadcasting standard, has an aspect ratio (the ratio of the width to the height of the reproduced picture) of 4: 3. Recently, however, interest in higher aspect ratios for television receiver systems, for example, ratios of 2: 1.16: 9 or 5: 3, since such higher aspect ratios are closer to or equal to the visual field of the human eye than is the case in FIG 3-aspect ratio of a conventional television receiver is the case. The video information signals with an aspect ratio of 5: 3 have received special attention because this ratio approaches that of a motion picture film, and such signals can then be transmitted and received without curtailing the picture information. However, widescreen television systems that simply send signals with a higher aspect ratio compared to conventional systems are incompatible with receivers that have the conventional aspect ratio. This makes a widespread introduction of widescreen systems difficult. It is therefore desirable to have a widescreen system compatible with conventional television receivers. Such a system is disclosed in U.S. Patent 4,782,383 to MA Isnardi entitled "Apparatus for processing high frequency edge information in a widescreen television system" published on November 1, 1988. There is an even greater desire for In such a widescreen compatible system, measures are taken to improve or enhance the resolution of the reproduced image in order to obtain additional image details. For example, a wide definition extended definition television (EDTV) system may include a progressive (progressive) device A system of this type is disclosed in a paper by MA Isnardi et al., entitled "Encoding for compatibility and recovery in the ACTV system", and in IEEE Transactions on Broadcasting, Vol.CBC-33, No.4 , December 1987, pages 116-123. The system uses signal time compression and intraframe processing of the image information of the middle and boundary portions.

Object of the invention

The aim of the invention is a comprehensive application of the widescreen system, wherein a reception is possible with conventional systems. This means that the widescreen signals are reproduced in both standard NTSC receivers and wide picture quality widescreen receivers.

Explanation of the essence of the invention

It is an object of the present invention to provide an arrangement for processing a television signal for displaying a wide picture having edge portion picture information and body picture information and an aspect ratio larger than that of a standard television picture, wherein signal time compression and intraframe signal processing are performed becomes.

In accordance with the principles of the present invention, an arrangement for improving systems of the type described is disclosed herein to achieve increased uniformity of spatial resolution between the reproduced center and edge portion information by substantially reducing or eliminating unwanted diagonal image artifacts in the rendered edge portion information ,

The arrangement in accordance with the principles of the present invention is disclosed in the context of a compatible widescreen EDTV television system employing time compression and intraframe signal processing, such as the averaging technique. The wide-screen EDTV signal includes a plurality of components, including a first major component containing the mid-portion and time-compressed edge portion information, and an additional second component containing the margin portion information. In the main component, only the sub-part information is subjected to intra-frame processing. The portion of currently compressed edge portion information of the main component is not subjected to intraframe processing.

In a disclosed preferred embodiment of a compatible widescreen EDTV television system j - \ accordance with the principles of the present invention, an original widescreen progressive scan signal of high resolution is coded so that four components are incorporated. The four components are processed separately before being reunited in a single signal transmission channel.

A first component is a 2: 1 interlaced main signal with a standard aspect ratio of 4: 3. This component consists of a middle portion of the wide-screen signal, which is time stretched to occupy nearly the entire active line duration of the 4: 3 aspect ratio, and of horizontal low-frequency edge portion information, which is temporally compressed in the left and right horizontal frame overscan portions. such information being hidden from view in a standard television receiver display. Only the middle part of this component is subjected to an intraframe message above a given frequency.

A second component is an additional signal in the 2: 1 interlaced format, which comprises the high-frequency information of the left and right edge portions, which are each temporally stretched up to half of the active line duration. In this way, the stretched edge part information occupies substantially the entire active line duration. This component is "mapped" to occupy the same period of time as the middle part of the first component, and it is assigned to an infra-frame.

Message.

A third component is a 2: 1 interlaced additional signal derived from the wide-screen signal source comprising the high frequency horizontal luminance detail information between approximately 5.0MHz and 6.0MHz.

This component is also mapped to occupy the same time period as the middle part of the first component, and it is subjected to an intraframe message. The intraframe-averaged second and third components undergo quadrature modulation of a phased alternating subcarrier which is combined with the intraframe-averaged first component.

An arbitrary fourth component is a 2: 1 interlaced additional "helper" signal that includes a temporal field difference of luminance detail information to help recover the missing image information in a widescreen EDTV receiver.

In a widescreen EDTV receiver, a composite signal consisting of the described four components is decoded into the four toilets. The decoded components are separately processed and used to develop a wide-band signal representing an image with increased resolution.

embodiments

Figure 1 provides a general overview of a compatible widescreen EDTV coding system including an arrangement according to the present invention:

Figure 1 a shows a detailed block diagram of the coder for the disclosed system; Figures 1 b through 1 e contain useful diagrams for understanding the operation of the disclosed system; Figures 2 to 5 show signal waveforms and diagrams useful in understanding the operation of the disclosed system;

Figure 13 is a block diagram of a portion of the decoding apparatus of the widescreen EDTV receiver, and Figures 6 through 12 and 14 through 24 illustrate aspects of the disclosed system in detail.

A system designed for that. Images with a large aspect ratio of, for example, 5: 3 over a standard, e.g. B. NTSC broadcast channel, should on the one hand achieve high quality image reproduction on a widescreen receiver and on the other hand greatly reduce visible noise or eliminate completely when the image is reproduced with a standard aspect ratio of 4: 3. The application of the signal compression technique to the fringes of an image has the advantage of the horizontal oversampling range of a standard NTSC television receiver; however, this can be at the expense of image resolution in the edge portions of a reconstructed wide-screen image. Since compression in time results in rotation at the level of the level, when processed in a standard television channel, only low-frequency components will be left which have a smaller bandwidth compared to that required for a wide-screen signal. Therefore, when the compressed edge portions of a compatible wide-screen signal are stretched in a wide-screen receiver, there is a noticeable difference between the resolution or high-frequency portion of the center portion of a reproduced wide-screen image and the edge portions, unless steps are taken to avoid this effect. This significant difference stems from the fact that low-frequency edge information is recovered while high-frequency information is lost due to the band-limiting effects of the video channel.

In the arrangement of Figure 1, the elements which correspond to those of the detailed arrangement of Figure 1 a, indicated by the same reference numerals. As shown in Fig. 1, an original wide-screen progressive scan signal having left, right and center part information is processed to develop four separate encoding components. These four components have been described above and are shown pictorially in FIG. The processing of the first component (including the time-expanded center-part information and the time-compressed low-frequency edge portion information) is performed such that in this example the resulting luminance bandwidth does not exceed the 4.2 MHz NTSC luminance bandwidth. This signal is color coded in standard NTSC format, and the luminance and chrominance components of this signal are appropriately prefiltered (eg, using field comb filters) to provide improved luminance / chrominance separation in both the standard NTSC format and NTSC format also in the widescreen receivers to ensure. The temporal extension of the second component (high-frequency information of the edge portions) reduces its horizontal bandwidth to about 1.16 MHz. This component is spatially uncorrelated with the main signal (the first component), and thus special security measures are taken to prevent its visibility on the standard NTSC receivers, as will be discussed.

The extended high frequency component of third component luminance information, ranging from 5.0 to 6.0 MHz, is first frequency shifted down to a frequency range of 0 to 1.0 MHz before further processing. The fourth component (temporal field difference helper signal) is mapped into the 4: 3 standard format to correlate with the main signal component, thereby obscuring its visibility on standard NTSC receivers, and is horizontal in bandwidth to 75OkHz limited.

As will be explained in more detail below, the first, second and third components are processed by respective Infraframe message devices 38, 64 and 76 (a type of vertical-temporal (VT) filter) to control the vertical-temporal crosstalk between the main and to eliminate additional signal components on a widescreen receiver. The sub-part information of the first component is subjected to an intra-frame message above approximately 1.5 MHz. The second and third intraframe-averaged components, labeled X and Z, undergo nonlinear amplitude compression prior to quadrature modulation of an alternating 3.108 MHz subcarrier ASC with a periodically (inverted) field phase in a block 80. In an adder stage 40, a modulated signal (M) from block 80 is added to the intraframe-averaged first component (N). A resulting output is a 4.2 MHz bandwidth baseband signal (NTSCF) which, together with a 75OkHz low pass filtered fourth component (VTN) from a filter 79, quadrature modulates an RF picture carrier in a block 57 to form an NTSC compliant RF Signal to be transmitted to a standard NTSC or progressive scan widescreen receiver over a single standard bandwidth broadcast channel.

Applying the time compression to the first component allows the low-frequency edge information to be completely injected into the horizontal oversampling area of a standard NTSC signal. The high-frequency edge part information of the second component and the high-frequency luminance detail information of the third component are spectrally divided with the standard NTSC signal through the video transmission channel in a manner transparent to a standard receiver by applying a quadrature modulation method to an alternating one Subcarrier in the block PO, as will be explained. When received with a standard NTSC receiver, only the middle portion of the main signal (the first component) is visible. The second and third components may cause a low amplitude interference pattern, but this is not noticeable at normal viewing distances and in normal settings of the image controls

 The fourth component is completely suppressed in receivers with synchronous video detectors. In receivers with

Envelope rectifiers process the fourth component, but they are not perceived because they are in the main signal

is correlated.

The main signal (component 1) has an active standard NTSC horizontal line interval of approximately 52 microseconds (μβ). Only the high frequency information of this component is subjected to an intraframe message above about 1.5MHz. The time-compressed low-frequency hand part information of this component is not subjected to the process of intra-frame message. It has been found that such selective intraframe processing of the main component improves the resolution of the diagonal edge portion image information by eliminating unwanted jagged diagonal artifacts which are sometimes attributed to "jags" which would otherwise be produced in a reconstructed image if the compressed edge portion information of the Main signal has been subjected to an intraframe message.

In this regard, it should be noted that the low frequency edge portion information of the main signal component has been temporally compressed with an edge compression factor (SCF) of approximately six. If such temporally compressed information is subjected to an intraframe message before being time stretched in the image retrieval receiver, the retrieved edge portion image information would have jagged diagonals because the horizontal frequency at which the intraframe message was started would then be jagged approximately SCF times lower than for the middle part. The diagonal image information is increasingly disturbed ("jagged") as the frequencies at which the intraframe message is being communicated fall, for example, if the main signal is subjected to intraframe signaling at frequencies above 1.5 MHz and the low frequency edge component information of component 1 is temporally compressed with a SCF factor of six, the intraframe message of edge information effectively starts at a much lower frequency of 25OkHz (1.5MHz / SCF), resulting in jagged diagonals, so jagged diagonals in the recovered edge portions would become more noticeable Since the component 1 does not undergo intraframe notification in the temporally compressed edge portions, the entire range of the original frequency in these ranges (0 to 700 KHz) retains the full vertical resolution without jagged diagonal artifact distortion.

Component 2, which includes the left and right high-frequency edge portion information, is mapped to occupy the same time period as the middle portion of component 1. In this way, the left and right edge sub-levels are time-stretched to execute the entire center sub-area, whereby the component 2 has an active horizontal scanning interval of approximately 50ps, which corresponds to the horizontal scanning interval of the center portion of the component 1. For this purpose, the margin expansion factor (SEF) is about 4.32, compared to an edge expansion factor of about 4.49, which would be required to stretch the left and right edge portion information of component 2 to the full active row duration of 52 μβ.

Both components 2 and 3 are mapped into the middle portion because of the infra-frame processing performed on component 1 and accessory components 2 and 3. As will be explained below, the intraframe message is a process that facilitates the separation of the two previously combined signal components, in this case, for example, the main signal N and the modulated additional signal M. Since the range of intraframe processing in component 1 has been reduced, to include only the mid-portion of 50 μβ, the mapping of modulation components 2 and 3 is similarly modified to include only the mid-portion. As mentioned above, component 3 is mapped to coincide with the mid-portion interval by linear temporal compression of the extended horizontal luminance information to 50μβ. The time compression of component 3 from 52 ps to 50μβ dispenses with a portion of the spatial correlation with major component 1, but it can be determined with certainty that the center and edge portions of the reconstructed image have a similar horizontal resolution. Although the spatial correlation between the components 1 and 3 is desirable, around the alternating subcarriers already contains uncorrelated information in the form of component 2. The measure of the spatial correlation given in component 3 is insignificant and is enhanced by the resulting similar horizontal resolution of the center and edge portions. The component 4 is not intraframe-averaged and remains unchanged by having the full active line duration of 52 με, coincident with the main signal. In the decoder described in conjunction with Fig. 13, the intraframe processing is performed only with respect to the center portion for separating the signals M and N. After component M is demodulated into subcomponents 2 and 3, components 2 and 3 are mapped to their original time slots, ie, to assume a full active line interval of 52 ps.

Figure 1 b illustrates the RF spectrum of the disclosed EDTV widescreen system including the additional information compared to the RF spectrum of a standard NTSC system. In the spectrum of the disclosed system, the edge sub-levels and the additional high-frequency horizontal luminance detail information extend approximately 1.16 MHz on each side of the alternate subcarrier frequency (ASC) of 3.108 MHz. The V-T helper signal information (component 4) extends 75 okHz on each side of the picture carrier frequency of the main signal.

A progressive scan widescreen receiver includes means for restoring the original progressive widescreen strobe signal. Compared with a standard NTSC signal, the reconstituted widescreen signal has left and right edge portions with a standard NTSC resolution and a center portion with a 4: 3 aspect ratio with higher horizontal and vertical detail of detail, especially in fixed areas of an image. Two fundamental considerations dominate the signal processing technique associated with the development and processing of the first, second, third and fourth signal components. These two considerations are compatibility with existing receivers and restorability in the receiver.

Full compatibility also means compatibility of the receiver and transmitter so that existing standard receivers can receive widescreen EDTV signals and produce standard reproduction without special adapters.

For example, compatibility in this sense requires that the image sampling format in the transmitter be substantially the same as the image sampling format in the receiver, or at least within the tolerance of the receiver-side image sampling format. Compatibility also means that special, nonstandard components are hidden physically or in terms of perceivable in the main signal when played back using a standard receiver. In order to achieve compatibility in the latter sense, the disclosed system uses the following methods to hide the additional components.

As already described above, the edge part depths in the normal horizontal oversampling area are one

Standard receiver physically hidden. The component 2, which contains little energy compared to the component of the edge part depths, and the component 3, which is normally a high-energy low-energy detail signal, are subjected to amplitude compression and quadrature modulation to an alternating subcarrier at 3.108 MHz, which is an interlaced scanning frequency (odd-numbered) Multiple of half the horizontal line frequency). The frequency. Phase and amplitude of the alternating subcarrier are chosen so that the visibility of the modulated alternating subcarrier signal is reduced as much as possible, for example by controlling the phase of the alternating subcarrier from field to field so that it changes every 180 ° from one field to the next, unlike the phase of the chrominance subcarrier from one field to the next. Although the components of the modulated alternating subcarrier lie entirely within the chrominance passband (2.0 to 4.2 MHz), the components of the modulated alternating subcarrier are noticeably hidden because they are reproduced as complementary field-flicker color fading from the human eye normal levels of chroma saturation is not perceived. In addition, nonlinear amplitude compression of the modulation components prior to amplitude modulation advantageously reduces instantaneous amplitude overshoots to an acceptably lower level. The component 3 is spatially correlated with respect to the information of the center region of the component 1, and it is spatially correlated with respect to the left and right information components of the component 1 spatially. This is done with the help of a Bildformatkcdierers, as will be explained.

Component 4, the helper signal, is also hidden as a result of the temporal expansion of the mid-span information to ensure adaptation to the 4: 3 standard format, thereby spatially correlating the fourth component with the main signal. Component 4 is suppressed in standard receivers with synchronous detectors while hidden in standard receivers with envelope detectors due to their spatial correlation with the main perception signal.

The recovery of components 1, 2 and 3 in a progressive scan widescreen receiver is made possible by the application of a process of intraframe processing in the transmitter and receiver. This process is carried out in the transmitter system of Figures 1 and 1 a by the associated elements 38,64 and 76 and in the receiver by associated elements, as will be explained. Intraframe averaging is a signal handling technique in which two visually correlated signals for mutual combining are pretreated so that they are later recognized quickly and accurately, for example, by means of a field memory device, such that they are free from vertical temporal crosstalk, too then, when it comes to image signals and movement is present in the images. The type of signal handling used for this purpose essentially involves the action of making two signals identical on a field basis, i. H. generate two samples with identical values one frame apart. Intraframe averaging is a suitable technique for achieving this goal, but other techniques may be used. The intra-frame averaging is basically a process of linear time-varying digital prefiltering and postfiltering to ensure the accurate recovery of the two visually correlated combined signals. Horizontal crosstalk is eliminated by guard bands between horizontal prefilters in the transmitter-side encoder and between horizontal post-filters in the receiver-side encoder.

Intraframe averaging is a form of paired pixel processing. The process of intraframe averaging in the time plane is illustrated generally in Figure 1c, where pairs of fields are made identical by dividing the average of the picture elements (A, B and C, D) by the 262 horizontal line periods (262H) lie, is formed. The mean replaces the original values in each pairwise group. Figure 1d illustrates the process of intraframe averaging in the context of the system of Figure 1. Beginning with components 2 and 3, pixel pairs (pixels) spaced apart by 262 H within a frame are averaged, and the average (e.g. B. X1, X3 and Z1, Z3) replaces the original pixel values. This vertical-time averaging occurs within a frame and does not exceed the frame boundaries. In the case of the component 1, the intraframe averaging is performed on the information of the center part above approximately 1.5 MHz so as not to affect the vertical detail information of low frequency. In the case of components 1 and 2, intraframe averaging is performed on a composite signal including the luminance (y) and chrominance (c) components throughout the chrominance band. The chrominance component of the composite signal survives intraframe averaging because the pixels spaced apart by 262 H are in phase with the color subcarrier. The phase of the new alternating subcarrier is controlled to be for 262 H apart pixels Therefore, if components 2 and 3 (after quadrature modulation) in unit 40 are added to component 1, the pixels separated by 262 H have the form (M + A). and (M - A), where M is a sampling pattern of the composite main signal above 1.5 MHz and A is a sampling pattern of the modulated additional signal.

Intraframe averaging substantially eliminates vertical-temporal crosstalk, even with motion in the picture. In this regard, the process of intraframe averaging produces idontic sampling patterns at intervals of 262H. In the receiver, it is a simple matter to accurately recover the information content of these sampling patterns by processing the 262H apart pixel scans within a frame such as will be explained; thereby the information of the main and auxiliary signal is recovered. In a decoder in the receiver, the original information which has undergone intraframe averaging can be recovered almost intact via the process of intraframe processing, since the well-correlated original information has been made substantially identical from field to field.

Also in the receiver is d *. RF channel quadrature demodulated using an RF synchronous detector. The component 4 is thereby separated from the other three components. The intraframe processing is used to separate the component 1 from the modulated components 2 and 3, and using the quadrature demodulation, the components 2 and 3 are separated from each other, as will be discussed with reference to FIG. After the four components have been recovered, the composite signals are decoded according to the NTSC standard and separated into the luminance and chrominance components. An inverse mapping of all components is made to restore the widescreen aspect ratio, and the high frequency portions of the edge portion information are combined with the low frequency portions to recover the complete resolution in the edge portions. The enhanced high-frequency luminance detail information is shifted to its original frequency range and added to the luminance signal, which is then converted to the progressive-scan format using temporal interpolation and the helper signal.

The chrominance signal is converted to the progressive scan format using temporal interpolation performed without additional assistance. Finally, the progressive scan format luminance and chrominance signals are converted to the analog form and matrices to produce the R, G, and B color image signals for display on a progressive scan widescreen display device.

Before the compatible widescreen coding system according to FIG. 1a is described in more detail, the signal waveforms A and B of FIG. 2 are considered. The signal A is a wide-screen signal corresponding to the aspect ratio 5: 3 which is to be converted to an NTSC standard compatible signal for a 4: 3 aspect ratio signaled by signal B. The wide-screen signal A includes a central portion designated primary image information occupying an interval TC and left and right marginal portions having secondary image information occupying the intervals TS. In this example, the left and right edge portions have substantially equal aspect ratios that are smaller than those of the dominant center portion that adjusts therebetween.

The wide picture signal A is converted into the NTSC signal B by completely compressing a certain edge part information into the horizontal oversampling areas with the time intervals TO. The standard NTSC signal has an active line interval TA (having a duration of 52.6 microseconds), oversampling intervals TO, a playback time interval TD containing the video information to be displayed, and a total horizontal line duration interval TH having a duration of 63.556 Includes microseconds. The intervals TA and TH are the same intervals for both the widescreen and the standard NTSC signals. It has been found that almost all consumer television receivers have an oversampling interval occupying at least 4% of the total active line time TA, i.e., at least 4% of the total active line time TA. H. 2% oversampling on the left and right margins. At an interlaced scan rate of 4 x fsc (where fsc is the frequency of the color subcarrier), each horizontal line interval contains 910 pixels (pixels), of which 754 form the active horizontal line image information to be rendered.

The widescreen EDTV system is shown in more detail in Figure 1a. Reference is now made to Figure 1a. A progressive scan 525 line and 60 field per second wide-area camera 10 provides a widescreen color signal with components R, G, and B, and in this example a 5: 3 broit image aspect ratio. An interlaced signal source could also be used except that a progressive scan source provides better results. A widescreen camera has a larger aspect ratio and a larger video bandwidth compared to a standard NTSC camera, where the video bandwidth of a widescreen camera is proportional to the product of its aspect ratio and the total number of lines per frame, among other factors. At the same scanning speed of the wide-screen camera, increasing the aspect ratio results in a corresponding increase in its video bandwidth as well as greater horizontal compression of the image information when the signal is reproduced on a standard 4: 3 aspect ratio television receiver. For these reasons, it is necessary to modify the wide screen signal if it is to be fully NTSC compatible. The color video signal processed by the coding system of Figure 1 contains both luminance and chrominance signal components. The luminance and chrominance signals contain both low and high frequency information, which in the following description will be referred to as "lows" and "highs". The progressive scan widescreen color video signals having a wide bandwidth and coming from the camera 10 are matrixed in a unit 12 to derive the luminance component Y and the color difference signal components I and Q from the color signals R, G and B. The progressive scan wideband signals are sampled at eight times the frequency (8 x fsc) of the chrominance subcarrier and individually converted from analog to digital (binary) form by separate analog to digital converters (A / D) in an A / D unit 14 before being filtered individually by means of separate vertical-temporal low-pass filters in a filter unit 16 to produce filtered signals YF, IF and QF. These signals each have the shape as indicated by the waveform A in FIG. The separate filters are linear, time-invariant 3χ3 filters of the type shown in Figure 10 and discussed below. These filters easily reduce the vertical-temporal resolution, particularly the diagonal vertical-temporal resolution, to undesirable interlaced scanning artifacts (e.g., flickering, jagged edges, and other fold-over effects) in the main signal (component 1 in Figure 1) after conversion of the progressive ones To prevent sampling in the interlace scan. The filters preserve almost the full vertical resolution in still parts of the image.

The center stretch factor (CEF) is a function of the difference between the width of an image rendered by a wide-screen receiver and the width of an image rendered by a standard receiver. The image width of a widescreen image with an aspect ratio of 5: 3 is 1.25 times larger than the image width of a standard reproduction with

an aspect ratio of 4: 3. This factor of 1.25 is a preliminary factor for the midsection stretch which must be adjusted to account for the oversampling range of a standard receiver and to account for intentional slight overlap of the area boundaries between the center and edge portions, as will be discussed These considerations prescribe a CEF of 1.19.

The progressive scan signals from the filter network 16 have a bandwidth of 0 to 14.32 MHz and are respectively converted into 2: 1 interlaced format signals by progressive scan interlaced (P / Z) converters 17a, 17b and 17 c, the details of which will be discussed in connection with FIGS. 22 and 23. The bandwidth of the output signals IF ', QF' and YF 'from the P / Z converters 17 a to 17 c amounts to a range of 0 to 7.16 MHz, since the horizontal scanning frequency for interlaced scanning signals half of the signals is progressive sampling. In which

Conversion process, the progressive scan signal is subsampled by making up half the available pixel

Scanning pattern is taken to generate the main signal in 2: 1 interlaced format.

Specifically, each progressive scan signal is converted to the 2: 1 interlaced format, by maintaining either the odd or even lines in each field, and reading the retained pels at a frequency of 4 x fsc (14.32 MHz), all subsequent digital processing of the interlaced scanning signals is performed at a frequency of 4 x fsc.

The network 17c also contains an error prediction network. An output signal of the network 17c, YF ', is in the

Interlaced sub-sampled luminance version of the prefiltered progressive scan component. Another output (luminance) signal of the network 17c, YT, contains the temporal information derived from the field difference information representing a temporal prediction or interpolation error between actual and predicted values of the luminance sampling patterns which are missing in the receiver The prediction is based on a time averaging of the amplitudes of the "before" and "after" picture elements that are available in the receiver, the signal YT, a luminance "helper" signal, contributes to this in effect restoring the progressive scan signal in the receiver and, in principle, accounting for an error that the receiver can expect to make with respect to non-stationary image signals, and facilitates the cancellation of such error in the receiver. In the still parts of an image the error is zero, and in the receiver the full recovery is done. It has been found that a chrominance helper signal is not practically necessary and that a luminance helper signal is sufficient for good results because the human eye is less sensitive to missing vertical or temporal chrominance details. FIG. 2 a illustrates the algorithm used for developing a helper signal YT. In the following, reference is made to FIG. 2 a. The picture elements A, X and B in the progressive scan signal occupy the same spatial position in an image. The black pixels, such as A and B, are transmitted as the main signal and are available in the receiver. A white picture element, for example X, is not transmitted and predicted by a frame averaging (A + B) / 2. That is, in the encoder, prediction is made for "missing" picture elements X by averaging the amplitudes of the "before" and "after" picture elements A and legs The predicted value (A + B) / 2 is subtracted from the actual value X. to generate a prediction error signal, corresponding to the helper signal, having an amplitude according to the expression X - (A + B) / 2 This term identifies the temporal field difference information except the information of the time frame averaging The helper signal is generated by means of a 750kHz The band limiting of the helper signal to 75OkHz is necessary to keep this signal from interfering with the next lower RF channel after this signal is modulated onto the RF picture carrier using an average of the sample patterns A and B, a similar prediction of the missing picture element X, and the prediction error is added to the prediction. That is, X is retrieved by adding the prediction error X - (A + B) / 2 at the time average (A + B) / 2. In this way, the helper signal facilitates the interlaced conversion to the progressive scan format.

The helper signal, which is advantageously generated by the disclosed temporal prediction algorithm, is a low energy content signal compared to a predictive signal generated by other algorithms, for example the row difference signal generation algorithm described by M. Tsinberg in an article " ENTSC Two Channel Compatible HDTV System ", published in IEEE Transactions on Consumer Electronics, Vol. CE-33, No.3, August 1987, pages 146 to 153. In idle areas of an image, the error energy is zero because the prediction A low-energy state manifests itself through static and virtually static images (such as a news bulletin with a reporter against a static background).

It has been found that the disclosed algorithm causes the least disturbing artifacts after image reconstruction in the receiver, and that the helper signal generated by this disclosed algorithm retains its usefulness after being band limited (filtered) to about 75 oKHz. The helper signal generated by the disclosed algorithm advantageously has zero energy at still image information, and as a result, a helper signal associated with a still image is not affected by filtering.

Even if the helper signal is not transmitted, the reconstruction of the widescreen is much better. In such a case, still parts of the image appear much sharper than in an NTSC standard image, however, moving parts become somewhat "softer" and may contain a "beating" artifact. Thus, a broadcasting station does not need to transmit the helper signal from the beginning, but may choose to improve the RF transmission at a later time. The disclosed temporal prediction system is useful for both progressive scanning and interlaced scanning at higher than the standard line frequencies, but it works best with a progressive scan source in which pixels A, X and B occupy the same spatial position in one picture, resulting in a perfect prediction for still pictures. However, the temporal prediction will be imperfect even in still areas of an image when the original widescreen comes from an interlaced scanning signal source. In such a case, the helper signal will have more energy and will introduce small artifacts in still parts of a reconstructed image. Although experiments have shown that the use of an interlaced signal source gives acceptable results with artifacts that are only noticeable upon closer inspection, a source of progressive scan signals will bring less artifacts and better results.

In the arrangement of Figure 1 a, the wide-format signals IF ', QF' and YF 'present in the interlaced format are filtered by the P / Z converters 17a to 17c respectively in an associated ™ vertical low-pass filter 19a, 19b and 19c, respectively Signal IF "with a bandwidth of 0 to 600KHz, a signal QF" with a bandwidth of 0 to 600KHz and a signal YF "with a bandwidth of 0 to 5MHz to produce These signals are then subjected to a process of image format encoding, each of these signals being brought into the 4: 3 format by means of an image format encoder including an edge center divider and processing unit 18. Briefly, the center of each widow line is time stretched and written into the The portion of the active line duration shown in FIG. 4 is represented with an aspect ratio of 4: 3. The time stretching effects a reduction of the bandwidth, so that the interlaced frequencies of the original B riding picture signal with dp; Bandwidth of the NTSC standard. The edge portions are split into horizontal frequency bands so that the component containing the high frequency components ("highs") of the color of the I and Q signals has a bandwidth of 83KHz to 600KHz (as shown for signal IH in Figure 7) ) and the component containing the "heights" of the luminance signal Y has a bandwidth of 70OKHz to 5.0MHz (as shown for signal YH in Fig. 6). The low frequency components ("lows") of the edge portions, i.e. the signals Y0, IO and Q0 developed in accordance with Figures 6 and 7, contain and become a DC component

mapped into the left and right horizontal oversampling areas of each line under temporal compression. The edge heights are processed separately. Details of this aspect ratio encoding process will be explained below.

For consideration of the details of coding described below, it will also be helpful to consider Figure 1e, which illustrates the process of coding components 1,2,3 and 4 in conjunction with the recovered center and margin information. The filtered interlaced signals IF ", QF" and YF "are processed by the edge part / center part separator and processing means 18 to produce three groups of output signals: YE, IE and QE; YO, IO and QO and YH, IH and QH) The first two sets of symbols (YE, IE, QE, and YO, IO, QO) are processed into a signal containing a full bandwidth midsection component and the luminance depths of the edge portions pressed into the horizontal oversampling areas Signal group (YH, IH, QH) is processed into a signal containing the edge part heights.

When these signals are combined, a NTSC-compatible widescreen signal with a 4: 3 aspect ratio is obtained. Details of the circuits forming the unit 18 are shown in Figures 6, 7 and 8 and will be described in connection with these Figures 6, 7 and 8.

The signals YE, IE and QE contain the complete center part information and have the same format as shown in FIG. 3 for the signal YE. The signal YE is derived from the signal YE "in short as follows: The wide-field signal YF" includes picture elements 1 to 754 which appear during the active line interval of the wide-screen signal and contain both edge-part and center-part information. The wideband component information (picture elements 75 to 680) is extracted as a center part luminance signal YC by means of a time demultiplexing method. The YC signal is time stretched by the center portion expansion factor 1.19 (i.e., 5.0 MHz: 4.2 MHz) to produce the NTSC compatible center sub-signal YE. The signal YE has an NTSC compatible bandwidth (0 to 4.2 MHz) due to the time expansion by a factor of 1.19. The signal YE occupies the picture display interval TD (Fig. 2) between the oversampling areas TO. The signals IE and QE are derived from the signals IF "and QF" and processed in a similar manner as the signal YE. The signals Y0, IO and Q0 contain the low-frequency parts ("depths") of the edge-part information, which are inserted in the left and right horizontal oversampling areas. The signals Y0, IO and Q0 have the same format as that for the signal Y0 is shown in Fig. 3. The signal Y0 is derived from the signal YF "in short as follows. The wide-field signal YF "contains left-hand edge information in the picture elements 1 to 84 and right-hand edge information in the picture elements 671 to 754. As will be explained, the signal YF" is low-pass filtered to produce a luminance depth signal having a bandwidth of 0 to 700KHz from which a left and a right edge part depth signal are then extracted by means of a time-demultiplexing method (signal YL 'in FIG. 3). The luminance depth signal YL 'is time-compressed to generate the edge portion depth signal YO having the compressed low-frequency information in the oversampling areas, that is, the picture elements 1 to 14 and 741 to 754. The compressed edge portion depth signal is proportional to the measure of Time compression increased bandwidth. The signals IO and QO are developed from the signals IF "and QF" and processed in the same way as the signal YO.

The signals YE, IE, OE and YO, IO and QO are represented by a border part / center part combining circuit 28, e.g. As a time division multiplexer, combined to produce signals YN, IN and QN with an NTSC-compatible bandwidth and a 4: 3 aspect ratio. These signals take the form of the signal YN shown in FIG. Combination circuit 28 also includes appropriate signal delays to equalize the durations of the signals to be combined. Such compensating signal delays are also found elsewhere in the system where signal transit times are required.

A modulator 30, a bandpass filter 32, a horizontal / vertical / temporal notch filter 34 and combining circuit 36 constitute an improved NTSC signal coder 31. The chrominance signal components IN and QN are modulated by the modulator 30 to a subcarrier SC of nominal NTSC color subcarrier frequency , 58MHz modulated to produce a modulated signal CN. The modulator 30 is of conventional design and will be described in conjunction with FIG. The modulated signal CN undergoes vertical (V) and temporal (T) dimension bandpass filtering in the two-dimensional (VT) filter 32 which removes crosstalk artifacts in the interlaced chrominance signal before applying that signal as a signal CP to a chrominance signal input of the combining circuit 36 becomes. The luminance signal YN undergoes band-stop filtering in horizontal (H), vertical (V), and temporal (T) dimensions by means of the three-dimensional H-V-T band notch filter 34 before being applied as a signal YP to a luminance input of the combining circuit 36. The filtering of the luminance signal YN and the chrominance color difference signals IN and QN is intended to ensure that the crosstalk between luminance and chrominance is substantially reduced after the subsequent NTSC coding. Multi-dimensional spatio-temporal filters such as the H-V-T filter 34 and the V-T filter 32 in Figure 1 a have a structure as shown in Figure 10 and described below.

The H-V-T band-stop filter 34 in Fig. 1a has the structure shown in Fig. 10b and removes the frequency components of upward-moving diagonals from the luminance signal YN. These frequency components are similar in appearance to components of the color subcarrier and are removed to produce a gap in the frequency spectrum where modulated chrominance information is inserted. The removal of the frequency components of upwardly moving diagonals from the luminance signal YN does not visibly degrade a reproduced image because it has been found that human Augo is virtually insensitive to such frequency components. Filter 34 has a cutoff frequency of approximately 1.5 MHz in order to obtain the vertical detail information in the l.uminanz not to be affected.

The V-T bandpass filter 32 reduces the chrominance band widths so that the modulated chrominance information of the edge portions can be inserted into the gap created by the filter 34 in the luminance spectrum, as described above. The filter 32 reduces the vertical and temporal resolution of the chrominance information, so that quiet and moving edges are easily blurred, but with little or no consequences, because human Augo is insensitive to such effects. The combiner circuit 36 provides at its output a mid-section / edge-subcircuit signal C / SL, which contains NTSC-compatible information to be reproduced, derived from the center portion of the wide-screen signal, and compressed edge sub-lows (both luminance and chrominance) derived from the Are derived in the left and true horizontal oversampling areas, by a viewer on an NTSC receiver

not to be seen. The compressed edge part depths in the oversampling area form part of the edge part information for widescreen reproduction. The other component, the margin heights, is developed by the processing unit 18, as described below. The edge part height signals YH (luminance heights), IH (I-heights) and QH (Q-heights) are shown in FIG. Figures 6, 7 and 8 show an arrangement for developing these signals, as will be described. The signals YH, IH and QH shown in FIG. 4 contain high-frequency information of the left edge part in the picture elements 1 to 84 and high-frequency information of the right edge part in the picture elements 671 to 754.

The mid-portion of the signal C / SL is processed in the intraframe averaging circuit 38 to produce a signal which is fed to the input of the adder 40. The intraframe-averaged signal N is substantially identical to the signal C / SL because of the highly visible correlation of the intraframe image information of the signal C / SL. The averaging circuit 38 averages the signal C / SL above approximately 1.5 MHz and helps to reduce or eliminate vertical time crosstalk between the main signal and the supplemental signals. The high pass frequency range of 1.5MHz and above, in which the intraframe averaging circuit 38 operates, has been chosen so that full intraframe averaging for information at 2 MHz and above takes place to prevent luminance vertical detail information from passing through the process intraframe averaging is affected. Horizontal crosstalk is eliminated by a 200 KHz guard band between a filter belonging to the intraframe averaging circuit 38 in the encoder 31 and a filter belonging to an intraframe processing unit in the decoder of Figure 13. FIG. 11b shows details of the intraframe averaging circuit 38 provided for the heights. FIGS. 11 and 13 will be explained below. The signals IH, QH and XH are brought into the NTSC format by means of an NTSC encoder 60, which is similar to the encoder 31. Specifically, the encoder 60 includes means of the type shown in Figure 9 and means for quadrature modulating the heights of the edge-portion chrominance information to the heights of the edge-portion luminance information at 3.58 MHz to produce a signal NTSCH comprising the Represents heights of the margin information in NTSC format. This signal is shown in FIG.

The use of multi-dimensional bandpass filtering in the NTSC encoders 31 and 60 advantageously allows the luminance and chrominance components to be separated from each other substantially free of crosstalk in the receiver when the receiver uses complementary multidimensional filtering to separate luminance and chrominance information , The use of complementary filters for the encoding and decoding of luminance and chrominance are referred to as cooperative processing; she is in detail by CH. Strolle in an article entitled "Cooperative Processing for Improved Chrominance / Luminance Separation", published in SMPTE Journal, Vol. 95, No. 8, August 1986, pages 782 to 789. Even standard receivers using conventional notch filters and row comb filters benefit from the use of such multi-dimensional prefiltering in the encoder by showing less crosstalk between chrominance and luminance.

The signal NTSCH is time-stretched in a unit 62 to produce a stretched fringe menu signal ESH having an active horizontal line interval of 50 ms, i. H. less than the default active NTSC line interval of approximately 52 ps. More specifically, as shown in Fig. 5, the stretching is performed by a "mapping" process in which the left edge subpixels 1 to 84 of the signal NTSCH are mapped to the picture element positions 15 to 377 of the signal ESH, ie, the heights of the left edge portion in FIG The signal NTSCH is stretched to occupy approximately half the line duration of the signal ESH, and the portion (pixels 671 to 754) of the signal NTSCH associated with the right edge portion is similarly processed The time expansion process reduces the horizontal bandwidth of the information of the signal ESH (compared to that of signal NTSCH) by a factor of 363/84.

The imaging process by which the temporal stretching occurs can be realized by means of the type shown in Figs. 12 to 12d and described below in connection with these figures. The signal ESH undergoes intraframe averaging in a network 64 of the type shown in Figure 11d to produce a signal X, as shown in Figure 5. This average signal X is substantially identical to the signal ESH because of the highly visible correlation of the image information of the signal ESH within one frame. The signal X is applied to a signal input of a quadrature modulator 80.

The signal YF 'is also filtered by a horizontal bandpass filter 70 having a bandwidth of 5 to 6.0 MHz. The output of the filter 70, containing horizontal luminance levels, is applied to an amplitude modulator 72 in which it modulates the amplitude of a 5 MHz carrier signal f _c . The modulator 72 has a low-pass filter on the output side with a cutoff frequency of approximately 1.0 MHz in order to obtain a signal in the band range 0 to 1.0 MHz at the output of the modulator 72. The upper ("folded over") sideband (5.0 to 6.0 MHz) of the modulation process is removed by the 1.0 MHz low pass filter, and effectively the horizontal luminance treble frequencies are in the range of 5.0 to 6.0 MHz through the

Amplitude modulation process and the subsequent low-pass filtering in the frequency range 0 to 1.0MHz shifted. Dio carrier amplitude should be high enough to preserve the original signal amplitudes after filtering by the 1.0 MHz low pass filter. That is, there is a frequency shift without affecting the amplitude. The frequency shifted horizontal luminance level signal from unit 72 is operated (time compressed) by means of a picture format encoder 74. That is, the encoder 74 encodes the frequency shifted horizontal luminance levels such that this signal has an active line interval of 50μsec, and thus less than the standard active NTSC interval of 52.6ps; this is done by using methods that will be explained in connection with FIGS. 6 to 8. When the input signal to the encoder 74 is compressed in time, its bandwidth increases from approximately 1.0 MHz to 1.1 MHz at the output of the encoder 74. The signal from the encoder 74 is amplified by means of a device 76 similar to that shown in FIG is similar to an intraframe averaging before it is applied as a signal to the unit 60. The intra-frame averaged signal Z is substantially identical to the signal from the encoder 74 because of the highly visible correlation of the image information within one frame of the signal coming from the encoder 74. The modulation signal X is a composite signal containing luminance and chrominance information, and the modulation signal has substantially the same bandwidth of approximately 0 to 1.1 MHz.

As will be explained in connection with FIG. 24, the unit 80 performs a non-linear amplitude compression after a gamma function at large amplitude excursions of the two additional signals X and Z, before these signals quadrature modulate an alternating subcarrier signal ASC. A gamma value of 0.7 is used whereby the absolute value of each sample is raised to the 0.7th power and multiplied by the sign of the original sample value. Gamma compression reduces the visibility of potentially large amplitude swings of the modulated signals in normal receivers and allows predicted recovery in the wide-screen receiver because the inverse function of the gamma function used in the encoder is predictable and can be easily realized in the decoder of the receiver.

The amplitude-compressed signals were then quadrature-modulated onto the 3.1075 MHz phased alternating subcircuit ASC whose frequency is an odd multiple of one-half horizontal line frequency (395 x H / 2). The phase of the alternating subcarrier is changed from one to the next field by 180 °, unlike the phase of the color subcarrier, which does not change from field to field. The field-alternating phase of the alternating subcarrier allows the modulating information of the additional signals X and Z to overlap with the chrominance information. Complementary-phase additional information components A1, -A1 and A3, -A3 of the modulated additional signal are formed. This makes it possible to separate the additional information in the receiver using a relatively uncomplicated field memory. The quadrature modulated signal M is added in the adder 40 with Hern signal N. The resulting NTSCF signal is a 4.2 MHz NTSC compatible signal.

The described nonlinear gamma function used in the coder serves to compress large amplitudes. It forms part of a non-linear compressing and expanding companding / expansion system which, as a further component for the expansion of the amplitude, contains a complementary gamma function in the decoder of a widescreen receiver, as will be described below. It has been found that the disclosed non-linear companding system significantly reduces the impact of non-standard additional information on standard image information without visibly degrading an image by noise effects. The companding system uses a non-linear gamma function to compress large amplitude excursions of the additional non-standard high-frequency wide-screen information in the encoder and a complementary non-linear gamma function to correspondingly stretch such high-frequency information in the decoder. The result is less interference with existing standard video information in the described compatible widescreen system, whereby the non-standard additional wide-screen information is split into low-frequency parts and high-frequency parts subjected to companding. In the decoder, nonlinear amplitude expansion of the compressed high frequency information does not result in excessively high noise. That is, large amplitudes of high frequency information are typically associated with high contrast image edges, and the human eye is insensitive to the noise at such edges. The described companding process also advantageously reduces cross-modulation products between the alternating subcarrier and the chrominance subcarrier so that associated visible beat phenomena are also reduced.

The luminance detail signal YT in FIG. 1 a has a bandwidth of 7.16 MHz and is coded by means of a picture format coder 78 to the 4: 3 picture format (for example in the same way as shown in FIG. 6) and then undergoes in a filter 79 horizontally low pass filtering 750KHz to produce a signal YTN. The edge subareas are low-pass filtered prior to their time compression to 125 KHz, which is done in an input-side low-pass filter of the Bildformatkoder 78 which corresponds to the input filter 610 of the device shown in Figure β, but a cut-off frequency of 125 kHz hi t. The heights of the edge parts are suppressed. Thus, the signal YTN is spatially correlated with the main signal C / SL.

The YTN and NTSCF signals are converted from the (binary) digital form to the analog form by digital / analog converters (A / D) 53 and 54 prior to being passed to an RF quadrature modulator 57 for transmission to a television RF Aufzumodulieren carrier signal. The RF-modulated signal is then applied to a transmitter 55 to radiate it via an antenna 56.

The alternating subcarrier ASC used in the modulator 80 is horizontally synchronized, and the frequency is chosen so as to ensure sufficient separation of edge-part and center-part information (e.g., 20 to 30 dB), and has no significant effect on an image which is reproduced with a standard NTSC receiver. The ASC frequency should preferably be an interlaced frequency with an odd multiple of the horizontal horizontal line frequency, so as not to cause disturbances which could affect the quality of a given image, as in the past.

The quadrature modulation performed in this way in the unit 80 advantageously allows the simultaneous transmission of two narrow-band signals, dio time expansion and modulation of high-frequency signals leading to oinor

Bandwidth reduction according to the narrow band requirements of quadrature modulation. The more that

Bandwidth is reduced, the less likely it is that interference between carrier and Modulus signals occur. In addition, the typically high-energy DC component) of the edge-portion information is forced into the oversampling area and not applied as a modulation signal. Thus, the energy of the modulation signal and therefore the associated interference is very reduced.

The encoded and transmitted via the antenna 56 NTSC-compatible widescreen signal is to be received by both NTSC receivers as well as three-party receivers, as shown in the figure 13.

Referring to FIG. 13, a broadcast widescreen enhanced wide-screen television signal is intercepted with an antenna 1310 and fed to the antenna input of an NTSC receiver 1312. The receiver 131? processes the compatible widescreen signal in a normal manner to reproduce a 4: 3 aspect ratio image with the wide part edge information (ie the "depths") pressed into the horizontal oversampling areas out of the view of the observer, and partially (ie The "treble") is included in the mod modulated auxiliary subcarrier signal which does not interfere with the operation of the standard receiver. The Wide Format Compatible Widescreen (EDTV) signal captured by the antenna 1310 is also applied to a progressive scan widescreen receiver 1320 which includes a Picture with a large aspect ratio of z, b. 5: 3 can play. The received widescreen signal undergoes first processing in an encoder unit 1322 including an RF turner and amplifier circuits, a baseband video signal generating synchronous video modulator (a quadrature demodulator), and analog to digital converter circuitry (A / D). to generate a baseband video signal (NTSCF) in binary form. The A / D converter circuits operate at a sample sampling frequency equal to four times the color subcarrier frequency (4x fsc).

The signal NTSCF is applied to an intraframe processing unit 1324 in which image lines are preprocessed in a frequency range above 1.7 MHz, each within a respective frame by 262 horizontal line periods (262H) auseinerder so the main signal N and the quadrature modulated additional signal M practically free of vertical and temporal crosstalk (VT crosstalk). Between the lying at 1.7MHz lower limit of the operating frequency of the unit 1324 and lying at 1.5MHz lower limit of the Betriobsfrequenz the unit 38 in the encoder of Figure 1a is a 200KHz wide security band against horizontal crosstalk. The recovered signal N contains information substantially visually identical to the image information of the main signal C / SL, because of the good correlation of the visual information within a frame of the original main signal C / SL, which detects the intraframe averaging in the encoder Figure 1 has experienced a.

The signal M is applied to a quadrature demodulator and amplitude dilation unit 1326 to demodulate the additional signals X and Z by means of an alternating subcarrier ASC whose phase changes from field to field, similar to the subcarrier signal ASC described in connection with Figure 1a. The demodulated signals X and Z contain information which is visually substantially identical to the image information of the signal ESH and the output of the unit 74 in Figure 1a, respectively, due to the correlation of the well-visible information within each one frame in these signals. which have undergone intraframe averaging in the encoder of Figure 1a. The unit 1326 also includes a 1.5 MHz lowpass filter to remove unwanted high frequency demodulation products at twice the frequency of the old rnie ^r forming the auxiliary carrier, and an amplitude expansion circuit to stretch the (previously compressed) demodulated signals using an inverse gamma function which is the inverse of the non-linear compression function used by the unit 80 in Figure 1 (ie has a gamma value of 1 / 0.7 = 1.423). A unit 1328 effects time compression of the color coded high frequency components of the edge portion information so that they again occupy their original time slots, thereby recovering the signal NTSCH. The unit 1328 compresses the signal NTSCH temporally by the same amount by which the unit 62 in Figure 1a has temporally expanded the signal NTSCH. An encoder 1330 decodes the signal Z in containing the horizontal high frequency components of the luminance (Y-treble) the widescreen format, by time stretching this signal by the same amount as the time compression of the corresponding component in the encoder of Figure 1a was and indicated in Figure 17 using the imaging techniques described therein.

A modulator 1332 outputs the signal from the decoder 1330 by means of amplitude modulation to a 5.0 MHz carrier f _c . The amplitude modulated signal is then high pass filtered in a filter 1334 at a cutoff frequency of 5.0 MHz to suppress the lower sideband. In the output of the filter 1334, the frequencies of the sub-part information of 5.0 to 6.0 MHz and the frequencies of the sub-part information of 5.0 to 6.0 MHz are recovered. The signal provided by the filter 1334 is given to an adder 1336.

The signal NTSCH from the compressor 1328 is applied to a unit 1340 to separate the luminance heights from the chrominance heights and to generate signals YH, IH, and QH. This can be done by means of the arrangement shown in FIG.

The signal N from the unit 1324 is separated into its luminance and chrominance components YN, IN and QN in a luminance / chrominance separator 1342. The separator 1342 may be similar to the separator 1340 and use an arrangement of the type shown in FIG.

The signals YH, IH, QH and YN, IN, QN are applied to the inputs of a Y-I-Q picture format decoder 1344, which decodes the luminance and chrominance components into the wide format. The edge part depths are time stretched, the center part information is temporally compressed, the edge part heights are added to the edge part depths, and the edge parts are combined with the middle part to form the ten-pixel wide overlap area, using the one illustrated in FIG principles. Details of the decoder 1344 are shown in FIG. The signal YF 'is fed to the adder 1336, where it is summed with the signal coming from the filter 1334. By this process wied'.i obtained information with extended high-frequency horizontal details of the luminance is added to the decoded luminance signal YF '.

The YF ', IF' and QF 'signals are converted to progressive scan format using framing converters 1350, 1352 and 1354, respectively, in the interlaced scanning format. The luminance image-to-noise converter 1350 also responds to the "helper" luminance signal YT from the image format decoder 1360, which decodes the decoded "helper signal" YTN. The encoder 1360 decodes the YTN signal into the widescreen format and has a structure similar to the structure shown in FIG.

Scaling converters 1352 and 1354 for the I and Q components, respectively, bring the interlaced signals into progressive format

Sampling in that they form the time average of lines which are one frame apart so as to obtain the information for the respective missing line of the progressive scan; this can be done by means of a device of the type shown in FIG.

The luminance image-frame converter 1350 is similar to the arrangement shown in Fig. 20 except that the signal YT is added as shown in the arrangement of Fig. 21. In this unit, a sample of the "helper" signal YT is added to a time average to help reconstruct a missing progressive scan pixel. The complete time details are recovered within the band of horizontal frequencies contained in the encoded row differential signal (750KHz, after encoding) Above this band of horizontal frequencies, the signal YT is equal to zero, so that the missing picture element is reconstructed by time averaging.The widescreen signals YF, IF and YF now in the progressive scan format are recorded by means of a digital / analogue converter 1362 in FIG have been brought into analog form before being applied to a video signal processor and matrix amplifier unit 1364. The video signal processor in unit 1364 includes signal amplification, DC level shifting, peaking, brightness control, contrast control, and others video signal treatments. The matrix amplifier 1364 combines the luminance signal YF with the color difference signals IF and QF to produce the video color signals R, G and B. These color signals are amplified in unit 1364 by driver amplifiers to a level suitable for directly driving a color capable wide-screen player, such as a video recorder. B. a widescreen picture tube, is suitable.

FIG. 8 shows an arrangement contained in the processing unit 18 of FIG. 1a for developing the signals YE, YO and YH from the wide-band wide-area signal YF. The signal YF "undergoes horizontal low-pass filtering in an input filter 610 at a cut-off frequency of 700KHz to produce the low-frequency luminance component signal YL which is fed to one input of a subtraction combining circuit 612. The signal YF" is applied to the other input of the Combination circuit 612 and a time-demultiplexing circuit 616, after being delayed in a unit 614 to compensate for the signal delay caused by processing in the filter 610. The combination of the delayed signal YF "and the filtered signal YL supplies the signal YH containing the high-frequency luminance components at the output of the combination circuit 612.

The delayed signal YF "and the signals YH and YL are supplied to separate inputs of the demultiplexer 616, which includes individual demultiplexer units (abbreviated DEMUX) 618, 620 and 621 to process the signals YF", YH and YL. The details of the demultiplexer 616 will be explained in connection with FIG. The demultiplexer units 618, 620 and 621, respectively, provide the full-bandwidth center sub-signal YC, the margin signal YH and the margin sub-signal YL 'as shown in Figs.

The signal YC is time stretched in a time expander 622 to produce the signal YE. This stretching is done with a center stretch factor that is sized to leave room for the left and right horizontal oversampling areas. This center part stretch factor (1.19) is the ratio of the desired width of the signal YE (pixels 15 to 740) to the width of the signal YC (pixels 75 to 680) as shown in FIG.

The signal YL 'is time compressed in a time compressor 628 by an edge compression factor to produce the signal Y0. The edge compression factor (6.0) is the ratio of the width of the corresponding part of the signal YL '(eg, the left picture elements 1 to 84) to the desired width of the signal Y0 (eg, the left picture elements 1 to 14) ) as shown in Fig. 3). The time stretchers 622, 624 and 626 and the time compressors 628 may be of a type as shown in FIG. 12 and described below.

The signals IE, IH, IO and QE, QH, QO are developed from the signals IF "and QF" in a similar manner as the signals YE, YH and YO with the aid of the device according to FIG. which shows a device for generating the signals IE, IH and IO from the signal IF "The development of the signals QE, QH and QO from the signal QF" takes place in a similar way.

Referring to FIGURE 1, wideband wide-band signal IF "is applied to demultiplexing circuit 716 after delay in unit 714 and also subtractively combined in subtractive combining circuit 712 with low-frequency signal IL from low-pass filter 710 to produce high-frequency signal IK IF "and the signals IH and IL are respectively decoded by demultiplexers 718, 720 and 721 in the demultiplexing circuit 716 so that the signals IC, IH and IL 'are obtained. The signal IC is time-expanded in a time-stretching means 722 to generate the signal IE, and the signal IL 'is time-compressed in a time compressor 728 to produce the signal IO. The stretching of the signal IC is done with a similar mid-section expansion factor as the described stretching of the signal YC, and the compression of the signal IL 'is done with a similar edge part compression factor as the described compression of the signal YL'.

FIG. 8 shows a demultiplex circuit 816 as may be used for the circuit 616 in FIG. 6 and the circuit 716 in FIG. Figure 8 shows the circuit in its application as the demultiplexer circuit 616 of Figure 6. The input signal YF "contains 754 picture elements which define the picture information The picture elements 1 to 84 define the left edge part, the picture elements 671 to 754 define the right edge part, and the pixels 75 to 680 define the central portion which slightly overlaps the left and right edge portions, and the signals IF "and QE" show a similar overlap As will be described, such an area overlap will allow the center portion and edge portions in the receiver The demultiplexing circuit 816 includes first, second and third demultiplexing units (DEMUX) 810, 812 and 814, respectively, for the information of the left edge portion and the center portion, respectively Each demultiplexer unit has an input "A" to which the signal YH or YF "and YL, respectively, and an input" B "for applying a blanking signal

(BLK). The blanking signal can be, for example, a logic value 0 or ground potential. The unit 810 extracts the output signal YM including the heights of the left and right edge portions from the input signal YH, as long as the select input SEL from a count comparator 817 receives a first control signal indicating the presence of the left edge portion pixels 1 to 84 and the first Display pixels 671 to 754 of the right edge portion. At other times, a second control signal from the count comparator 81 7 causes the blanking signal BLK from the input Ban to be coupled to the output of the unit 810 instead of the signal YH from the input A. The unit 814 and a count comparator 820 similarly operate to derive the edge sub-depth signal YL 'from the signal YL. The unit 812 couples the signal YF "from its input A to its output to produce the mid-section signal YC only when a control signal from a count comparator 818 indicates the presence of the mid-section pixels 75-680.

Dio count comparators 817, 818 and 820 are synchronized with the video signal VF "by a pulse signal from the output of a counter 822 responsive to a clock signal of four times the color subcarrier frequency (4x fsc) and a horizontal line sync signal H derived from the video signal YF". Each output pulse of the counter 822 corresponds to a pixel position along a horizontal line. The counter 822 has an initial header beginning with the count -100 corresponding to the 100 pixels from the beginning of the negative directed horizontal sync pulse at the time Ths to the end of the horizontal blanking interval at which the picture element 1 appears at the beginning of the horizontal line display interval. Thus, counter 822 is counted to "1" at the beginning of the line refresh interval Other counter arrangements may be developed The principles employed by demultiplexer circuit 816 may also be used in multiplexing circuits to perform a reverse signal combining operation, such as that described in US Pat B. runs in the edge part middle part combination circuit 28 of Figure 1a.

FIG. 9 shows details of modulator 30 in encoders 31 and 60 of FIG. 1a. Referring to Figure 9, signals IN and ON appearing at four times the chrominance carrier frequency (4x fsc) are respectively applied to the signal input of an associated latch (latch) 910 and 912, respectively. Dio latch circuits 910 and 912 also receive clock signals of frequency 4x fsc to supply the signals 2χ fsc supplied to an inverting switching signal input of latch circuit 910 and a non-inverting switching signal input of latch circuit 912. The signal outputs of the latch circuits 910 and 912 are combined into a single output line on which the signals I and Q appear alternately and go to the signal inputs of a non-inverting latch 914 and an inverting latch 916. These latches are clocked at the frequency 4xfsc and each of them receives a chrominance carrier frequency switching signal fsc, the first at an inverting input and the second at a noninverting input. The non-inverting latch 914 provides at its output in alternating sequence signals I and Q of positive polarity, and the inverting latch 916 provides at its output in an alternating sequence signals I and Q of negative polarity, ie signals -I, -Q. The outputs of the latch circuits 914 and 916 are combined into a single output line on which the signals I and Q appear alternately and in pairs alternately between opposite polarities, that is in the sequence, Q, -I, -Q ... etc., so as to form the signal CN. This signal is filtered in the filter 32 before being combined in the unit 36 with a filtered version of the luminance signal YN to produce the encoded NTSC signal C / SL in the form Y + I, Y + Q, Y-I, Y-Q ₁ Y + I, Q + Q ... etc. FIG. 10 shows a filter for vertical-temporal filtering (vertical-temporal filter or abbreviated VT-filter) by setting weighting coefficients a 1 to a9 into a VT band-pass filter , a VT notch filter or a VT low pass filter. The table in Figure 10a shows the weighting coefficients for VT bandpass filtering and VT bandstop filtering as used in the system disclosed herein. An HVT notch filter, such as a filter 34 of Figure 19, and HVT bandpass filters included in the decoding system of Figure 13, for example, each comprise the combination of a horizontal lowpass filter 1020 and a VT bandstop filter 1021 as shown in Figure 10b and the combination of a horizontal bandpass filter 1030 and a VT bandpass filter 1031 as shown in Figure 10c. In the HVT bandstop filter of Figure 10b, the horizontal low pass filter 1020 has a given cutoff frequency and provides a filtered low frequency signal component. This signal is subtractively combined in a combining circuit 1023 with a delayed version of the input signal from a delay unit 1022 to produce a high frequency signal component. The low-frequency component is using a? Delay network 1024 by one frame period before being applied to an additive combiner circuit 1025 to provide an output signal that has undergone HVT notch filtering. The VT filter 1021 has the weight coefficients indicated in FIG. 10a for the VT bandstop filter. An HVT bandpass filter, as z. For example, as shown in the encoder of Figure 13, Figure 10c consists of a horizontal bandpass filter 1030 having a given cut-off frequency in cascade with a VT bandpass filter 1031 having the weight coefficients indicated in the table of Figure 10a for VT bandpass filters.

The filter of FIG. 10 includes a plurality of cascaded memory units (M) 1010a-1010h to progressively receive signal delays at individual taps 11-19 and provide a total filter delay. The signals taken at the taps are respectively supplied to an input of associated multiplier circuits 1012a to 1012i. Another input of each multiplying circuit receives a prescribed weighting signal a 1 to a 9, depending on the nature of the filtering operation to be performed. The nature of the filtering process also dictates the delays introduced by the storage units 1010a-1010h.

Horizontal dimension filters contain pixel memory elements such that the total filter delay is shorter than one horizontal line period (1H). Vertical dimension filters contain only line storage elements, and temporal dimension filters contain only full-screen storage elements. Thus, a three-dimensional HVT filter consists of a combination of pixel memory elements (<1H), row memory elements (1H), and frame memory elements (> 1H), while a VT filter contains only the last two types of memory elements , The weighted tapped (mutually delayed) signals from the elements 1012a through 1012i are combined in an adder stage 1015 to provide a filtered output signal.

Such filters are non-recursive filters with finite impulse response (so-called Fl-R filters). The nature of the delay introduced by the memory elements depends on the type of signal to be filtered, and in the present example also on what degree of crosstalk between the luminance, the chrominance and the high-frequency edge portion signals can be tolerated. The sharpness of the Filtorgrenzkennlinie and thus the sharpness of the filtering is improved by increasing the number of cascaded memory elements.

FIG. 10d shows one of the separate filters in network 16 of FIG. 1a. This filter includes cascaded memory units (delay units) 1040a to 104Od and associated multiplier circuits 1042a to 1042e having designated weighting factors a 1 to a 5, which receive the signals from taps 11 to 15. Further, a combination circuit 1045 is provided which sums the weighted output signals of the multiplier circuits a 1 to a5 to form an output signal therefrom.

FIG. 11a shows an intraframe averaging device suitable for the intraframe averaging devices 64 and 76 of FIG. 1a. A composite video signal is applied to the delay network having the 262-H delay elements 1110 and 1112 and also to one input of a multiplexer (MUX) 1115 which is switched to a 30-Hz switching signal at the field frequency. The 30 Hz MUX switch signal is vertically synchronized in response to vertical interval sync pulses associated with the composite input video signal. Another input of the multiplexer (MUX) 1115 receives a signal from an output of the delay element 1112. In addition, one combines

Combining circuit 1118 outputs from the multiplexer 1115 and from a point of center tap between the

Delays 1110 and 1112 after these signals have been evaluated by an averaging factor Ίι . This weighting factor may be provided by geegienten matrix networks within the combination circuit 1118 or by means of Signalmultiplizierschaltungen, which are each arranged in the input signal paths of the combination circuit 1118.

The signals "Y1 + C1" and "Y1 + C2" are composite color video signals at 262H intervals in successive first and second fields, and the signal "M1" is an intra-frame averaged output signal as shown in FIG. 1, for example During a first field, the multiplexer 1115 is in an input position "1" and transmits the signal Y2 + C2 to the combiner 1118, in which it is coupled to the signal Y1 + C? the center tap is summed to produce an output-side average signal M1. In the next field, the center tap between the delay elements 1110 and 1112 contains the signal value Y2 + C 2 and the multiplexer 1115 is in position "2" for selecting the signal path from the output of the delay element 1112 containing the signal value Y1 + C1, thereby obtaining the same averaging value The described device produces identical pixels spaced 262 H apart and is not limited in the application of a matte process Any weight values may be used to generate a desired weighted combination of pixel pairs and delays other than 262H may be used (along with an associated change in the switching frequency of the multiplexer), depending on the requirements of a particular system. Averaging device, which is suitable as an intraframe averaging device 38 of Figure 1 a. Figure 11b contains the arrangement of Figure 11a except that the combination circuit 1128 combines the subtractive and not the additive signal combination, and that a filter 1130, a gate 1132, and a combiner 1134 are included. In short, for the output of the combinational circuit 1128, a difference of the fields and not an average value is representative as in the arrangement of Figure 11a. This difference is essentially an erasure term added backwards to the signal Y1 + C1 in the combiner 1134 for canceling the difference between successive fields to ensure that the portions of the successive fields are averaged identically. Filter 1130 filters the cancellation term from the output of combiner 1128 to limit the averaging process to a desired frequency range. The gate 1132 is controlled to detect when the averaging process occurs in an image interval, in this case in the middle portion, except for the temporally compressed edge portions.

In particular, the input signal paths to combiner 1128 have signal weight coefficients of ¹ A and -V j, as shown, such that the output from combiner 1128 equals the difference in the information content of the input signals to combiner 1128, which is 262 H in time in adjacent fields are separated from each other. The complementary weighting coefficients may be provided to the combiner circuit 1128 by using the signal multiplying circuits in the respective input paths, or by the arrangement of the combinational circuit 1128 as the differential amplifier. The output signal from combination circuit 1128 is filtered in a 1.5 MHz horizontal high pass filter 1130 before being applied to an electronic transmission gate 1132. Gate 1132 is responsive to a switching control signal for passing the high frequency signal from the output of a filter 1130 only during the middle portion of the main line (component 1). At this time, the gate 1132 is open (conducting). Gate 1132 is closed (nonconducting) during the time-compressed edge portions of the main signal, for example during the illustrated positive pulse intervals of the control signal. The output from gate 1132 is summed in combination circuit 1134 with the composite video signal appearing at the point of center tap between delay elements 1120 and 1122. The control signal of the gate is vertically synchronized with the associated composite input video signal, triggered by the vertical interval sync pulse. The control signal of the gate is also synchronized horizontally. The horizontal synchronization can be achieved by responding to the horizontal line synchronizing pulse component of the composite input video signal including a picture element counter to determine the timing of the positive pulse components of the gate control signal following each horizontal line synchronizing pulse. Between a horizontal line sync pulse and the first picture element, a predetermined time interval can easily be accommodated.

In the following, reference is again made to FIG. 1 d together with FIG. 11 b. As shown, when multiplexer 1125 is in position 1 and gate 1132 is closed, only the composite video signal Y1 + C1 appears from the center tap between elements 1120 and 1122 at the output of combiner 1134

When the output of the combining circuit 1134 is the unchanged compressed edge portion information of the composite video signal Y1 + C 1 belonging to the field 1. The output of the combining circuit 1134 is the unchanged compressed edge portion information of the composite video signal Y2 + C2 for the subsequent field 2 when the multiplexer 1125 occupies position 2.

When multiplexer 1125 is in position 1 for field 1 and gate 1132 is closed during the mid-portion interval between the edge sub-intervals, the output of combining circuit 1134 contains the signal components Y1 + C1 and M1. The component Y1 + C1 contains unchanged, d. H. not intraframe averaged, the midsection information at and below approximately 1.5MHz. The component M1 contains the intraframe-averaged midsection information above approximately 1.5 MHz. When the multiplexer 1125 is in position 2 during the succeeding field 2 and the gate 1132 is closed during the mid-portion interval, the output from the combiner 1134 includes the intraframe-averaged component 1 as discussed above and the component Y2 + C2. The latter component contains the unmodified (not intraframe averaged) midsection information at and below approximately 1.5 MHz. FIG. 12 shows a raster imaging device that can be used for the time-stretching and time-compression devices in FIGS. 6 and 7. The waveforms referred to herein in Figure 12a illustrate the imaging process. FIG. 12a shows an input signal waveform S with a middle part between the picture elements 84 and 670 which, by means of a time-stretching process, accesses the picture element positions 1 to 754 of FIG

Output waveform W should be displayed. The final image monsters 1 and 670 from the waveform S are opened directly

End picture elements 1 and 754 of the waveform Wab. The intervening pixels are not imaged directly in a position ratio of 1: 1 as a result of the temporal strain, in some cases not even in an integer positional relationship. The latter case applies z. For example, when the pixel position 85, 33 of the waveform S corresponds to the integer pixel position 3 of the output waveform W. The pixel position 85, 33 of the signal S contains an integer part (85), and a fraction DX (0.33), and the pixel position 3 of the waveform W contains an integer part (3) and a fraction (O).

In the arrangement of Figure 12, a pixel counter 1210 operating at frequency 4x fsc provides an output-side write address signal M representing pixel positions (1 ... 754) of an output raster. The signal M is fed to a programmable read-only memory (PROM) 1212 which contains a look-up table of programmed values which depend on the nature of the raster image to be performed, e.g. 8. Whether the image is compressing or stretching. In response to the signal M, the PROM memory 1212 provides an output read address (READ ADDRESS) signal N representing an integer and an output DX representing a fraction equal to or greater than zero but less than 1. In the case of a 6-bit signal DX (2 ⁶ = 64), the signal DX shows the fractions 0, Vm, Ve>"Vu ... <* /».

The PROM memory 1212 permits a stretched or compressed representation of a video input signal S as a function of the stored values of the signal N. Thus, in response to integer values of the pixel position signal M, a programmed value of the read address signal N and a programmed value of the fractional signal DX are provided. For example, to perform signal expansion, the PROM memory 1212 is designed to provide the signals N at a slower rate than the signals M. Conversely, if compression is to occur, the PROM memory 1212 provides the N signals at a faster rate as the signals M.

The video input signal S is delayed by cascaded picture element delay elements 1214a, 1214b, and 1214c to produce video signals S (N + 2), S (N + 1), and S (N), which are mutually delayed versions of the video input signal , These signals are applied to video signal inputs of associated dual port memories 1216a-1216d, as known in the art. The signal M is applied to the write address input of each of the memories 1216a to 1216d, and the signal N is applied to the read address input of each of the memories 1216a to 1216d.

The signal M determines where incoming video signal information is to be written into the memories, and the signal N determines which values are to be read from the memories. The memories are designed so that information can be written into one address while simultaneously reading out another address. The output signals S (N-1), S (N), S (N + 1) and S (N + 2) from the memories 1216a to 1216d show a time-expanded or time-compressed format depending on the read / write operation of the memories Memory 1216a to 1216d, which depends on how the PROM memory 1212 is programmed.

The signals S (N-17), S (N), S (N + 1), and S (N + 2) from the memories 1216a to 1216d are processed in a four-point linear interpolator, the peaks 1220 and 1222 (FIG. Peak filter), a PROM memory 1225, and a linear two-point interpolator 1230; Details of these parts are illustrated in Figures 12b and 12c. The peak filters 1220 and 1222 receive three signals from the group of signals S (N-1), S (N), S (N + 1), and S (N + 2) in the shown selection and also receive a peak signal PX. The value of the peak signal PX changes from 0 to 1 as a function of the value of the signal DM as shown in Figure 12d and is output from the PROM memory 1225 in response to the signal DX. The PROM memory 1225 contains a look-up table and is programmed to provide a given value of PX in response to a given value of DX.

The peak filters 1220 and 1222 provide two mutually delayed video signals S '(N) and S' (N + 1) which have been peaked to the two-position linear interpolator 1230, which also receives the signal DX. The interpolator 1230 provides a (compressed or stretched) video output W defined by the term

W = S '(N) + DX [S' (N + 1) -S '(N)].

The described function of four-point interpolation and peaking is advantageously approximated to a (sin X) / X interpolation function with good resolution of the high-frequency details.

Figure 12b shows details of the peak filters 1220 and 1222 and the interpolator 1230. Referring to Figure 12b, the signals S (N-1), S (N) and S (N + 1) are applied to a weighting circuit 1240 in the peak filter 1220, in These signals (in the order listed) are weighted with peak coefficients - '/ · », V2 and -' /«. As shown in Fig. 12c, the weighting circuit 1240 includes multiplying circuits 1241a to 1241c for receiving the signals S (N-1), S (N) and S (N + 1), respectively

with the associated peak coefficient -'Λ or V2 or - ¹ A to multiply. The output signals of the multiplier circuits 1241a through 1241c are summed in an adder stage 1242 to produce a peak signal P (N) which is multiplied in a further multipath circuit 1243 by the signal PX to provide a peak signal which is then applied in an adder stage Is summed with the signal S (NJ) to provide the peak signal S '(N) The peak filter 1222 has a similar structure and operation.

In the two-point interpolator 1230, the signal S '(N) is subtracted from the signal S' (N + 1) by means of a subtractor 1232 to produce a difference signal which is multiplied in a multiplier circuit 1234 by the signal DX. The output of multiplier circuit 1234 is summed in an adder stage 1236 with signal S '(N) to produce output signal W.

FIG. 15 shows details of the intraframe processing device 1324 of FIG. 13. The decoding device of FIG. 15 basically corresponds to the coding device of FIG. 11b. A composite input video signal for the processing means 1324 in Fig. 15 includes in a first field the signal components "Y1 + C1" and "M1 + A1". In a subsequent second field, the input signal contains the components "Y2 + C2" and "M1-A1". The components Y1 + C1, M1 and Y2 + C2, M1 are components supplied by the intraframe processing device 38, as described in more detail in connection with Figure 11b. The components + A1 and -A1 represent the alternating

Subcarrier signal, which with the component 2 and the component 3 of the intraframe-averaged information of the units 64th

and 76 is modulated for the respective subsequent fields. In this context, reference is made to FIGS. 1 a and in particular to FIG. 1 d.

The intraframe processor of Figure 15 operates in substantially the same manner as the Figure 11b arrangement previously discussed. If the multiplexer 1525 is in position 1, the component of the field difference is formed at the output of the combination circuit 1528. After filtering by high pass filter 1530 and gate combination by unit 1532, a component -A1 is obtained which, when combined in a combining circuit 1534 with signal Y1 + C1, M1 + A1, clears the modulated auxiliary subcarrier component (KAI) to generate a recovered main signal Y1 f-C1, M1. The component Y1 + C1 of the recovered main signal is not changed below the cut-off frequency of the 1.7 MHz high-pass filter 1530, and the component M1 represents the intraframe-averaged center part information above approximately 1.7 MHz. The field difference canceling term (-A1) after inversion in the unity gain amplifier 1535 is the recovered modulated additional signal A1.

The recovered main signal Y1 + C1, M1 corresponds to the signal N in FIG. 13 and is further processed in a network 1342 as explained. The recovered additional signal A1 corresponds to the signal M in FIG. 13 and is demodulated in a network 1326.

Figure 16 illustrates the operation of the network 1324 as shown in Figure 15 for the next succeeding one

Field. In this case, the signal Y2 + C2, M1-A1 is developed between the delay elements 1520 and 1522,

and the multiplexer 1528 occupies the position 2 for the received signal Y1 + C1, M1 + A1. A recovered main signal Y2 + C2, M1 is generated at the output of the combinational circuit 1534, and an opposite phase-modulated additional signal -A1 is recovered.

In FIG. 18, an H-V-T bandpass filter 1810, which is constructed like that of FIG. 10c and has a passband of 3.58 + 0.5 MHz, passes the signal NTSCH to a subtractive combining circuit 1814, which also includes the FIG

Signal NTSCH receives after passing through a delay compensating delay 1812. The

separated luminance level signal YH appears at the output of combiner 1814. The filtered NTSCH signal from filter 1810 is quadrature demodulated in demodulator 1816 in response to chrominance subcarrier signal SC to produce chrominance levels IH and QH.

In Fig. 19, the signals YN, IN and QN are divided into the compressed edge part depths Y0, IO, Q0 and the extended center part signals YE, IE, QE by means of an edge middle part signal separator (time demultiplexer) 1940. *** " The demultiplexer 1940 may operate using the principles of the demultiplexer 816 of the previously discussed figure. Signals YO, IO, and QO are time-stretched by a time-expansion means 1942 by an edge-strain factor (corresponding to the edge compression factor in the encoder of Figure 1 a) to restore the original ranking of the edge-portion depths in the wide-screen signal; these recovered edge part depths are labeled YL, IL and QL. Similarly, in order to make room for the edge portions, midrange signals YE, IE, and QE in a time compressor 1944 are time-compressed about a mid-portion compression factor (corresponding to the mid-scale expansion factor in the encoder of Figure 1 a) to the original spatial Restore alignment of the mid-section signal in the wide-screen signal; the recovered mid-section signals are labeled YC, IC and QC. The time compressor 1944 and the time expander 1942 may be of a type previously described in FIG.

The spatially recovered edge portion heights YH, IH, and QH are combined with the spatially recovered edge portion depths YL, IL, and QL in a combining circuit 1946 to produce reconstructed edge portion signals YS, IS, and QS. These signals are combined with the reconstructed center-part signals YC, IC and QC in a combiner 1960 to form a fully reconstructed wide-field luminance signal YF 'and fully reconstructed wide-field color difference signals IF' and QF '. The assembly of the edge part and middle part signal components is done in a manner in which a visible edge at the boundaries between the middle part and the edge parts is virtually eliminated, as will be apparent from the following description of the combiner 1960 shown in FIG.

In FIG. 20, the interleaved IF '(or QF') signals are delayed by 263H by element 2010 before being applied to one input of a dual port memory 2020. This delayed signal experiences another delay of 262 H in a 2012 element before being added to the input signal in an addition stage 2014. The output of the adder 2014 is applied to a frequency bisecting network 2016 before being applied to one input of a dual port memory 2018. The reading of the data from the memories 2020 and 2018 takes place at a speed of 8x fsc, while the data is written at a speed of 4x fsc. The outputs of the memories 2018 and 2020 are applied to a multiplexer 2022 to provide output signals IF (QF) in the format

to produce progressive scanning. In Fig. 20, there are also the waveforms of the interlaced input signal (two lines of which one includes the picture element scanning patterns C and the other of which the picture element scanning patterns X) and the progressive scan format output having the same picture element scanning patterns C and X shown.

FIG. 21 shows an arrangement that can be used as the transducer 1350 for the signal YF 'in FIG. The interlaced signal YF 'is delayed by elements 2110 and 2112 before it is combined with the instantaneous version in adder 2114. The delayed signal from element 2110 will stick to a dual port memory 212 '. The output of the adder 2114 is applied to a frequency bisecting network 2116, the output signal of which is added to the signal YT in an adder 2118. The output signal you addition stage 2118 is supplied to a Poppelanschluß memory 2122. The memories 2120 and 2122 are operated at a write frequency of 4x fsc and at a read frequency of 8x fsc and provide output signals to a multiplexer 2124 which provides the progressive scan signal YF.

FIG. 14 shows a peripheral part-middle combiner, which is e.g. B. can be used for the element 1960 in Figure 19. The combiner illustrated in FIG. 14 includes a network 1410 that generates the full-bandwidth luminance signal YF 'from the edge-portion luminance component YS and the center-portion luminance component YC, and an I-signal combiner 1420 and a Q-signal combiner 1430 constructed as shown in FIG and operation are similar to the network 1410. The center area and the edge portions are intentionally overlapped by some picture elements, e.g. B. 10 picture elements. Si with the center and edge signals have some picture elements in redundant fashion throughout the encoding and transmission process prior to their merge.

In the widescreen receiver, the center part and the edge parts are recovered from the respective signals concerned, however, because of the experienced temporal stretching, temporal compression and filtering of the signals associated with each area some pixels are distorted or distorted at the boundaries between edge parts and middle part. The waveforms of signals YS and YC shown in Figure 14 show the overlap areas (OL) and the corrupted pixels (CP, somewhat exaggerated for purposes of illustration). If there were no overlapping areas of the individual parts, the falsified picture elements would collide, so that a hem would be visible. It has been found that an overlap area of 10 pixels is wide enough to compensate for three to five falsified boundary pixels. The red-inked picture elements advantageously allow a mutual blending of edge and middle parts in the overlapping area. A multiplier circuit 1411 multiplies the edge sub-signal YS by a weight function W in the super-loacting area as indicated by the associated waveform before applying the signal YS to a combination circuit 1415. Similarly, a multiplying circuit 1412 multiplies the center sub-signal YC η lit of a complementary weighting function (1-W) in the overlapping areas as shown with the associated waveform before passing the signal YC to the combining circuit 1415. These weight functions have a linear ramp-shaped characteristic in the overlap areas and have values between 0 and 1. After the evaluation, the edge-part and center-part picture elements in the combination circuit 1415 are summed so that each reconstructed picture element is a linear combination of edge-part and center-part picture elements.

The weight functions should preferably approach the value 1 near the inner limits of the overlap area and the value 0 near the outer limits. This ensures that the corrupted picture elements have relatively little influence on the boundaries of the reconstructed areas. The illustrated linear ramped weight function fulfills this requirement. However, the weight functions need not necessarily be linear, nonlinear weight functions curvilinear or rounded in the end regions (i.e., near the weight points 1 and 0) may also be used. Such a weighting function can be easily achieved by filtering a linear ramped weight function of the type shown.

The weight functions W and (1-W) can be easily generated by a network including a look-up table responsive to an input signal indicative of pixel positions and a subtractive combination circuit. The pixel positions in the overlap area between marginal and center portions are known, and the look-up table may be appropriately programmed to provide output values of 0 to 1 corresponding to the weight function W in response to the input signal. The input signal can be developed in various ways, e.g. By a counter synchronized with each picture line sync pulse. The complementary weight function (1-W) can be obtained by subtracting the weight function W from 1.

FIG. 22 shows an arrangement that can be used as the image converter 17c for the YF signal in FIG. 1a to convert this progressive scan format signal to the interlaced format. Figure 22 also shows a diagram of a portion of the progressive scan input signal YF with the sampling patterns A, B, C and X in the plane containing the vertical (V) and temporal (T) dimension, as shown in Figure 2a is shown. The progressive scan signal YF is subjected to a delay of 525 H in each of elements 2210 and 2212 to produce relatively delayed scan patterns X and A from the scan pattern B. The sample patterns B and A are summed in an adder stage 2214 before being fed to a frequency bisector network 2216. An output of the network 2216 is subtractively combined in a network 2218 with the scan pattern X to produce the signal YT. This signal is applied to an input of a switch 2220 which is operated at a frequency equal to twice the line scan frequency of the interlaced scan. Another input of the switch 2220 receives the delayed signal YF from the output of the delay element 2210. The output of the switch 2220 is applied to a dual port memory 2222 which is operated at a write speed of 8x fsc and a 4geschwindigkeit fsc lot speed to an output to deliver the signals YF 'and YT in interlaced format.

FIG. 23 shows an arrangement which can be used to implement the image-scanning converters 17a and 17b in FIG. 1a. Referring to Figure 23, the progressive scan format signal IF (or QF) is applied to a 525 H delay element 2310 before being fed to a dual port memory 2312 operating at a read speed of 4x fsc and a write speed of 8x fsc. to produce the IF output signal IF '(or QF') in interlaced format. Also shown in FIG. 23 are waveforms of the progressive scan format input signal

comprising a first row containing the scanning patterns C and a second row including the scanning patterns X and the interlace output signal (the first row with the scanning pattern C stretched by scanning at the half line frequency). The dual port memory 2312 provides at its output only the first line (scan pattern C) of the input signal in stretched form.

FIG. 24 shows details of the unit 80 of FIG. 1a. The signals X and Z are respectively fed to the address input of an associated nonlinear amplitude compression means 2410 and 2412, respectively. Compressors 2410 and 2412 are programmable read only memories (PROM memories), each of which contains a look-up table of programmed values corresponding to the desired non-linear gamma compression function. This function is shown in FIG. 24 under the unit 2412 by a curve showing the output instantaneous values as a function of the input-side instantaneous values. The compressed signals X and Z from the data outputs of the units 2410 and 2412 are respectively given to the signal input an assigned multiplier circuit 2414 and 2416, respectively. The reference inputs of the multiplier circuits 2414 and 2416 receive the alternating subcarrier ASC in a quadrature phase relationship shifted by 90 ° with respect to each other, ie. H. in sine or cosine form. The output signals of the multiplier circuits 2414 and 2416 are combined in a combinational circuit 2420 to provide the quadrature-modulated one. To generate signal M. In the decoder of Figure 13, the compressed signals X and Z are retrieved by a conventional quadrature demodulation technique and undergo complementary nonlinear amplitude expansion in associated lookup PROM memories programmed with words, wek he complementary to the values in the PROM Stores are 2410 and 2412 respectively.

Claims (12)

¹ . An arrangement for processing a television signal to display a wide image having edge portion image information and body image information and an aspect ratio greater than that of a standard television image, the apparatus comprising: means responsive to said representative signal for developing a first component in which the peripheral image information is compressed in time into an over-sampling area of said first component and which contains the body image information, characterized by: first means (38) for intraframe processing of the body image information without said timing compressed edge part information. An arrangement according to claim 1, characterized in that said first means (38) subject said body information, without said boundary information, to intraframe communication. 3. Arrangement according to claim 1, characterized in that said first means processed Infi aframe information is generated containing mutually exclusive groups with substantially identical intraframe pixel information. An arrangement according to claim 1, characterized by means responsive to said representative signal for developing a second component (X) containing the additional image information, second means (64) for intraframe processing of said second component, and means for combining (80, 40) the processed first and second components in a complementary manner to produce a combined signal (M). An arrangement as claimed in claim 4, characterized in that said first means (38) includes intraframe message means and generates processed intraframe information including mutually exclusive groups having substantially identical intraframe pixel information and said second means (64 ) contains an intraframe message and contains processed intraframe information containing mutually exclusive groups with substantially identical intraframe pixel information. A device according to claim 4, characterized by means responsive to said third component developing representative signal containing said additional image information, third means (76) for intraframe processing said third component, means (80) for Modulating a phased alternating subcarrier other than a chrominance subcarrier with said infra-frame-processed second (X) and third (Z) components, and wherein said combining means (40) comprises said first component and said modulated subcarrier combined to produce a composite signal (NTSCF). 7. Arrangement according to claim 6, characterized in that said second (X) and third (Z) components include a non-standard television information. 8. An arrangement according to claim 6, characterized in that said third component (Z) contains the high-frequency luminance detail information without the low-frequency luminance information. 9. Arrangement according to claim 6, characterized in that said alternating subcarrier has a phase which reverses from one field to the next. An arrangement according to claim 6, characterized in that said first (N) second (X) and third (2) components and said alternating subcarrier (ASC) are baseband signals. 11. Arrangement according to claim 6, characterized in that said composite signal modulates an RF carrier. 12. An arrangement according to claim 1, characterized in that said time-compressed image information comprises the low-frequency information without the high-frequency information.