DD275540B5 - Partial discharge sensor - Google Patents

Description

The invention relates to the field of partial discharge measurement in the on-site diagnosis of electrical equipment, such as transformers, generators, motors, switchgear including solid-insulated and SF ₆ switchgear, cable networks, etc., without activation of these objects. In addition to the cyclical or permanent status diagnosis, this also provides opportunities for anomalous early detection and thus the triggering of shutdowns of endangered objects in the event of a disruptive breakdown.

In addition to the classical and standardized partial discharge measurement technique ("partial discharge measurement", IEC publication [1981]), a small probe and a more favorable useful-to-noise ratio are sought for on-site diagnostics, for which it is known to compensate the interference signals in bridge circuits. Zaengl, W .; Klauss, A .: On-site surveillance of potential transformers by means of DP measurement CIGRE symposium, Vienna 1987, Section 700-01] or to hide the time-limited interference components with electronic discriminator circuits [Lemke, E .: Diagnosis of electrical equipment on the basis of very broadband partial discharge measurements ELEKTRIE, Berlin 38 [1984] 10, pp. 387-398) A 50 Hz test AC voltage based method for detecting internal TE pulses is described in DE-OS 2 721 353 Disturbances, eg external discharges, which occur in the region of the maximum of the test voltage, are caused by special electronic K Short suppressed and thus not used for the evaluation. WP 253 333 discloses an arrangement for suppressing the HF interference voltages superimposed on the partial discharge signals, which consists essentially of a push-pull transistor circuit with diodes and a controllable resistor between the two bases for generating a quiescent current. To the control electrode of the controllable resistor is connected via an isolating capacitor and a rectifier circuit with optimal timing and basic level setting the output of the Nutz-Störsignalgemisch amplifying control amplifier and a decoupling diode of the control pulse so that when the active control signal, only the electrical partial discharge pulses are transmitted through the push-pull transistors.

DE-OS 3 630 026 discloses an ultrasonic sensor for the indication of TE from a partially flat trained, fixed in several curvatures and attached to the specimen optical fiber loop which is coupled at one end with a monochrome light source and at the other end with a light sensor. The ultrasonic waves caused by TE cause elastic changes in the radii of curvature of the light guide, thus coupling and loss or modulation of the light, which is used for indication and evaluation.

DE-PS 3 408 256 discloses a measuring probe for detecting the electromagnetic radiation field resulting from the ignition of TE with frequency components well above 100 MHz by means of a broadband antenna which consists of a three-electrode system connected to one another via a compensation network. On the side facing the test specimen, a frame-shaped compensation electrode is arranged between a measuring electrode and an opening provided with a comprehensive housing-like reference electrode, and within the reference electrode differential amplifiers are integrated whose rise time is very small compared to the duration of TE pulses and nanosecond lies. The arrangement of the electrodes results in a good straightening effect and by their interconnection increased interference signal reduction of far fields.

The condition monitoring described above requires the constant readiness for operation of the TE probe. Since the troubled Störsignalunterdrückung can only be fully effective in potential-free operation (without mains connection), a cyclic replacement of internal batteries or their recharging is inevitable with permanent condition monitoring, so that in addition to the complex maintenance intermittent service interruptions can not be excluded. Consequently, a second probe would have to be available for each measurement point to ensure redundancy if a critical error occurs at the moment of maintenance.

Also, the circuit complexity for the previously known TE probe (DD-WP 214 461) is not insignificant, but their possible uses on the o. G. Range of damage prevention go far beyond.

Finally, for the purpose of permanent condition monitoring, the TE probe would have to be permanently installed on the object to be protected. Problems arise in terms of ensuring accessibility for the purpose of battery replacement. In addition, size and weight of the known TE probe do not allow easy attachment to the object. For outdoor objects, the problem of climate protection is added. Finally, in view of selective fault detection, it would be desirable to make measurements at several vulnerable locations within electrical equipment (eg in the transformer, i.e. under oil). For this purpose, the previously known TE probe is not suitable because it can not be hermetically sealed. Due to the required rapid transhipment of parasitic stray capacitances, the wideband amplifiers customary in the prior art for amplifying the high-frequency TE signals (> 100 MHz) consciously require small working resistances (<100 ohms), with switching times in the nanosecond range) and thus large operating currents (approx 10 mA per transistor), which hinders battery operation over a sufficient period of operation.

In classical rectifier circuits (including in avalanche diodes) the threshold is about 2 to 10% of the final control, so that only dynamic range of 1:50 to 1:10 can be achieved. The need for a fixed operating point adjustment in the classical rectifier by means of a voltage divider also increases the power consumption. It is an object of the invention to realize a TE sensor for TE diagnosis including TE anomaly early detection, as well as the accidental protection of high-quality electrical high-voltage equipment. The aim is:

- simplest circuit design,

- low number of components,

- feasibility as integrated circuit,

- low energy consumption,

- low weight and small dimensions,

- possibility of hermetic encapsulation,

high immunity to interference,

- high dynamics of the modulation without measuring range switching,

- simple adaptation options on the test object (if necessary implementation),

Reactivity only to short-time pulses in the nanosecond time range,

- TE signal detection on metal surfaces with a distance of the measuring points of only 10 cm.

The above-mentioned objectives are in part directly linked with each other and complement each other. Based on the above objective, the object is to form the TE signals recorded by means of an adapter adapted to the test object while reducing the noise level and in a processing unit so that the output signals at a higher level and thus reduced risk of further interference of a subsequent signal processing in the control center (eg control room) can be supplied. According to the invention the object is achieved in that the adapter is connected via a signal line and two charge storage capacitors each with the interconnected by Hochohmwiderstand bases of two complementary bipolar transistors whose emitters are each connected to a pole of the operating voltage and their collectors are interconnected and each have a coupling capacitor and in each case one emitter-base path which is complementary to one another in the blocking direction and in each case one charging capacitor are connected to one pole of the operating voltage. In addition, each charging capacitor terminal is preferably connected to the base per mutually complementarily constructed emitter follower, wherein in each case the connection point of the coupling capacitor and the emitter-base path are respectively connected via clamping resistors to the pole of the operating voltage to which the collector of the associated emitter follower is located. Of the two in the emitter branches lying working resistors are amplified by series capacitors, the amplified and stretched TF signals, to an output jack, to which the signal processing stage is connected.

This circuit can be combined with various adapters, which are shown in more detail in the embodiments. As a result, the solution has a wide range of applications.

Instead of a battery power supply or buffering the same, it is also possible to feed via the output signal from the control room a supply current, according to the invention a contact of the output jack is connected via a resistor to the positive and the other contact via a resistor to the negative operating voltage line and at the end the output signal line in the control room chokes a power supply is connected.

To avoid oscillation tendencies and unbalanced control is provided that a low-inductance smoothing capacitor is connected via the connection points of the transistor emitter with the operating voltage potentials. To increase the detection limit for the TE sensor, it is possible according to the invention that between the signal line and the bases of the two complementary bipolar transistors an equally constructed second complementary transistor arrangement with charge storage capacitors is inserted as a precursor. The evaluation in the center is expediently carried out with digital signals, it is expedient that instead of the emitter follower as a voltage follower connected OP amplifiers are arranged, whose outputs are connected to a unipolar pulse train generating arrangement.

The transmission of the TE signals of a high-voltage potential TE sensor to the center is preferably carried out in that the output jack is connected to the modulation input of a downstream light emitter coupled via optical fiber or a focused beam to one of the signal processing stage upstream light receiver with demodulator.

A simple analog signal evaluation and assessment of critical isolation errors provides that in the signal processing stage, a measured value is formed, which corresponds to the mean value of the square of the TE impulse charges. Appropriate capacitive coupling of the signal processing stage requires a Nullpegelklammerung, in which an op-amp with its non-inverting input to reference potential and with its inverting input via a high-pass capacitance at the output of the processing unit via a high-passpole resistor to reference potential via a diode in parallel with a resistor to the output of OP amplifier and are connected to the input of an integrator.

The integration forms a signal level corresponding to the mean square value of the TE impulse charge. In this case, an economic variant provides that the integrator consists of a resistor-capacitor resistor series circuit, wherein the output signal is tapped off across the capacitor, or that an active as an integrator switched OP amplifier instead of or with the resistor-capacitor resistor Row circuit are arranged. The combination of the resistor-capacitor resistor series circuit also prevents, due to their low-pass behavior considered by the output jack, from influencing the TE sensor by external high-frequency interference signals which can be picked up by the signal line from the environment as a result of the antenna effect. In the following embodiment, the invention will be explained in more detail with reference to seven figures. Show it

Fig. 1: the basic principle of the TE sensor technology

2: the adaptation by means of a magnet and the use of the probe housing as an antenna FIG. 3: the adaptation during mechanical fastening and simultaneous electrical contacting of the probe housing on the

FIG. 4: an overall current flow diagram of the TE sensor FIG. 5: a circuit example for analog pulse conditioning FIG. 6: two variants of a pulse transformer FIG. 7: a circuit example for averaging the square of the TE -Impulsladungen.

Fig. 1 shows the basic principle of the partial discharge sensor technique. The partial discharges TE radiated from the various test objects PO can be tested in a variety of ways at the most varied places u. a. also removed at the metallic conductive vessels MG by means of adapters AD, amplified and shaped in the processing unit VE and evaluated after transmission to the central unit SV.

In the simplest case, the adapter AD is realized by the metallic vessel MG of the test object PO for the part itself, the active measuring zone being the area between the metal-tapped tapping points MP ₁ -MP ₂ . The distance MP1-MP2 determines the threshold. It can be reduced to less than 10 cm.

Another possibility of TE signal detection is the single-pole contact with the test object, in which case the opposite pole is realized by the housing of the sensor itself, which acts as a radiator (antenna) for the rapid electromagnetic compensation processes in TE signals. For single-pole contacting, the use of magnetic clamps HM is also expedient for iron vessels MG (FIG. 2).

On the other hand, it is also possible to use an inverse arrangement, i. H. the metallic housing SG of the sensor is electrically and mechanically connected to the test object housing MG, and the opposite pole is by an additional radiator SR (antenna), the z. B. disk-shaped, realized (Fig. 3).

Preferably, for the latter arrangement, a coaxial cable KL is used in conjunction with the adapter to simulate a fictitious short circuit for signals lasting more than a few tens of nanoseconds. Its inner conductor is connected on the input side to the signal line SL and on the output side to the metallic housing SG of the sensor (FIG. 4). The outer conductor of the cable KL is connected on both sides with the sensor housing SG. The optimal length of the cable KL is between 10 and 50 cm. With regard to the universal use, it is expedient to completely integrate this RF cable KL as part of the adapter into the circuit (FIG. 4). In principle, instead of the coaxial cable KL also modified waveguides can be used, such. B. pulse transformer or parallel stripline.

Fig. 6 shows schematically the possibility to develop a coaxial line KL on a Manifer shell core or by a ring core MK and connect as a pulse transformer.

In variant A, the two inputs MP ₁ ; MP _{2 is} connected to the beginning and end of the inner conductor IL _{1 of} the coaxial line KL, while the beginning or end of the outer conductor AL ₁ via the signal line SL to the charge storage capacitors C ₅₁ ; C ₅₂ ) and the end or beginning of the outer conductor AL _{1 is} connected to the ground of the circuit arrangement.

In variant B, on the other hand, the two inputs MP ₁ ; MP _{2 is} connected to the outer conductor AL _{2 of} the beginning and end of the coaxial line KL, while the inner conductor IL _{2 are connected} to the signal line SL and ground. This pulse transformer with a manifold shell core MK and few turns of the coaxial line KL represents a space-saving arrangement, which causes a sharp separation of disturbing pulses.

For signal amplification in the nanosecond time range, concepts are used in the classical circuit technology whose power consumption is due to the required rapid transhipment of parasitic stray capacitances in the range of some 10 mA (operating voltage about 10 V). In addition, considerable expenditure is required to rectify nanosecond pulses of low amplitude (response threshold eg 10 mV) with simultaneously high dynamics (modulation eg up to 5 V) in classical circuit technology, as a result of which the power consumption also increases considerably. An unexpected in the field of nanosecond pulse technology effect with respect to lowest power consumption (<1 mA) and high dynamics is achieved by the inventive arrangement in which a counter circuit of a bipolar transistor T ₂ npn-type with a pnp transistor T ₁ in the Way, that the collectors of both transistors directly and the base of each transistor T ₁ ; T _{2 are} connected via a high-ohmic resistor RB in MOhm range (Fig. 4). The emitter of the pnp transistor T is connected to the positive operating voltage pole (+) and the emitter of the 2n pn transistor T _{2 is} connected to the negative operating voltage pole (-). The input signal line SL is across the charge storage capacitors C ₅₁ and C ₅₂ , whose capacitance is in the range of a few 10 pF, depending on the base of the transistors T ₁ ; T ₂ connected. The connection point of the collectors of both transistors T ₁ , T ₂ is on the one hand connected via a coupling capacitor C _K1 in the range of less than 10 nF with a clamping resistor R _K1 in the order of 50 kOhm, with its other terminal to the negative operating voltage (-) is applied. The connection point of coupling capacitor C _K1 and resistor R _K1 is connected to the emitter of a pnp transistor T ₃ whose base is connected via a charging capacitor CLI of the capacity of the order of 100 pF to the positive operating voltage (+). At the junction of charging capacitor C _L1 and the base of T ₃ , the base of a pnp transistor T _{4 is connected} in collector circuit with the load resistor R _A1 .

In an analogous manner, a second measuring branch is realized, consisting of the corresponding components C _K2 , R _K2 , C _L2 , R _A2 and the npn transistors T ₅ and T ₆ . The values of the resistors R _A1 and R _A2 are of the order of 10 kOhm. The crossover frequency of all transistors is above 500 MHz.

At the connection point of transistor T ₄ and resistor R _A1 and of transistor T ₆ and resistor R _A2 , a series capacitor C _R1 and C _{R2 is} connected in each case, the mating terminals are connected together. At the connection point of the components C _R1 , C _R2 , the coaxial output socket КЗ is connected, on the one hand, the signal propagation to the subsequent signal processing stage SV and possibly a supply of the operating voltage of about 8 to 10 V is ensured by then with a contact via a resistor RR, the size of about 1 kOhm, to the positive (+), and connected to the other contact via the isolation resistor R ₇ to the negative (-) operating voltage line.

At the end of the output signal line connected to the output signal socket КЗ in the control room, the signal processing stage SV is connected via a capacitor and the power supply via a choke. The outer conductor of the output signal line is clamped there to reference potential (ground).

The smoothing capacitor C _G is designed as a ceramic disk capacitor and has a capacity of about 100 nF. It is to avoid oscillation tendencies and unbalanced controls in short-term pulses directly to the connection point of the emitter of transistor T ₁ with the negative operating voltage (-) and connected to the junction of the emitter of transistor T ₂ with the positive operating voltage (+), wherein the lead length on both sides of the capacitor C _{G is} to be limited to less than 2 cm

 The basic mode of operation of the circuit according to FIG. 4 is the following:

By means of the signal adapter AD in conjunction with the coaxial cable KL, the maximum duration of the TE signals detectable on the test sample vessel MG is limited in such a way that the highest measuring sensitivity is achieved with pulse rise times in the nanosecond time range. This purpose is fulfilled in particular by the short-circuit line KL, unless a fictitious short circuit for long-term signals already exists due to the contacting with the metallic vessel MG of the test object. As a result, the useful-interference signal distance can be considerably improved, since usual interference signals have, among other things, rise times in the range of a few 100 to 1000 ns and are thus considerably longer than the above-mentioned value of a few ns for maximum measuring sensitivity.

As a result of the countercurrent connection of the transistors T ₁ and T ₂ with simultaneous connection of the bases of these transistors via a high-resistance resistor (> 1 MOhm), this transistor arrangement is, as it were, in the "quiet" state, ie the power consumption is far below 1 mA. When a short-time pulse of a few ns duration occurs whose voltage amplitude exceeds about 1 mV, with positive polarity of the pulse, a charge fraction is taken over by the storage capacitor C ₅₁ and injected into the base of the transistor T ₁ . After transient charge storage in transistor T ₁ , after the input pulse has almost decayed, a collector current surge in the transistor T _{1 is} triggered. The transistor T ₂ remains in the "snooze" state when the positive polarity is considered, ie no additional current flow through transistor T2 is excited Accordingly, at the point of connection of the collectors of the transistors T ₁ and T ₂ a voltage change occurs which is decisive for the discharge of the transistor storage capacitor C _L1 results given by the current path from the collector of the transistor T ₁ via coupling capacitor C _K1 and a limited avalanche breakdown of the emitter-base path of the transistor T ₃ for charging capacitor C _L1. in contrast, the reverse biased transistor T ₅ remains at the considered According to the current amplification and thus also the charge amplification of transistor T ₁ , a signal amplitude which exceeds the input signal is reached at the charging capacitor C _L1 , above all the duration of the signal at the charging capacitor C _L1 as a result of the termination of the avalanche breakdown and again incipient Locking phase of transistor T _{3 is} greatly stretched. This results in a rapidly rising and almost linearly sloping signal edge, ie, a triangular pulse is obtained, which includes in its area the square value of the pulse charge. This characteristic is advantageously suitable for alarming a critical TE intensity in the sense of anomaly early detection.

Analogous conditions exist with opposite polarity of the input signal. In this case, the transistor T ₂ becomes active and transistor T ₅ suffers the limited avalanche breakdown while transistor T ₃ remains in the off state. When using transistors with a crossover frequency of over 500 MHz, this proportional to the input signal memory effect is detectable into the sub-nanosecond time domain.

It should be noted that this circuit arrangement according to the invention differs substantially from the "classical broadband amplifier", in which the working resistances are intentionally kept small in contrast to the present circuit (magnitude <100 ohms with switching times in the nanosecond range) and also the operating current of each transistor in the range of approx 10 mA.

The inventive peak value storage with transistors T ₃ and T _{5 also} differs from conventional circuit arrangements. Thus, a rectifier effect already takes place at a voltage swing of only 10 mV, which is maintained in the transistors used with a crossover frequency of over 500 MHz, even with switching times of a few ns, ie the reverse recovery time is extremely low. In contrast, in conventional rectifier circuits (including in avalanche diodes), the response threshold rises to 10 to 50 times, so that instead of a dynamics of the modulation of 1: 500 only values of 1:50 to 1:10 at the specified operating voltage of U _{B are} less than or equal to 10 V reachable. The physical effect is based on the avalanche breakdown of the reverse-biased emitter-base path of transistors T ₃ ; T ₅ . It is particularly advantageous that no fixed bias must be set by means of a voltage divider. In contrast, in the case of classic rectifier circuits, an operating point setting is only possible on the one hand by voltage dividers, which increases the power consumption of the circuit.

In addition, this measure is effective only when a diode current is excited, so that the blocking characteristic deteriorates. Due to the simplicity of the circuit arrangement, in particular by the small number of components, the sensor can also be implemented in integrated circuit technology. In addition, hermetic encapsulation is provided, so that the sensor can be installed not only outside but also inside electrical equipment (eg under oil). This results in a further improvement of the signal-to-noise ratio, since, on the one hand, the TE signals no longer have to penetrate the metallic encapsulation with corresponding attenuation and, on the other hand, this encapsulation, as a Faraday cage, considerably attenuates the interference signals in the interior. By mounting several sensors in high-quality equipment (eg transformers), a multi-position measurement can also be advantageous for fault localization. The smallness of the sensor is also beneficial for the pinpoint localization of TE-damaged bar windings of large motors and generators as evidenced by experimental studies. It should be noted that the sensor can of course also be operated with internal battery, with a service life of at least one month can be achieved. This period of time seems sufficient to allow a damaged resource to remain operational for a period of time until a replacement unit is procured. Due to permanent condition monitoring, total failure with all consequential damage can be avoided. If an increase in the detection limit for the TE detection is desired, a circuit expansion by a pre-stage, which is analogous to the input circuit gem. Fig. 4 with the components C ₅₁ , C ₃₂ , R _B , T ₁ and T _{2 is} realized.

The circuit arrangement according to the invention can also be combined with an active broadband antenna as a pickup. As active broadband antenna is preferably the known three-electrode system with differential amplifier arrangement.

For further signal processing in the control center, the digital technology is available. For this purpose, an analog pulse conditioning when using circuit components of the TE sensor is recommended. An embodiment is shown in FIG. 5. This circuit arrangement, in which instead of the emitter follower T ₄ ; T _{6 are used} as a voltage follower switched operational amplifier, generated from bipolar needle pulses unipolar triangular pulses, which allow by means of a threshold trigger in a simple manner, an amplitude-time conversion and thus digitization.

If the entire partial discharge sensor is battery-powered to high-voltage potential, the transmission of the amplified and shaped TE signals to the evaluation devices is preferably to be carried out by means of focused light or laser beam or via high-voltage-resistant optical fiber cable. In this case, the partial discharge sensor according to FIG. 5 is connected with its signal output K1 to a modulation input of a non-illustrated light transmitter, which is likewise at high-voltage potential. Partial discharge sensor and light emitter can then be arranged preferably also in the interior of encapsulated, oil or protective gas-filled electrical equipment at high voltage potential, directly on the possible origin of partial discharges.

Fig. 7 shows a simple analog circuit arrangement for obtaining the mean square value from the unipolar triangular pulses as supplied from the circuit arrangement according to Fig. 5 to the output socket K1. The evaluation of this mean square value is a good and cheap alternative to the digital evaluation and allows a better assessment of a critical insulation error, in contrast to the applicable standards (IEC 270), because large TE pulses are particularly emphasized by the square value and also by averaging a dense Sequence of TE pulses also lead to a high output signal.

The circuit arrangement preferably consists of OP amplifier stages, which are connected via a high-pass filter C ₁ ; R-, to the output jack KI of Fig. 5 are connected. The decoupling via the high-pass capacitor C ₁ prevents lower-frequency noise or operating point displacements of the upstream stage from leading to the displacement of the output signal level and thus to the distortion of the measured value. The time constant of the high-pass C ₁ * R ₁ is approx. 1 ms.

The first stage causes with the diode D ₁ and the output resistance R ₂ in the feedback branch of the OP amplifier VI, a clamping of the zero level, which would otherwise move in response to the pulse train. The second OP stage V2 represents an active integrator, whose integration capacitor C located in the feedback branch, with the series resistance R _{3, has} a time constant C; * R ₃ approx. 10 ms and with its adjustable parallel resistor R ₄ a time constant C | ♦ R _{4 forms} approx. 50 ms.

The signal level thus formed corresponds to the mean square value of the TE pulse charges.

The output side resistor-capacitor resistor series circuit, in which the output jack K2 is connected via the capacitor, is another passive integration member that can take over the function of the active integrator with appropriate dimensioning, so that the second operating stage in a economic variant can be omitted.

In addition, this passive integrator prevents due to the low-pass behavior influencing the TE sensor by external high-frequency noise, which could be coupled due to the antenna effect on the output signal line from the environment in the output jack K2.

Claims (9)

1. partial discharge sensor, in particular for the on-site diagnosis, the anomaly-early detection and the accidental protection of electrical equipment, without their activation, consisting of a respective test object adapted adapter (AD) for partial discharge signals, a processing unit (VE) and a signal processing stage (SV), characterized in that the adapter (AD) via a signal line (SL) and two charge storage capacitors (C _S1 ; C _S2 ) each with the by means of high resistance (R _B ) interconnected bases of two complementary bipolar transistors (T ₁₇ - T ₂ ) is connected such that their emitters are in each case connected to one pole of an operating voltage (+; -) and whose collectors are connected to one another and via a respective coupling capacitor (C _K1 ; C _K2 ), one mutually complementary emitter base polarized in the reverse direction Line (T ₃ ; T ₅ ) and one charging capacitor each (C _L1 , C _L2 ) connected to each pole of the operating voltage In addition, each charging capacitor terminal (C _n ; C _L2 ), is connected to the base per mutually complementary emitter follower (T ₄ , T ₆ ), that in each case the connection point of the coupling capacitor (C _K1 ; C _K2 ) and the emitter-base path (T ₃ ; T ₅ ) respectively is connected via terminal resistance (R _K1 , R _K2 ) to the pole of the operating voltage, at which the collector of the associated emitter follower (T4; T6) is located, and that of the two in the emitter branches working resistances (R _A1 , R _A2 ) each series capacitors (C _R1 , C _R2 ) decouple the amplified and extended TE signals to an output socket (КЗ) to which the signal processing stage (SV) is connected. 2. Partial discharge sensor according to claim 1, characterized in that for supplying a supply current via the output signal line, a contact of the output socket (КЗ) of the processing unit (VE) via a resistor (R _R ) with the positive (+) and the other contact the Output socket (КЗ) via a separation resistor (R ₁ -) to the negative (-) operating voltage line is connected, wherein at the end of the output signal line in the control room next to the signal processing stage (SV) choked a power supply is connected. 3. Partial discharge sensor according to claim 1, characterized in that to avoid oscillation tendencies and unbalanced controls a low-inductance smoothing capacitor (C _G ) via the connection points of the transistor emitter (T ₁ , T ₂ ) with the operating voltage potentials (+, -) is connected. 4. Partial discharge sensor according to claim 1, characterized in that for increasing the TE detection limit between the signal line (SL) and the bases of the two complementary bipolar transistors (T ₁ , T ₂ ) an equally constructed second complementary transistor arrangement with charge storage capacitors is inserted as a precursor , 5. partial discharge sensor according to claim 1, characterized in that for digital evaluation in place of the emitter follower (T ₅ ; T ₆ ) arranged as a voltage follower OP amplifiers are arranged, whose outputs are connected to a threshold trigger. 6. partial discharge sensor according to claim 1 or 5, characterized in that the output socket (K1) is connected to the modulation input of a downstream light emitter which is coupled via optical fiber or a focused beam to one of the signal processing stage (SV) upstream light receiver with demodulator , 7. Partial discharge sensor according to claim 6, characterized in that to form a measured value corresponding to the mean value of the square of the TE impulse charges, an OP amplifier with its non-inverting input (+) to negative operating voltage (U _B -) and its inverting input (-) via a high-pass capacitance ( _C1 ) at the output (KD of the processing unit (VE), via a high-pass resistor (R1) to the negative (-) operating voltage (U _B -), via a diode (D ₁ ) in parallel with a resistor (R ₂ ) connected to the output of the Op-Amp (VD and to the input of an integrator. 8. Partial discharge sensor according to claim 7, characterized in that the integrator a resistor-capacitor resistor series circuit, wherein the output signal is tapped across the capacitor. 9. Partial discharge sensor according to claim 8, characterized in that an active integrator switched Op-amplifier (V2) between the output of the Op-amplifier (VD and the resistor-capacitor resistor series circuit is arranged. For this 7 pages drawings