CN211374870U - Current detector - Google Patents
Current detector Download PDFInfo
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- CN211374870U CN211374870U CN201922050150.4U CN201922050150U CN211374870U CN 211374870 U CN211374870 U CN 211374870U CN 201922050150 U CN201922050150 U CN 201922050150U CN 211374870 U CN211374870 U CN 211374870U
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- current
- power supply
- magnetic field
- feedback current
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- G—PHYSICS
- G01—MEASURING; TESTING
- G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
- G01R15/00—Details of measuring arrangements of the types provided for in groups G01R17/00 - G01R29/00, G01R33/00 - G01R33/26 or G01R35/00
- G01R15/14—Adaptations providing voltage or current isolation, e.g. for high-voltage or high-current networks
- G01R15/20—Adaptations providing voltage or current isolation, e.g. for high-voltage or high-current networks using galvano-magnetic devices, e.g. Hall-effect devices, i.e. measuring a magnetic field via the interaction between a current and a magnetic field, e.g. magneto resistive or Hall effect devices
- G01R15/202—Adaptations providing voltage or current isolation, e.g. for high-voltage or high-current networks using galvano-magnetic devices, e.g. Hall-effect devices, i.e. measuring a magnetic field via the interaction between a current and a magnetic field, e.g. magneto resistive or Hall effect devices using Hall-effect devices
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- G—PHYSICS
- G01—MEASURING; TESTING
- G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
- G01R19/00—Arrangements for measuring currents or voltages or for indicating presence or sign thereof
- G01R19/0092—Arrangements for measuring currents or voltages or for indicating presence or sign thereof measuring current only
Abstract
The utility model provides a current detector. A current sensor (100) is provided with: the magnetic field sensor includes a magnet core (102) that converges a magnetic field generated by conducting a current (Ip) to be detected to a primary conductor (104), a fluxgate sensor (106) that outputs a detection signal corresponding to the intensity of the magnetic field converged in the magnet core (102), and a secondary winding (L1) that conducts a feedback current supplied from a power supply (+ Vcc) and causes the magnet core (102) to generate a magnetic field in a direction opposite to the magnetic field converged in the magnet core (102). In particular, the current sensor (100) is provided with: the magnetic field control circuit is characterized by comprising a control circuit (110) for controlling the feedback current to a value which enables the magnetic field converged by the magnetic core (102) and the magnetic field generated by the secondary winding (L1) to be balanced, and a current conduction circuit for conducting the feedback current to the secondary winding (L1) by using the maximum potential difference of the power supply.
Description
Technical Field
The present invention relates to a servo-type current detector, and more particularly to a current detector of a type that supplies a feedback current to a secondary winding wound around a magnetic core.
Background
As such a current detector, there is known a current sensor in which a current flowing to a feedback coil is controlled by detecting a magnetic field generated in a magnetic core of a magnet by a hall element and by PWM-controlling a switching element by amplifying the detection signal by a digital amplifier. In this conventional technique, a dc power supply (+/-) and a feedback coil are half-bridge connected between two switching elements, and the two switching elements are alternately turned on and off by PWM control, whereby the polarity of a current flowing through the feedback coil can be switched.
The above-mentioned prior art is described in, for example, a publication (JP2014-228418A1) issued by the Japanese patent office. Thus, further details of the prior art can be found in this publication.
The above prior art is excellent in that: by using the digital amplifier, the number of components and power consumption can be reduced compared to an analog amplifier, and the inductance of the feedback coil is also used as an output filter.
On this basis, from the viewpoint of the measurement performance of the current sensor, the measurement range of the current depends purely on the supplied power supply voltage (+ voltage or-voltage) and the resistance value of the feedback loop. Therefore, if the resistance value is strictly fixed, it is considered that maximizing the power supply voltage that can be used when supplying the feedback current directly relates to an increase in the measurement range. However, in the environment in which the current sensor is used, the power supply voltage is often predetermined (given), and it is not practical to increase the current measurement range by a simple voltage increase.
SUMMERY OF THE UTILITY MODEL
An object of the utility model is to provide a practical technique that can increase the current measurement scope.
In order to achieve the above object, the present invention adopts the following solution. Note that the parentheses in the following description are merely for reference, and the present invention is not limited to these.
The utility model relates to a current detector possesses: the current detector includes a magnet core that converges a magnetic field generated by conducting a current to be detected to a primary conductor, a detection element that outputs a detection signal corresponding to the intensity of the magnetic field converged in the magnet core, and a secondary winding that conducts a feedback current supplied from a power supply and generates a magnetic field in a direction opposite to the magnetic field converged in the magnet core, and is characterized by comprising: a control circuit for controlling the feedback current conducted to the secondary winding to a magnitude at which a magnetic field converged by the magnetic core and a magnetic field in the opposite direction are balanced; and a current conduction circuit that conducts the feedback current controlled by the control circuit to the secondary winding using a maximum potential difference of the power supply.
Preferably, in the current detector, the control circuit outputs a differential signal of two systems as a control signal for the current conducting circuit, and the current conducting circuit has a circuit configuration including: the maximum potential difference of the power supply is connected by branching into two systems, including a combination of switching elements of the two systems, and the secondary winding bridges the intermediate points of the combination of switching elements of the two systems, the combination of switching elements of the two systems being: by inputting the differential signals of the two systems, the switching element on the high potential side in one system is turned on and the switching element on the low potential side is turned off, and the switching element on the high potential side in the other system is turned off and the switching element on the low potential side is turned on, or the switching element on the high potential side in one system is turned off and the switching element on the low potential side in the other system is turned on and the switching element on the high potential side in the other system is turned off.
Preferably, in the above current detector, the current-on circuit has a circuit configuration of: when the differential signals of the two systems are ac-outputted, in any combination of the switching elements of the two systems, NPN transistors are arranged in groups on the high potential side and PNP transistors are arranged in groups on the low potential side, or N-channel MOSFETs are arranged in groups on the high potential side and P-channel MOSFETs are arranged in groups on the low potential side.
Preferably, in the current detector, the current detector further includes a detection resistor that converts the feedback current into an output voltage, and the current conduction circuit conducts the feedback current to the detection resistor in a state of being connected in series to the secondary winding.
Preferably, in the above current detector, the current conducting circuit conducts the feedback current at a maximum voltage of the unipolar power supply in a case where a unipolar power supply is used as the power supply, and conducts the feedback current at a maximum voltage between the bipolar power supplies in a case where a bipolar power supply is used as the power supply.
Preferably, in the above current detector, the current conducting circuit conducts the feedback current at a maximum voltage of the unipolar power supply in a case where a unipolar power supply is used as the power supply, and conducts the feedback current at a maximum voltage between the bipolar power supplies in a case where a bipolar power supply is used as the power supply.
The utility model provides a current detector. The current detector of the present invention is a so-called closed-loop, servo-type current detector, which uses a magnet core to converge a magnetic field generated by conduction of a detected current and detects the magnetic field intensity, and controls a feedback current so that the magnetic field converged by the magnet core and the magnetic field in the opposite direction generated by a secondary winding wound around the magnet core are equalized, thereby performing voltage conversion on the feedback current at this time by using a detection resistor and outputting the converted voltage as an output signal (current detection value).
In this case, as described above, the measurement range of the current to be detected depends on the voltage of the power supply and the resistance value of the feedback loop, but the current detector of the present invention is formed as a circuit configuration in which the feedback current is conducted to the secondary winding using the maximum potential difference of the power supply voltage. Therefore, even if the power supply voltage is a predetermined (predetermined) voltage in the use environment of the current detector, the feedback current can be passed by effectively utilizing the voltage of the power supply to the maximum. This effectively increases the current measurement range within a predetermined power supply voltage range.
In the circuit configuration of the above-described conventional technique and the like, the midpoint of the two dc power supplies (+ power supply and — power supply) is set to the ground level, and therefore, the power supply voltage of the feedback current that can flow when each switching element is turned on is only the midpoint of the two poles.
In contrast, the circuit configuration of the present invention is advantageous in that, when the same dc power supply (±) is used, the feedback current can be passed to the secondary winding by using the maximum potential difference between the positive and negative poles exceeding the midpoint between the two poles. Further, even when the power supply is unipolar (+ Vcc), the feedback current can flow at the maximum voltage of 0V to + Vcc. Thus, the predetermined power supply voltage can be effectively used to the maximum extent in the use environment, and the measurement range of the current to be detected can be increased.
As described above, according to the present invention, the current measurement range can be increased.
Drawings
Fig. 1 is a block diagram schematically showing a configuration of a current sensor according to an embodiment.
Fig. 2 is a circuit diagram showing the configuration of the bridge circuit in detail.
Fig. 3A and 3B are diagrams showing an operation example when a feedback current is conducted to the secondary winding through the bridge circuit.
Fig. 4 is a diagram showing a circuit configuration of a half-bridge circuit as a comparative example.
Fig. 5A and 5B are diagrams showing an operation example of a half-bridge circuit of a comparative example.
Fig. 6 is a block diagram showing a configuration of a current sensor according to an application example.
Fig. 7A to 7D are graphs showing temporal changes in the feedback current obtained in the present embodiment and the comparative example, in which the power supply voltage is set to the unipolar power supply (+ Vcc).
Fig. 8A to 8D are graphs showing temporal changes in feedback current obtained in an application example in which the power supply voltage is set to the bipolar power supply (± Vcc) and in a comparative example.
Detailed Description
The present embodiment will be described below with reference to the drawings. In the following embodiments, a fluxgate-type current sensor is exemplified as an example of the current detector, but the present invention is not limited thereto, and a hall IC-type current sensor may be used.
Fig. 1 is a block diagram schematically showing a configuration of a current sensor 100 according to an embodiment. The main components of the current sensor 100 include a magnet core 102, a fluxgate sensor 106, a secondary winding L1, a control circuit 110, a bridge circuit 120, a detection resistor Rs, and an output circuit 130. In fig. 1, only main components are illustrated, and other components are appropriately omitted.
[ magnetic core ]
For example, the magnet core 102 has a C-ring shape, and an air gap 102a is formed in a part thereof. The magnet core 102 passes the primary conductor 104 through the inside thereof, thereby converging the magnetic field (magnetic path) generated by the detected current Ip (±). Alternatively, by conducting the detected current Ip to the primary conductor 104, a (convergent) magnetic field is generated in the magnet core 102.
[ detecting element ]
The fluxgate sensor 106 is disposed in the air gap 102a of the magnet core 102. The fluxgate sensor 106 has a probe coil 106c wound around the fluxgate core 106a, and outputs a detection current corresponding to the intensity of the magnetic field generated (converged) on the magnet core 102.
[ Secondary winding ]
The secondary winding L1 is wound around a part of the magnet core 102 in the circumferential direction, and the secondary winding L1 turns on the feedback current (±) to generate a magnetic field in the opposite direction (direction to cancel out) the magnetic field generated by turning on the detected current Ip (±). The winding direction and the number of turns N of the secondary winding L1 can be set as appropriate according to the use conditions of the current sensor 100 and the like.
[ control Circuit ]
The control circuit 110 controls the magnitude and direction (polarity) of the feedback current flowing to the secondary winding L1. That is, when the current sensor 100 is used, the control circuit 110 outputs a feedback current to the secondary winding L1 to generate a magnetic field in the opposite direction when the magnetic field is generated around the primary conductor 104 by conducting the detected current Ip (±) and the magnetic field is converged by the magnet core 102, and controls the feedback current to a magnitude at which the detection signal (detection current) output from the probe coil 106c disappears. The disappearance of the detection signal from the probe coil 106c means that the magnetic field generated by the conduction of the detection current Ip (±) is balanced (equalized) with the magnetic field in the opposite direction generated by the secondary winding L1. When the polarity of the detected current Ip changes periodically, the polarity of the feedback current changes periodically in accordance with the change.
[ Current conduction Circuit ]
[ detection resistance, output Circuit ]
The sense resistor Rs is connected in series with the secondary winding L1 in the conduction direction of the feedback current. The detection resistor Rs converts the voltage of the feedback current, and the output circuit 130 performs signal output processing on the feedback current to obtain an output voltage Vout. Thus, the waveform (±) of the output voltage Vout substantially matches the waveform of the detected current Ip (±), and therefore, substantially becomes a value related to the magnitude of the detected current Ip (±).
[ details of bridge circuits ]
Fig. 2 is a circuit diagram showing the configuration of the bridge circuit 120 in detail. Here, the secondary winding L1 and the detection resistor Rs are shown embedded in the circuit.
The bridge circuit 120 branches the power supply (+ Vcc) and the Ground (GND) into two systems for connection, one of which includes a combination of two transistors Q3, Q2 (switching elements), and the other of which includes a combination of the remaining two transistors Q4, Q1 (switching elements). In addition, the signal a is input as a common base current to the combination of the transistors Q3 and Q2 of the one system, and the signal B is input as a common base current to the combination of the transistors Q4 and Q1 of the other system.
Fig. 2 shows an example in which a unipolar power supply (+ Vcc) is used, but bipolar power supplies (+ Vcc and-Vcc) may be used. In this case, as shown by the bracket in fig. 2, the Ground (GND) is at a potential of a negative power supply (-Vcc) (the same applies hereinafter).
The bridge circuit 120 is configured as a circuit in which a combination of transistors Q3 and Q2 and a combination of transistors Q4 and Q1, which are disposed so as to be branched into two systems, are bridged to each other at an intermediate point by a secondary winding L1 and a detection resistor Rs (so-called H full bridge). In each system, the transistors Q3 and Q4 located on the high potential side are NPN type, and the transistors Q1 and Q2 located on the low potential side are PNP type.
[ working examples ]
Fig. 3A and 3B are diagrams showing an operation example when the feedback current is conducted to the secondary winding L1 through the bridge circuit 120. An operation example of the current sensor 100 will be described below.
[ step 1]
For example, when the detected current Ip (+) is conducted to the primary conductor 104 shown in fig. 1, the generated magnetic field is converged by the magnet core 102.
[ step 2]
The magnetic flux in the fluxgate sensor 106 (probe coil 106c) disposed in the air gap 102a is interleaved by the magnetic field generated (converged) in the magnet core 102.
[ step 3]
The control circuit 110 outputs a signal a and a signal B for canceling the magnetic flux crossing the fluxgate sensor 106 (probe coil 106 c). The polarity of the signal A, B at this time is determined according to the polarity of the detected current Ip. As described above, the signal A, B is a differential signal, and when one is a positive signal, the other is a negative signal.
[ step 4]
FIG. 3A: in this example, since the signal B is positive (+), and the signal a is negative (-), the NPN transistor Q4 and the PNP transistor Q2 in the bridge circuit 120 are turned on. That is, the transistor Q2 on the low potential side in one system is turned on, and the transistor Q4 on the high potential side in the other system is turned on. In addition, the other transistors Q3 and Q1 remain off.
[ step 5]
Accordingly, a feedback current in the forward direction flows through the secondary winding L1 and the detection resistor Rs along the path indicated by the thick arrow in fig. 3A due to the maximum potential difference between the power supply voltage (+ Vcc) and the Ground (GND).
[ step 6]
At this time, the magnitude (current value) of the feedback current flowing is controlled by the control circuit 110. That is, the control circuit 110 determines that the signal A, B of the feedback current of the magnitude necessary to cancel the magnetic field crossing the fluxgate sensor 106 by the detected current Ip is passed by the magnetic field in the reverse direction generated by the secondary winding L1, and controls the base currents flowing to the transistor Q4 and the transistor Q2. Thus, the magnitude of the feedback current in the forward direction is controlled, and the detection resistance Rs converts the feedback current into a detection voltage and takes the detection voltage as the output voltage Vout (+).
[ step 7]
FIG. 3B: in the case where the polarity of the detected current Ip is reversed (-), the above is reversed. That is, the bridge circuit 120 receives a negative signal (-) as the signal B and receives a positive signal (+) as the signal a. In this case, the NPN transistor Q3 and the PNP transistor Q1 are turned on, and the transistor Q4 and the transistor Q2 are kept off this time. That is, the transistor Q3 on the high potential side in one system is turned on, and the transistor Q1 on the low potential side in the other system is turned on.
[ step 8]
Accordingly, the negative feedback current flows through the secondary winding L1 and the detection resistor Rs along the path indicated by the thick arrow in fig. 3B due to the maximum potential difference between the power supply voltage (+ Vcc) and the Ground (GND).
[ step 9]
Here, the magnitude (current value) of the feedback current flowing is also controlled by the control circuit 110, and the control circuit 110 determines that the signal A, B of the feedback current of the magnitude necessary to cancel the magnetic field crossing the fluxgate sensor 106 by the detected current Ip is flowing by the magnetic field in the opposite direction generated by the secondary winding L1, thereby controlling the base currents flowing to the transistor Q3 and the transistor Q1. Thus, the magnitude of the feedback current in the negative direction is controlled, and the detection resistance Rs converts the feedback current into a detection voltage and takes the detection voltage as the output voltage Vout (-).
Next, the current sensor 100 of the present embodiment will be described based on comparison with a comparative example.
Fig. 4 is a diagram showing a circuit configuration of a half-bridge circuit 520 as a comparative example. The half-bridge circuit 520 of the comparative example is a so-called half-bridge circuit in which a power supply (+ Vcc) and a Ground (GND) are connected by a combination of transistors Q1, Q2 of one system, and a secondary winding L1 is connected at the midpoint thereof. A detection resistor Rs is connected to the tip of the secondary winding L1, and the tip of the detection resistor Rs is grounded via a current source V2 of an intermediate potential (+ Vcc/2). Further, as shown by brackets (-Vcc) in fig. 4, a bipolar power supply may also be used in the comparative example. The transistors Q1 and Q2 are driven by a single system signal.
Fig. 5A and 5B are diagrams illustrating an operation example of a half-bridge circuit 520 according to a comparative example.
FIG. 5A: for example, when a + signal is input in accordance with the polarity of the detected current Ip, the transistor Q2 is turned on, and the transistor Q1 is kept off. This causes a feedback current in the forward direction to flow through the secondary winding L1 along a path indicated by a thick arrow line. The maximum value of the feedback current that can flow at this time depends on the intermediate potential (+ Vcc/2) of the power supply voltage, not the maximum potential difference (+ Vcc) of the power supply voltage.
FIG. 5B: when a signal is input when the polarity of the detected current Ip is inverted, the transistor Q1 is turned on, and the transistor Q2 is turned off. Thus, the feedback current in the negative direction flows to the secondary winding L1 along the path shown by the thick arrow line, but the maximum value of the feedback current that can flow is still determined by the intermediate potential (-Vcc/2) of the power supply voltage, not the maximum potential difference (-Vcc) of the power supply voltage.
[ calculation examples of comparative examples ]
Here, an attempt was made to verify the half-bridge circuit 520 of the comparative example using specific values. The following numerical values are appropriately selected values, and are merely examples.
The calculation conditions are as follows.
Power supply voltage: +12V
Resistance of secondary winding L1: 30 omega
Detection resistance Rs: 2 omega
The maximum value of the feedback current (±) is calculated based on the above calculation conditions as follows.
Maximum value (power supply voltage 12V × (1/2))/(total resistance value of feedback loop 30 Ω +2 Ω) — 0.1875A
Further, there is actually a voltage drop in the transistors Q1, Q2, etc., but it is ignored here.
It can be seen that the maximum value of the feedback current depends on the power supply voltage and the total resistance of the feedback loop, but only half of the 12V power supply voltage can be used in the half-bridge circuit 520 of the comparative example. The low available power supply voltage means that the maximum value of the feedback current flowing is suppressed to be low, and therefore means that the range that can be measured by the current sensor 100 is limited accordingly.
Assuming that the half-bridge circuit 520 is changed to the bipolar power supply (+ -12), the maximum voltage of 12V can be used, but under the present circumstances that the usage environment (system) of the current sensor 100 is simplified, the use of the bipolar power supply is considered to be less and less, and from the viewpoint of convenience, the demand for a unipolar power supply as the power supply voltage will be increased in the future.
In view of the above, the present embodiment has been disclosed as a preferred example of a circuit configuration that can make the most effective use of a power supply voltage even if the power supply voltage is a predetermined (predetermined) power supply voltage depending on the usage environment without simply increasing the available power supply voltage.
[ example of calculation in the present embodiment ]
Hereinafter, an example of calculation when the bridge circuit 120 of the present embodiment is used will be described. Here, it is assumed that the calculation conditions are the same as those of the comparative example.
Power supply voltage: +12V
Resistance of secondary winding L1: 30 omega
Detection resistance Rs: 2 omega
In the case of the present embodiment, the maximum value of the feedback current (±) can be calculated as follows.
Maximum value of (power supply voltage 12V)/(total resistance value of feedback loop 30 Ω +2 Ω) is 0.375A
In addition, there is a voltage drop in the transistors Q1 to Q4 and the like in practice, but it is ignored here.
[ test results ]
As is clear from the above calculation examples, when the bridge circuit 120 of the present embodiment is used, the power supply voltage (+12V) can be used under a condition close to the maximum value thereof, as compared with the half bridge circuit 520 of the comparative example. Thus, a current about 2 times as large as that of the half-bridge circuit 520 of the comparative example can be passed through the feedback loop. Therefore, the measurement range of the current sensor 100 can be increased by a unique circuit configuration without simply increasing the power supply voltage itself.
[ application example ]
Fig. 6 is a block diagram showing a configuration of a current sensor 200 according to an application example. The difference from the current sensor 100 according to the embodiment is that the power supply voltage is bipolar (+ Vcc and-Vcc), and the other configuration is the same as that of the embodiment.
The operation of the current sensor 200 according to the application example is as follows. With appropriate reference to figure 3.
[ step 1A ]
For example, when the detected current Ip (+) is conducted to the primary conductor 104 shown in fig. 6, the generated magnetic field is converged by the magnet core 102.
[ step 2A ]
The magnetic flux in the fluxgate sensor 106 (probe coil 106c) disposed in the air gap 102 is interleaved by the magnetic field generated (converged) in the magnet core 102.
[ step 3A ]
The control circuit 110 outputs a signal a and a signal B for canceling the magnetic flux crossing the fluxgate sensor 106 (probe coil 106 c). The polarity of the signal A, B at this time is determined according to the polarity of the detected current Ip. As described above, the signal A, B is a differential signal, and when one is a positive signal, the other is a negative signal.
[ step 4A ]
FIG. 3A: as in the first embodiment, the signal B is positive (+), and the signal a is negative (-), so that the NPN transistor Q4 and the PNP transistor Q2 in the bridge circuit 120 are turned on. In addition, the other transistors Q3 and Q1 remain off.
[ step 5A ]
Accordingly, a feedback current in the forward direction flows through the secondary winding L1 and the detection resistor Rs along the path indicated by the thick arrow in fig. 3A due to the maximum potential difference (± Vcc) between the bipolar power sources.
[ step 6A ]
The magnitude (current value) of the feedback current flowing at this time is controlled by the control circuit 110, and the control circuit 110 determines that the signal A, B of the feedback current of the magnitude necessary to cancel the magnetic field crossing the fluxgate sensor 106 by the detected current Ip is flowing by the magnetic field in the opposite direction generated by the secondary winding L1, thereby controlling the base currents flowing to the transistor Q4 and the transistor Q2. Thus, the magnitude of the feedback current in the forward direction is controlled, and the detection resistance Rs converts the feedback current into a detection voltage and takes the detection voltage as the output voltage Vout (+).
[ step 7A ]
FIG. 3B: when the polarity of the detected current Ip is inverted (-) as described above, the bridge circuit 120 receives the negative signal (-) as the signal B and receives the positive signal (+) as the signal a. In this case, the NPN transistor Q3 and the PNP transistor Q1 are turned on, and the transistor Q4 and the transistor Q2 are kept off this time.
[ step 8A ]
Accordingly, the negative feedback current flows through the secondary winding L1 and the detection resistor Rs along the path indicated by the thick arrow in fig. 3B due to the maximum potential difference (± Vcc) between the bipolar power sources.
[ step 9A ]
Here, the magnitude (current value) of the feedback current flowing is also controlled by the control circuit 110, and the control circuit 110 determines that the signal A, B of the feedback current of the magnitude necessary to cancel the magnetic field crossing the fluxgate sensor 106 by the detected current Ip is flowing by the magnetic field in the opposite direction generated by the secondary winding L1, thereby controlling the base currents flowing to the transistor Q3 and the transistor Q1. Thus, the magnitude of the feedback current in the negative direction is controlled, and the detection resistance Rs converts the feedback current into a detection voltage and takes the detection voltage as the output voltage Vout (-).
[ Effect of application example ]
In the case of the current sensor 200 of the application example, the following advantages are also provided.
(1) Even if the usage environment is a bipolar power supply, the feedback current can be made to flow by the maximum potential difference (± Vcc) between them, and therefore, the voltage of each power supply (+ Vcc and-Vcc) may be 1/2 in one embodiment, as long as the measurement range is set to be approximately the same as that in one embodiment. For example, in the above calculation example, it is sufficient to use +6V and-6V as the bipolar power source, and the increase in the system can be suppressed accordingly.
(2) Alternatively, when the voltages of the power supplies (+ Vcc and-Vcc) are made the same as those of the one embodiment, the available voltage is 2 times that of the one embodiment. Therefore, the maximum value of the feedback current can be increased by about 2 times, and the measurement range can be further increased.
[ test result list ]
The advantages of the current sensor 100 according to the present embodiment and the bridge circuit 120 used in the application example thereof have been described above by comparing with the half-bridge circuit 520 of the comparative example. The advantages of the embodiment and its application examples described above will be described in the following.
[ verification based on comparison of the present embodiment with comparative example ]
Fig. 7A to 7D are graphs showing temporal changes in the feedback current obtained in the present embodiment and the comparative example, in which the power supply voltage is set to the unipolar power supply (+ Vcc). Fig. 7A and 7C are waveforms of the feedback current obtained in the present embodiment, and fig. 7B and 7D are waveforms of the feedback current obtained in the comparative example. In addition, the waveforms were all obtained by simulation.
[ control time using AC signal ]
As is clear from a comparison of the waveform of fig. 7A and the waveform of fig. 7B, although the feedback current also has an alternating waveform when the control signal (signal A, B in the present embodiment) is alternating, the maximum value Ifb1(±) of the feedback current is obtained at a sufficiently high wave height in the present embodiment, whereas the maximum value Ifb2(±) is obtained only at a low wave height in the comparative example. The wave height of the present embodiment is about 2 times that of the comparative example. This means that the maximum value of the feedback current of about 2 times that of the comparative example is obtained in the present embodiment, and the result is consistent with the above description.
[ control time using DC signal ]
As is clear from a comparison between the waveform of fig. 7C and the waveform of fig. 7D, although the feedback current is also dc when the control signal (signal A, B in the present embodiment) is dc, the maximum value Ifb1(+) of the feedback current is obtained at a sufficiently high level in the present embodiment, whereas the maximum value Ifb2(+) is obtained only at a low level in the comparative example. The current value of the present embodiment is about 2 times that of the comparative example. This means that the maximum value of the feedback current of about 2 times that of the comparative example is obtained in the present embodiment, and the same is true as described above.
[ verification based on comparison of application example with comparative example ]
Next, fig. 8A to 8D are graphs showing temporal changes in the feedback current obtained in the application example and the comparative example in which the power supply voltage is set to the bipolar power supply (± Vcc). Fig. 8A and 8C are waveforms of the feedback current obtained in the application example, and fig. 8B and 8D are waveforms of the feedback current obtained in the comparative example. In addition, the waveforms were all obtained by simulation.
[ control time using AC signal ]
As is clear from a comparison of the waveform of fig. 8A and the waveform of fig. 8B, although the feedback current also has an alternating current waveform when the control signal (signal A, B in the application example) is alternating current, the maximum value Ifb3(±) of the feedback current is obtained at a sufficiently high wave height in the application example, whereas the maximum value Ifb4(±) is obtained only at a low wave height in the comparative example. Here, the wave height of the application example is about 2 times that of the comparative example. This means that the maximum value of the feedback current of about 2 times that of the comparative example was obtained in the application example, and still agrees with the above description.
[ control time using DC signal ]
As is clear from a comparison of the waveform of fig. 8C and the waveform of fig. 8D, the feedback current is also dc when the control signal (signal A, B in the application example) is dc, but the maximum value Ifb3 of the feedback current is obtained at a sufficiently high level in the application example, whereas the maximum value Ifb4 is obtained only at a low level in the comparative example. The current value of the application example is about 2 times that of the comparative example. This means that the maximum value of the feedback current of about 2 times that of the comparative example is obtained in the application example, and the description is also consistent with the above description.
As described above, the current sensor 100 according to the present embodiment and the current sensor 200 according to the application example have the following effects.
(1) The measurement range can be increased only by the configuration of the bridge circuit 120 without changing a predetermined (predetermined) power supply voltage in a use environment.
(2) Under the present situation of the system unipolar electric power generation in the use environment, the requirements of simplification of the system, the good degree of convenience and the like can be accurately satisfied.
(3) As in the application example, even a bipolar power supply can be used, and in this case, each power supply voltage can be set lower (half) than before.
The present invention is not limited to the above embodiments and application examples, and can be implemented by being variously modified. For example, the configuration of the bridge circuit 120 can be modified as follows.
(1) MOSFETs may be used instead of the bipolar transistors Q1 to Q4. In this case, the NPN transistors Q3 and Q4 may be N-channel MOSFETs, and the PNP transistors Q1 and Q2 may be P-channel MOSFETs. In addition, the MOSFET may use the control signal as a gate voltage and drive at its enhancement region.
(2) In addition, the bipolar transistor and the MOSFET may be combined and arranged in the same H-bridge. In this case, the combination mode is arbitrary, and one may be a bipolar transistor and the other three may be MOSFETs, or two may be bipolar transistors and the other two may be MOSFETs, or three may be bipolar transistors and the other may be MOSFETs.
(3) The control signal (the signal A, B) may be input in combination of two transistors, or may be input to each of the transistors Q1 to Q4.
(4) In addition, when the combination of the bipolar transistor and the MOSFET used in the H-bridge is modified as in (2), the input method of the control signal can be appropriately changed according to the combination of the elements at that time.
Note that the configurations shown in the drawings in the embodiments and application examples are merely preferable examples, and it is needless to say that the present invention can be appropriately implemented by adding various elements to the basic configuration or by replacing some of them.
Claims (6)
1. A current detector is provided with: a magnetic core for converging a magnetic field generated by conducting a current to be detected to a primary conductor, a detection element for outputting a detection signal corresponding to the intensity of the magnetic field converged in the magnetic core, and a secondary winding for conducting a feedback current supplied from a power supply and generating a magnetic field in a direction opposite to the magnetic field converged in the magnetic core,
the current detector is characterized by comprising:
a control circuit for controlling the feedback current conducted to the secondary winding to a magnitude at which a magnetic field converged by the magnetic core and a magnetic field in the opposite direction are balanced; and
and a current conduction circuit for conducting the feedback current controlled by the control circuit to the secondary winding using a maximum potential difference of the power supply.
2. The current detector of claim 1,
the control circuit outputs differential signals of two systems as control signals for the current conducting circuit,
the current conducting circuit comprises the following circuits:
the connection is made by branching the maximum potential difference of the power supply into two systems, including a combination of switching elements of the two systems, and the intermediate points of the combination of the switching elements of the two systems are bridged by the secondary winding,
the combination of the switching elements of the two systems mentioned above means: by inputting the differential signals of the two systems, the switching element on the high potential side in one system is turned on and the switching element on the low potential side is turned off, and the switching element on the high potential side in the other system is turned off and the switching element on the low potential side is turned on, or the switching element on the high potential side in one system is turned off and the switching element on the low potential side in the other system is turned on and the switching element on the high potential side in the other system is turned off.
3. The current detector of claim 2,
the current conducting circuit comprises the following circuits:
when the differential signals of the two systems are ac-outputted, in any combination of the switching elements of the two systems, NPN transistors are arranged in groups on the high potential side and PNP transistors are arranged in groups on the low potential side, or N-channel MOSFETs are arranged in groups on the high potential side and P-channel MOSFETs are arranged in groups on the low potential side.
4. Current detector according to any of claims 1 to 3,
the current detector is also provided with a detection resistor for converting the feedback current into an output voltage,
the current conduction circuit conducts the feedback current to the detection resistor in a state of being connected in series with the secondary winding.
5. Current detector according to any of claims 1 to 3,
the current conducting circuit conducts the feedback current at a maximum voltage of the unipolar power supply in a case where a unipolar power supply is used as the power supply, and conducts the feedback current at a maximum voltage between the bipolar power supplies in a case where a bipolar power supply is used as the power supply.
6. The current detector of claim 4,
the current conducting circuit conducts the feedback current at a maximum voltage of the unipolar power supply in a case where a unipolar power supply is used as the power supply, and conducts the feedback current at a maximum voltage between the bipolar power supplies in a case where a bipolar power supply is used as the power supply.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP2018223095A JP2020085753A (en) | 2018-11-29 | 2018-11-29 | Current detector |
| JP2018-223095 | 2018-11-29 |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CN211374870U true CN211374870U (en) | 2020-08-28 |
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| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN201922050150.4U Expired - Fee Related CN211374870U (en) | 2018-11-29 | 2019-11-22 | Current detector |
| CN201911159060.7A Pending CN111239462A (en) | 2018-11-29 | 2019-11-22 | Current detector |
Family Applications After (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN201911159060.7A Pending CN111239462A (en) | 2018-11-29 | 2019-11-22 | Current detector |
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| JP (1) | JP2020085753A (en) |
| CN (2) | CN211374870U (en) |
Family Cites Families (9)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| FR2703467B1 (en) * | 1993-03-29 | 1995-06-30 | Mecagis | Zero flow Hall effect current sensor intended in particular for motor vehicles and electric scooters. |
| CH689465A5 (en) * | 1993-11-02 | 1999-04-30 | Lem Liaisons Electron Mec | Bidirectional electric current sensor for magnetic circuit in electric vehicle |
| DE4423429A1 (en) * | 1994-07-05 | 1996-01-11 | Vacuumschmelze Gmbh | Current sensor based on the compensation principle |
| JP2002228689A (en) * | 2001-01-30 | 2002-08-14 | Denso Corp | Magnetic balance type current sensor |
| TWI339008B (en) * | 2007-12-05 | 2011-03-11 | Ite Tech Inc | Class-d amplifier and multi-level output signal generated method thereof |
| JP4769883B2 (en) * | 2009-03-10 | 2011-09-07 | 株式会社ユー・アール・ディー | DC current sensor |
| JP6499821B2 (en) * | 2013-05-23 | 2019-04-10 | 株式会社タムラ製作所 | Current sensor |
| JP6711764B2 (en) * | 2014-06-18 | 2020-06-17 | デンツプライ シロナ インコーポレーテッド | 2-wire ultrasonic magnetostrictive driver |
| JP6520896B2 (en) * | 2016-11-16 | 2019-05-29 | Tdk株式会社 | Inductance element for magnetic sensor and magnetic sensor comprising the same |
-
2018
- 2018-11-29 JP JP2018223095A patent/JP2020085753A/en active Pending
-
2019
- 2019-11-22 CN CN201922050150.4U patent/CN211374870U/en not_active Expired - Fee Related
- 2019-11-22 CN CN201911159060.7A patent/CN111239462A/en active Pending
Also Published As
| Publication number | Publication date |
|---|---|
| CN111239462A (en) | 2020-06-05 |
| JP2020085753A (en) | 2020-06-04 |
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