CN1354905A - Narrow band-pass tuned resonator filter topologies having high selectivity, low insertion loss and improved out-of band rejection over extended frequency ranges - Google Patents

Abstract

A tuned resonator circuit topology is disclosed that permits implementation of narrow band-pass filters having high loaded Q and optimal coupling(for low insertion loss)using a parallel tuned resonator topology at freqencies in the 1 to 2 GHz range and beyond. The topology consists of a mirror image of the parallel tuned circuit about the signal line of a conventional parallel tuned circuit to effect a cancellation of virtually all of the induced currents between the inductive elements of the resonators. This reduction in induced currents reduces the magnetic coupling between the resonators, thereby offsetting the increase in overall coupling between the resonators as frequency increases, and thereby serves to maintain optimal coupling between the resonators as the frequency of operation increases. Moreover, the mirror image topology increases the parallelism between the inductive elements in the resonators, thereby decreasing the inductance values and permitting an increase in capacitance values. Increasing the capacitancevalues of the resonators effectively offsets the decrease in the loaded Q as frequency is increased. The topology works for any number of parallel resonators. As the resolution of the manufacturing process decreases(e.g. from printed circuit board to integrated circuit processes), the range of operating frequencies are scaled with the increase in resolution.

Description

The tuning resonator filter topology in arrowband that on extending bandwidth, has high selectivity, low insertion loss and improved no band resistance

Background of invention

1. technical field

The present invention relates to be applicable to the tuning resonator filter topology in advantageously used arrowband on high frequency (HF), very high frequency(VHF) (VHF), hyperfrequency (UHF) and the microwave section, more particularly, relate to and can on interested frequency range, keep an optionally high capacity Q who is suitable for increasing, minimize the Best Coupling of inserting loss (insertion loss) with improved no band resistance, and with the simple and cheap relatively the sort of topology of repeatable accuracy manufacturing highly.

Background technology

A kind of strict and harsh especially context about the signal processing circuit such as filter appears in the broad band multicarrier Signal Processing.For example have the base-band television signal that is equivalent to about 5-6MHz bandwidth and generally mix (modulation) with RF (radio frequency) carrier signal, thus place it in 50 to 100MHz or bigger scope in the RF channel on, to realize frequency division multiplex (FDM).Such as other such application of microwave communication can need 1-2GHz and outside the opereating specification of (beyond).Need to handle simultaneously the application of the broadband signal that comprises a plurality of channels, such as the transmission of television broadcasting and reception (by aerial or via optical fiber/coaxial cable), needing can occur filter only (promptly to fall into those frequencies in the narrow passband by the sub-fraction of total bandwidth, generally be an interested channel) situation, suppress remaining frequencies on the total bandwidth (promptly fall into stopband in those frequencies) simultaneously.This general narrow-band pass filter that uses is realized.According to being used for the concrete system design of implementing, these filters may need to be operated on the RF frequency identical with interested RF channel, perhaps may convert at the RF channel with rising or descend, be that typical certain other frequency (intermediate frequency or IF) is located to wideband flexible system (agile system).

May inject or produce noise and picture signal and various tedious parasitic signal at each point of handling, therefore often according to the sensitivity of application, calling band pass filter, to suppress (decay) out of band signal be tangible low level.For example, even be attenuated the signal that reaches 60dB and still can in the video transmission of being received, see.Therefore, fully decay occur any not on desirable carrier wave base band signal modulated be very important.This needs band pass filter to have very high selectivity (promptly often, ideally, the sub-fraction of the total bandwidth by comprising interested baseband signal only), in passband, have few or (promptly without any energy loss, low insertion loss), and to other frequency in the stopband keep necessary attenuation amplitude.And because in broadband application, be so little (being approximately 1-2%) comparatively speaking by the fraction of the occupied total bandwidth of baseband signal, so such filter must produce necessary frequency response with high accuracy, and must keep this response (i.e. this response should not drifted about) in later time.In addition, they must be impregnable comparatively speaking for the RF noise from the coupling between outside source and themselves each element.At last, always wish that this filter is inexpensive and easily with high repeatable accuracy manufacturing.

The known technology that several realization band pass filters are arranged.As previously described, its selectivity of the Q value of filter indication; The selectivity of filter is transformed into stopband by the response of filter from passband to be had and how soon limits.The Q value of filter is high more, and then the frequency response from the band connection frequency to the stop-band frequency is fallen steep more.Because its Q value of the input and output load effect of filter, more useful and practical tolerance are Q (that is Q, of its " in circuit " or load _L).The Q of filter _LApproximate the inverse of the fraction bandwidth (fractionalbandwidth) of its frequency response greatly, this generally is lower than on response curve between those points (that is the half-power point of this response) of the peak value 3dB of this response measures.Therefore, the Q of the filter by 1% fraction bandwidth _LBe approximately 100.The narrow-band pass filter that is used for the The Wideband Signal Processing application needs high Q often _LValue be inserted into (insertion) loss (that is, the signal amplitude in the passband of should not decaying significantly) and embody again simultaneously, and signal attenuation should be expired application requirements in stopband.

Be used to realize that a kind of known technology of band pass filter involves (lumped) the LC element that uses lump so that generate typical filter based on low pass to the logical converter technique of band.Can synthesize several different topologys and generate desirable Bandpass Filters response.For the broadband signal purpose of handling in VHF and the UHF frequency band, the shortcoming of this filter is a lot, and the element (coil inductance especially) of lump that the most serious is is in the influence that very easily is subjected to ghost effect far above the frequency place of 100MHz.And what circuit element must be cascaded, to obtain high Q _LThe complexity that is worth needed transfer function.Therefore, such filter has occupied valuable interval, makes manufacturing cost high relatively.

The another kind of known technology of realizing filter adopts helical resonator.Adopt the filter of helical resonator to be coupled, and can produce a lot of The Wideband Signal Processing and use required high Q by magnetic and/or electric capacity ground _LResponse with low insertion loss.But they are not suitable for the frequency well below 150MHz, and reason is the resonator that the super large inductance value can need to be lower than the sort of frequency.This inductor is impracticable maybe can not to construct.Even under higher frequency, they also have sizable mechanical structure (they need protection correct operation and reduction be subject to the character of RF noise effect), this point make make their more expensive comparatively speaking (even in enormous quantities).They are subject to ambient vibration and the influence that floats very much, and make handle in they generally need adjusted value with guarantee they accurately resonance on correct frequency.

Make up band pass filter and also have another kind of known technology, adopt the coaxial transmission line that not only can be used as cylindricality, but also can be used as the dielectric resonator that magnetic that the strip transmission line that is clipped in two printings between the ground shield realizes and/or electric capacity ground are coupled.These resonators are short-circuited transmission lines, thereby develop their performance, make its resonance on as their the concrete frequency of function of length with respect to the wavelength (for the wavelength X of resonance frequency, the length of this line is generally λ/4) of institute's input signals transmitted.This resonator can produce high Q _LValue with the response that obtains to have a lot of The Wideband Signal Processing and use required fraction bandwidth characteristic (that is, 1-2%).But because trace (trace) length increases when desirable resonance frequency reduces, such resonator is not suitable for any non-UHF (that is, between about 400MHz and several GHz).Because being increased to one, the length of this transmission line hinders size, so they become the cost obstacle of HF and VHF application.Even 1 to 2GHz, these need respectively to implement about 2 to 1 inches trace length, and this remains too big and wastes important area.And this will be not the technology of scale high-resolution (for example integrated circuit) rightly, reason is that 1/4th needed length that obtain wavelength concerning such technology are too big magnitudes.At last, long like this quarter-wave resonance device is easy to be transmitted and receive The noise.

The another kind of famous circuit topology that is used to produce Bandpass Filters response is magnetic coupling, double tunning resonant circuit.The band pass filter of Shi Xianing is with respect to other prior art discussed here like this, and manufacturing cost is the most cheap (they can be made with several sharing money one).The realization of these known up till now filters can't reach and produce the big Q value have such as the response of required fraction bandwidth of the so a lot of application of The Wideband Signal Processing and low insertion loss (in general, they reached be no better than 15% fraction bandwidth or bigger bandwidth).In the following discussion, those of ordinary skill in the art can understand the reason of their shortcoming in these are used.

In Fig. 1 a, illustrate the common topology of series connection double-tuned circuit 10, in Fig. 1 b, illustrate the common topology of double-tuned circuit 100 in parallel.This series connection double-tuned circuit has input resonator circuit 12, and its magnetic coupling is to output resonator circuit 14.Equally, double-tuned circuit 100 in parallel has the input resonator circuit 120 of magnetic coupling to output resonator circuit 140. Input resonator 12 and 120 is coupled to respectively by power supply V _S18,180 with relevant source impedance R _SThe input power supply of 16 and 160 simulations (modeled). Output resonator 14 and 140 is coupled to by resistance R respectively _LThe output load impedance of 15 and 150 simulations.

The input and output resonator 12 and 14 of series connection double-tuned circuit 10 is conduct capacitor C separately _S111 and C _S213 with separately inductor L ₁17 and L ₂The lump series connection forms between 19.These two series tuning resonators 12 and 14 and two tuned resonators 120 in parallel and 140 are become the function of the physical proximity (physical proximity) between their inductor by magnetic coupling, create mutual inductance M21 thus between them.

Wherein k is a value for the geometry of inductance element and the coupling coefficient of the function of the physical proximity mutually.Therefore coupling coefficient k has reacted the percentage of potential mutual coupling total between two resonators.Two inductors 17 and 19 or 170 and 190 lean on closely more, and the value of k is big more so, so the mutual inductance between the inductor is big more; Equally, they are from must be far away more, and mutual inductance degree so is more low, by lower k value reflection.

Double-tuned circuit 100 in parallel is theoretical antithesis of series connection double-tuned circuit 10, therefore operates quite similar.The resonator 120 and 140 the capacitor C of double-tuned circuit 100 in parallel by lump _P1110 and C _P2130 with inductor L ₁170 and L ₂190 parallel respectively being connected to form.Resonator 120 and 140 is also become the function of the physical proximity between their inductor by magnetic coupling, create mutual inductance M210 thus between them.The mutual inductance of parallel-tuned circuit is by same formula Provide how much same consideration indications that its k value is discussed by the front.

Fig. 2 illustrates three kinds of typical response for the double tunning resonant circuit (serial or parallel connection) of different coupling coefficient k.Response 22 obtains during in resonance frequency in two resonator Critical Coupling of this circuit, and wherein resonance frequency is such point, shows minimum insertion loss and at the best of breed of the average selectivity at circuit resonant frequencies place at this point.Response 24 illustrates at double-tuned circuit 10 and 100 input and output resonator separately and is under the undercoupling (under-coupled) their response.This situation approaches to take place in 0 o'clock for the k value, and the k value approaches 0 and can realize by the resonator of circuit is removed.When being in undercoupling, the Q of these circuit _LValue increases (minimizing of fraction bandwidth), also increases but insert loss, and this is undesirable.Response 26 the input and output resonator two inductors be close to and make them take place when becoming super coupling (over-coupled) (that is, the k value of approaching 1).Response 26 is portrayed by two maximums on any one side of resonance frequency, but this circuit shows minimum Q _LValue (so their maximum fraction bandwidth).From these responses, as can be seen, at obtainable Q _LMaximum and insert between the loss, exist to be suitable for the compromise of double-tuned filter embodiment.For a known frequency, this compromise function as the mutual inductance M between the resonator of this filter embodiment works.Best coupling clearly takes place or near this critical range, reason is that it provides optimal compromise between Stopband Performance and insertion loss.

Notice that when frequency increases it is very important that the total inductance coupling between the resonator also increases this point.This is because the function of mutual inductance M (for the function of the geometrical property of resonator and the degree of approach) is not only in the total inductance coupling between the resonator, and is the function of the induction reactance of the direct proportion function (direct function) (that is ω M) for frequency.Therefore, when increasing frequency for known M value, the inductance coupling high between the resonator increases, and in fact this circuit becomes super coupling.To a point of determining, can compensate the increase of this coupling by the interval between the direct increase inductor, thereby reduce M by reducing k.But, the 1GHz scope and above in frequency under to increase be unpractical at interval.

The Q of series-tuned circuit _LBe defined as roughly at resonance frequency (ω ₀L) locate the reactance X of tuning circuit network divided by load of being coupled to it or source impedance.Therefore, the Q of output resonator 14 _LFor

For given resonance frequency omega ₀, can be by increasing L ₂Value increase Q _L(certainly, in order to increase total Q about series connection double tunning resonator _L, also should be by increasing L ₁ Value input resonator 12 is carried out identical processing).The problem of this method is, to the inductor L that can make and implement by should cost ₁And L ₂Size have physical constraints.In addition, work as L ₁And L ₂Value when increasing, the parasitic shunt capacitance relevant with the inductor (being typically coil) of lump value worsens the filter freguency response at the frequency place on 200MHz.At last, because resonance frequency by formula

(for output resonator 14) determined, so C _S2Value must reduce to keep ω in suitable ratio ₀Value.Also exist and accurately to build how little C _S2The physical constraints of aspect.

Fig. 3 illustrates and has k, C _S111 and C _S213 and L ₁17 and L ₂The series connection double-tuned circuit 10 of Fig. 1 of 19 values is designed to advance Q _LValue, holding circuit Best Coupling when the 400MHz resonance frequency simultaneously.Fig. 4 a and 4b illustrate the analog response of the circuit 30 of the component value with indication as shown in Figure 3.The many frequency (MHz) and decay (dB) to value indication point 1-4 of the bottom of Fig. 4 a and 4b are worth, with indicate on the response curve the same.The unacceptable performance of filter when the response shown in the scale that provides by Fig. 4 a illustrates high-frequency in processing television signals is used.The less scale that is provided by Fig. 4 b shows 3dB fraction bandwidth (so the Q into about 16% _LApproximation be 6.25).As previously discussed, this point is unacceptable to a lot of The Wideband Signal Processing application.

The Q of parallel-tuned circuit _LBe defined as roughly multiply by load and the source impedance that is coupled to it in the admittance (admittance) of this network of resonance frequency place.So Q of tuning output resonator 140 in parallel _LBe ≌ ω ₀C _P2R _LTherefore as can be seen, in order to increase the Q of tuning output resonator in parallel _L, can increase C _P2And R _LValue.Because signal can shunt to ground by the splitter component of parasitism, so R _LCan not be increased to far above 100 ohm.Increase C _P2Require L ₂Make very for a short time.Use has the unusual difficulty of inductor of lump of the about 5nH of known technology manufacturing of acceptable accuracy, reason be such inductor to change for how much, very responsive on vertically particularly.In addition, repeatedly obtain and keep between such a plurality of small coils correct coupling almost be impossible.These small coils need have little gap to keep Best Coupling (generally be in or near Critical Coupling) between them, and this coupling coefficient is very responsive to this closely spaced size.When the fraction bandwidth of needs 1%, such element and change in size can not be tolerated.

Fig. 5 illustrates and has k, C _P1110 and C _P2130 and L ₁170 and L ₂The double-tuned circuit in parallel 100 of Fig. 1 of 190 values has the Best Coupling that is designed to when the 400MHz resonance frequency and advances (push) Q _LThe L of value than the ratio of C.Fig. 6 a and 6b illustrate the analog response of the circuit 50 of the component value with indication as shown in Figure 5.The many of the bottom of Fig. 6 a and 6b are worth value indication frequency (MHz) and the decay (dB) for a 1-4, with indicate on the response curve the same.Response shown in the scale that provides by Fig. 6 a illustrates the unacceptable performance of stopband median filter, even it moves more symmetrically under the high-frequency with respect to the series-tuned circuit 30 of Fig. 3.The a plurality of coil values that promptly are used in this example of prior art are being advanced to the limit, and the bandwidth of this filter still is not too narrow to and is fit to a lot of degree of using.The less scale that is provided by Fig. 6 b shows 3dB fraction bandwidth (so the Q into about 15.5% _LApproximation be 6.45).As previously discussed, this point is to a lot of needs 1 to 2% (that is Q, _LValue be within 50 to 100 scopes) the The Wideband Signal Processing of fraction bandwidth to use be unacceptable.

Therefore, those of ordinary skill in the art will appreciate that be provided at be about at interval 50 to 2000MHz or bigger bandwidth on a lot of The Wideband Signal Processing use in necessity of band pass filter circuit of necessary characteristic.Those characteristics just provide the high Q of high selectivity and fraction bandwidth _LLow insertion loss in the decay of height in value, the stopband, the passband and the cheap and manufacturing repeatedly of tuned resonator circuit that can resemble prior art.

Disclosure of an invention

Therefore the purpose of the first embodiment of the present invention provides a kind of band pass filter that adopts double tunning magnetic coupling resonator topology in parallel, and it can be realized than the Q that used such topology to obtain in the past _LBe worth in fact higher Q _LValue.

Another purpose of first preferred embodiment is to realize having and can and having the topology of high-order repeatable accuracy, higher Q with cheap cost manufacturing _LValue.

The further purpose of first preferred embodiment is to quite immunity of the RF noise in its environment.

The purpose of the second embodiment of the present invention remains the higher Q of realization _LValue, it has low insertion loss and have precipitous falling between passband and stopband when lower and higher frequency, add a unique add ons to first embodiment.

The purpose of the third embodiment of the present invention remains the higher Q of realization _LValue, its (when lower and higher frequency) has low insertion loss and have precipitous falling between passband and stopband, adds a unique add ons to known series connection double resonance magnetic coupling resonator topology.

The purpose of the fourth embodiment of the present invention is, though at higher UHF frequency band range promptly at about 500MHz with between more than the 2GHz, also can use the non-obvious again circuit topology of easy manufacturing and cost-effective novelty, realize the Q that wishes _LValue, absolute bandwidth and insertion loss.

Consider detailed description of the present invention, to ability and those of ordinary skill, these purposes and other purpose can be clearly.

First preferred embodiment of band pass filter of the present invention adopts double tunning resonator topology in parallel, it is very short on electric by using (be approximately resonance frequency wavelength 1%) transmission line realize high Q as resonator by its magnetic-coupled very little inductance element _LValue.Transmission line is manufactured with the metal trace of precisely controlled physical dimension, realizes necessary inductance value by it.Dielectric constant with the thick printed circuit board material of 1.5mm is 4.65.This trace is with having the thick copper production of 0.018mm.Then according to keeping locating little charged sensor (microstripinductor) physically to obtain about coupling coefficient of 0.01 to 0.02 (k) to giving the required value of frequency Best Coupling.One end of transmission line trace is coupled on the series capacitors, and the other end terminates to ground.Can be approximately with accuracy ± 2% accurately produce the inductor be low to moderate about 0.5nH.

In second preferred embodiment of band pass filter of the present invention, by in each magnetic-coupled resonator, adding the double tunning resonator in parallel that a coupling capacitor is revised first preferred embodiment, this capacitor with and magnetic-coupled microstrip transmission line inductor shunt capacitance in parallel in series be coupled together, and have more less than with the value of the shunt capacitance of magnetic-coupled microstrip transmission line inductor parallel connection.

In the 3rd preferred embodiment of the present invention, by in each resonator, adding the series connection double tunning resonator topology that shunt capacitance is revised prior art, this shunt capacitance in parallel with the coupling of the series element of these two resonators, and have greater than with the value of the electric capacity of induction reactance series connection.This induction reactance preferably uses air windings (air coil) or other known lump inductive reactive element to realize.

The second and the 3rd embodiment can both be directly by series connection or by-pass capacitor with veractor or the alternative resonator of other known controlled capacitance, as electronic tuning unit.

The fourth embodiment of the present invention discloses compensation and perplexed the increase of first three inductance coupling high of implementing and the topology that Q reduces when tuning frequency surpasses 1GHz.This topology comprises the mirror image of previously disclosed each reflection about each resonator of the tuning parallel resonator topology of their holding wires separately.In fact the mirror image of each resonator is used to offset two mutual inductances between the resonator, thus compensation otherwise will obviously increase the inductance coupling high that increases with frequency.In addition, effective induction reactance value that the character in parallel of the inductor of reflection reduces each resonator reaches more than 50%, and making to increase the C of each resonator _PValue is so that with the decline of the load Q of the increase compensating circuit of frequency.

The inductor element of each resonator and its mirror image can be used as the wall scroll metal tape and realize that perhaps they preferably are embodied as the bar of several parallel connections, so that further reduce effective induction reactance of each resonator, and do not have the corresponding increase of inductance coupling high aspect.Bar with parallel connection is realized the convenience that inductor element brings, so that these adjust effective induction reactance value of each induction reactance value by adding short circuit metal, thereby allows filter in test by tuning.Certainly, people can reduce induction reactance by the width that increases bar, but inductance coupling high can correspondingly increase along with the minimizing of L value, and the inductance coupling high of exceeding increase by adjustment is come the tuned filter circuit.In addition, for each resonator with produce induction reactance as their mirror image of parallel-connection structure and allow to create short circuit, so that tuned filter in test by between many, adding metal.Certainly, people can adopt laser reconditioning to realize same target.

Any preferred embodiment can be put with different configurations,, be offset any common-mode noise that from environments, to introduce by inductor to make that by putting inductor the mesh current direction is opposite.The preferred embodiment can also be put by balance-balance and balanced-unbalanced configuration mode.Any preferred embodiment can physically be put its resonator at any non-specific position all relevant mutually.Parallel (parallel with 0 degree or 180 degree directions) or the special circumstances of vertically opposite position such as resonator are best, although other direction (such as 45 degree etc.) other topological flexibility can be provided, and the other degree of freedom that control coupling coefficient k is provided.The component value of the resonator of any preferred embodiment both can be put symmetrically, also can asymmetricly put, and both can be used for the frequency response that impedance conversion also can be used for adjusting filter.At last, a plurality of resonators in any preferred embodiment can cascade together so that increase the complexity of transfer function, thereby increase Q _LValue and passband are to the slope of stopband or fall.

Brief description of the drawings

Fig. 1 a is the diagram of the series connection double tunning magnetic coupling resonator topology of prior art.

Fig. 1 b is the diagram of the double tunning magnetic coupling resonator topology in parallel of prior art.

When Fig. 2 is coupling coefficient k change, the diagram of three kinds of typical response of the resonator of Fig. 1 a and 1b.

Fig. 3 has the limit component value of the known realization that is fit to this resonator to realize maximum Q _LThe example of series resonator of Fig. 1 a.

The analog response of the resonator of the prior art that Fig. 4 a is to use the big scale that is suitable for frequency (40MHz/div) and decay (10dB/div), be fit to Fig. 3.

The analog response of the resonator of the prior art that Fig. 4 b is to use the less scale that is suitable for frequency (10MHz/div) and decay (1dB/div), be fit to Fig. 3.

Fig. 5 has the limit component value of the existing techniques in realizing that is fit to this resonator to realize maximum Q _LThe example of parallel resonator of Fig. 1 b.

The analog response of the resonator of the prior art that Fig. 6 a is to use the big scale that is suitable for frequency (40MHz/div) and decay (10dB/div), be fit to Fig. 5.

The analog response of the resonator of the prior art that Fig. 6 b is to use the less scale that is suitable for frequency (10MHz/div) and decay (1dB/div), be fit to Fig. 5.

Fig. 7 is that the first embodiment of the present invention, practical small-sized ground connection microstrip transmission line are realized a very little but accurate effectively example of the parallel resonator of induction reactance.

Fig. 8 a is little vertical view that has the physical representation of imitating inductive reactive element of the present invention.

Fig. 8 b is an example of the parallel resonator of Fig. 7, and wherein inductance element is divided into three in parallel little bands shown in Fig. 8 a so that realize low effective induction reactance of resonator.

Fig. 9 a is to use the big scale that is suitable for frequency (40MHz/div) and decay (10dB/div), the analog response that is fit to the resonator of Fig. 8 b.

Fig. 9 b is to use the less scale that is suitable for frequency (10MHz/div) and decay (1dB/div), the analog response that is fit to the resonator of Fig. 8 b.

Figure 10 a is to use microstrip transmission line as block inductive reactive element and have a diagram of the in parallel tuning resonator circuit of the additional capacitive element that is connected between resonator and the input/output signal.

Figure 10 b is to use the diagram of physical embodiments of the in parallel tuning resonator of printed circuit board (PCB) manufacturing technology, Figure 10 a.

Figure 11 illustrates the embodiment of circuit of Figure 10, provides the component value of the narrow-band pass filter that can realize 70MHz.

Figure 12 a is to use the big scale that is suitable for frequency (40MHz/div) and decay (10dB/div), the analog response that is fit to the resonator of Figure 11.

Figure 12 b is to use the less scale that is suitable for frequency (10MHz/div) and decay (1dB/div), the analog response that is fit to the resonator of Figure 11.

Figure 13 is an example of the in parallel tuning resonator of Figure 10 a, has the component value that can realize the 400MHz narrow-band pass filter.

Figure 14 a is to use the big scale that is suitable for frequency (40MHz/div) and decay (10dB/div), the analog response that is fit to the resonator of Figure 13.

Figure 14 b is to use the less scale that is suitable for frequency (10MHz/div) and decay (1dB/div), the analog response that is fit to the resonator of Figure 13.

Figure 15 is an embodiment of the in parallel tuning resonator of Figure 10 a, has the component value that can realize the 800MHz narrow-band pass filter.

Figure 16 a is to use the big scale that is suitable for frequency (40MHz/div) and decay (10dB/div), the analog response that is fit to the resonator of Figure 15.

Figure 16 b is to use the less scale that is suitable for frequency (10MHz/div) and decay (1dB/div), the analog response that is fit to the resonator of Figure 15.

Figure 17 illustrates an embodiment of the in parallel tuning resonator of Figure 10 a, and for it, the inductance element of each resonator is realized with little band of three parallel connections, so that further reduce the induction reactance value of resonator.

Figure 18 a is to use the big scale that is suitable for frequency (40MHz/div) and decay (10dB/div), the analog response that is fit to the resonator of Figure 17.

Figure 18 b is to use the less scale that is suitable for frequency (10MHz/div) and decay (1dB/div), the analog response that is fit to the resonator of Figure 17.

Figure 19 is an embodiment of the in parallel tuning resonator of Figure 10 a, has the resonator of three parallel connections that can realize the 400MHz narrow-band pass filter.

Figure 20 a is to use the big scale that is suitable for frequency (40MHz/div) and decay (10dB/div), the analog response that is fit to the resonator of Figure 19.

Figure 20 b is to use the less scale that is suitable for frequency (10MHz/div) and decay (1dB/div), the analog response that is fit to the resonator of Figure 19.

Figure 21 adopts balanced-unbalanced transformer to realize an embodiment of the in parallel tuning resonator 400MHz narrow-band pass filter, Figure 10 a.

Figure 22 a is to use the big scale that is suitable for frequency (40MHz/div) and decay (10dB/div), the analog response that is fit to the resonator of Figure 21.

Figure 22 b is to use the less scale that is suitable for frequency (10MHz/div) and decay (1dB/div), the analog response that is fit to the resonator of Figure 21.

Figure 23 be to use air windings as inductance element, have a diagram of the series tuning resonator of the building-out condenser that is connected in parallel between the input/output signal resonator.

Figure 24 is the diagram with embodiment of series tuning resonator component value, Figure 23 of realizing the 70MHz narrow-band pass filter.

Figure 25 a is to use the big scale that is suitable for frequency (40MHz/div) and decay (10dB/div), the analog response that is fit to the resonator of Figure 24.

Figure 25 b is to use the less scale that is suitable for frequency (10MHz/div) and decay (1dB/div), the analog response that is fit to the resonator of Figure 24.

Figure 26 is the diagram with embodiment of series tuning resonator component value, Figure 23 of realizing the 400MHz narrow-band pass filter.

Figure 27 a is to use the big scale that is suitable for frequency (40MHz/div) and decay (10dB/div), the analog response that is fit to the resonator of Figure 26.

Figure 27 b is to use the less scale that is suitable for frequency (10MHz/div) and decay (1dB/div), the analog response that is fit to the resonator of Figure 26.

Figure 28 is the diagram with embodiment of series tuning resonator component value, Figure 23 of realizing the 800MHz narrow-band pass filter.

Figure 29 a is to use the big scale that is suitable for frequency (40MHz/div) and decay (10dB/div), the analog response that is fit to the resonator of Figure 28.

Figure 29 b is to use the less scale that is suitable for frequency (10MHz/div) and decay (1dB/div), the analog response that is fit to the resonator of Figure 28.

Figure 30 is a table, and it provides the block inductance value of equivalence for the resonator of Fig. 8 b, 11,13,15,17,19 and 21 described each embodiment.

Figure 31 adopts the tuned resonator circuit in parallel of Figure 10 a to realize an example of 400MHz oscillator.

Figure 32 a is as embodiment tuned resonator in parallel, mirror image topology of the present invention who is applied to Figure 10 a.

Figure 32 b be have more than two as being applied to, an embodiment of the mirror image topology of the tuned resonator in parallel of resonator that the inductance element of each resonator adopts many cascade in parallel.

Figure 32 c illustrates the symmetric property of mirror image topology, as the resonator of the cascade that is applied to Figure 32 b.

Figure 33 a-d illustrates for the faradic of mirror image topology of the present invention and progressively determines.

Figure 34 a illustrates an embodiment as the mirror image topology of the cascade circuit that is applied to Figure 32 b, uses the printed circuit board (PCB) treatment technology to realize and have the component value of the narrow-band pass filter of realizing 1015.75MHz.

Figure 34 b is to use the response of the actual measurement of the resonator big scale, Figure 34 a that is suitable for frequency (100MHz scope) and decay (10dB/div).

Figure 34 c is to use the response of the actual measurement of the resonator less scale, Figure 34 b that is suitable for frequency (6MHz scope) and decay (1dB/div).

Figure 34 d is to use the response of the actual measurement of the resonator very big scale, Figure 34 a that is suitable for frequency (3GHz scope) and decay (10dB/div).

Figure 34 e is to use the return loss (return loss) that is suitable for the 100MHz scope scale and the actual measurement of resonator scale, Figure 34 a of (5dB/div) decaying.

Realize optimal mode of the present invention

It below is the detailed description of the preferred embodiments of the present invention.As previously discussed, the double tunning resonator of Fig. 3 and Fig. 5 can not obtain the required Q of a lot of broadband applications _LValue increases their Q even increase the LC ratio _LValue also is like this.For the double tunning resonator topology in parallel of Fig. 1 b and 5, restriction is that the value of L can not be reduced to above about 5nH.

In first preferred embodiment of the present invention shown in Figure 7, the metal trace that is formed by copper on printed circuit board (PCB) is taken on the inductor L of double tunning resonator 70 in parallel ₁72 and L ₂74.These metal trace are coupling in by-pass capacitor C respectively _P176 and C _P2An end points of 78; Their another end points separately terminate to ground.Use this technology, can obtain to be low to moderate the effective inductance value of 0.5nH with accuracy ± 2%.Therefore, can notice the Q of double tunning resonator in parallel _LValue can further be increased to and surpass the value that can directly be obtained by prior art, and immediate cause is that inductance value can accurately be reduced to below the 5nH, and it allows to increase C _P176 and C _P278 value.

Because the impedance of very small-sized inductor is very low, so microstrip transmission line as this novelty of lumped inductance device element and the added advantage of non-obvious usage is the current i that flows near resonance frequency the time ₁And i ₂(75) very big.The electric current that increases can promote the utilization of the energy that transmits between a plurality of resonators.Therefore, the coupling of the total inductance of circuit will be bigger for given M, even make under the situation of undercoupling, also allow filter to be coupled best.So Q _LValue can be the higher grace (courtesy) of undercoupling, will reduce owing to higher electric current but insert loss.In addition because the small inductor value causes by little physical size with about the very little physical appearance of PCB, so they for the RF noise (with interchangeably, radiation) neurological susceptibility obviously lower with respect to the inductor element of the lump of prior art.They are easily with high accuracy and high multiplicity manufacturing and cheap for manufacturing cost.At last, its example of this topological sum all can carry out scale according to the resolution of the manufacture process that is adopted.Therefore, when the resolution of the process of making printed circuit board (PCB) can be restricted to about 5mm with the minimum length of inductance bar, on silicon, make these topologys with the resolution that allows and can cause quite little inductor, and quite little effective inductance value.

Fig. 8 a illustrates to build thereon inductor element L ₁72 and L ₂The a fraction of vertical view of the PCB of 74 (Fig. 7).In the preferred embodiment, these inductor elements form as copper microstrip trace 82 and 84 respectively on the end face 81 of PCB 80.This is with slightly and uses well-known precipitated metal and etching technique manufacturing.This slightly with physical dimension (that is, high by 86, wide 87), the spacing between them 89 determine the effective inductance of these elements, and the measuring of mutual inductance M73 that is given as the function of coupling coefficient k.The thickness of trace is 0.018mm preferably.The thickness of PCB or height 85 1.5mm preferably are by the material structure with dielectric constant of 4.65.This slightly with termination end be grounding to the ground plane 88 of PCB 80 through through hole 802.Through hole 802 has himself the self-induction (diameter according to this hole is about 0.1nH) that must consider when implementing.If desired, provide a plurality of ground holes will reduce the total inductance in these holes.Ground plane 88 generally is formed on the back side of PCB, but can be positioned at end face or the inside of PCB 80.

In a preferred embodiment, little band can by shown in little band the inside fraction 83 of etching away metal be split into a plurality of little band in parallel.This provides the extra degree of freedom in control aspect the inductance effective value of coupling coefficient k.For example, by the microstrip line (shown in Fig. 8 a) that adopts three parallel connections, each has, and 2mm is wide, 5.5mm long, as each parallel connection combination that all has the inductance element of bigger inductance value, can realize the effective inductance of about 0.72nH.The effective inductance that is realized by such parallel connection combination approximates 1/n L, and wherein n is the quantity of little band of each parallel connection that inductance value L is arranged.The benefit of wall scroll of using little band in parallel rather than having a width of the width sum that equals n bar in parallel is, obviously reduces for bracing also in the increase aspect the coupling corresponding to the wide increase of bar.But there are some physical constraints about little band quantity that can in the parallel connection combination, adopt.One is the echo that successively decreases (return) about the extra bar of each interpolation, and another is such fact: when the overall width of inductor bar increased, this impedance can begin to show with distribution mode rather than lump mode.The circuit that the filter of the three-way inductance element of employing Fig. 8 a is realized carries out diagram with the component value among Fig. 8 b.

By with the response ratio of the simulation of Fig. 9 a and Fig. 9 b (the present invention) output response and Fig. 6 a and Fig. 6 b (prior art), the topology of the existing techniques in realizing that exceeds the lumped inductance device element (Fig. 5) that utilizes prior art is shown, utilizes the improved response of the double tunning resonator topology of little charged sensing unit.When the first embodiment of the present invention is 400MHz in resonance frequency, obtain to be approximately 25 Q _L(with about 4% fraction bandwidth), and the Q of prior art under the same frequency _LBe approximately 6.5 (with about 15.5% fraction bandwidth).Attenuation outside a channel also obviously improves.

Those of ordinary skill in the art will appreciate that in the magnetic coupling resonator with microstrip transmission line as effective inductor component be novel and non-obviously, this obviously be different from microstrip transmission line as resonator use the prior art usage.When its length was the proper proportion (be generally wavelength 1/4th) of center or resonance frequency, microstrip transmission line relied on the natural resonance of transmission line as the usage of resonator.The present invention adopts length only to be little band of 0.5% to 10% of the wavelength of interested resonance frequency.They can take on the lump sensing element effectively in the mode of transmission-line efficiency, rather than take on distributed impedances.As previously mentioned, using transmission line as resonator about interested broadband application, the transmission line of overlength length when needing to suppress low frequency.

Figure 10 a illustrates the second embodiment of the present invention, wherein in series adds building-out condenser in the parallel tuning input 432 of the resonator of the topology (Fig. 7) of first preferred embodiment of the present invention and output 434 and (is respectively C _S1431 and C _S2433).With respect to by-pass capacitor C _P176 and C _P278, C _S1431 and C _S2433 is very little.And the interpolation of such series capacitor may not be intuitively to one skilled in the art, C _S1431 and C _S2433 interpolation has improved the response of the band pass filter of first preferred embodiment in fact significantly.The interpolation of two elements that are dirt cheap has changed over six grades of filters with this band pass filter from the level Four filter.This can compare by the improved topology of the transfer function that will derive about the conduct that parallel double tunning topology of the present invention (Fig. 7) realizes and Figure 10 a finds out.

The transfer function of the topology of Fig. 7 is provided by following formula $H\left(s\right)={\mathrm{<figure-callout\; id="0"\; label="g"\; filenames="A99815887.900251.png,A99815887.900281.png"\; state="\{\{state\}\}">g</figure-callout>}}_0\&CenterDot;\frac{s^3}{({\mathrm{<figure-callout\; id="2"\; label="s"\; filenames="A99815887.900241.png,A99815887.900331.png"\; state="\{\{state\}\}">s</figure-callout>}}^2+a_{1}s+b_1)({\mathrm{<figure-callout\; id="2"\; label="s"\; filenames="A99815887.900241.png,A99815887.900331.png"\; state="\{\{state\}\}">s</figure-callout>}}^2+a_2+b_2)}$ 。Transfer function about the enhancing topology of Figure 10 a is $H\left(s\right)=g_P\&CenterDot;\frac{s^3}{({\mathrm{<figure-callout\; id="3"\; label="s"\; filenames="A99815887.900281.png,A99815887.900361.png"\; state="\{\{state\}\}">s</figure-callout>}}^3+c_1{\mathrm{<figure-callout\; id="2"\; label="s"\; filenames="A99815887.900241.png,A99815887.900331.png"\; state="\{\{state\}\}">s</figure-callout>}}^2+d_{1}s+e_1)({\mathrm{<figure-callout\; id="3"\; label="s"\; filenames="A99815887.900281.png,A99815887.900361.png"\; state="\{\{state\}\}">s</figure-callout>}}^3+c_2{\mathrm{<figure-callout\; id="2"\; label="s"\; filenames="A99815887.900241.png,A99815887.900331.png"\; state="\{\{state\}\}">s</figure-callout>}}^2+d_{2}s+e_2)}$ 。(s=complex frequency (being σ+j ω) wherein, g ₀And g _pBe constant, a ₁, b ₁, a ₂, b ₂, c ₁, d ₁, e ₁, c ₂, d ₂And e ₂Be multinomial coefficient).By with slope from

Change to $\frac{1}{{\mathrm{<figure-callout\; id="3"\; label="s"\; filenames="A99815887.900281.png,A99815887.900361.png"\; state="\{\{state\}\}">s</figure-callout>}}^3},$ Falling of stopband when the limit of transfer function of adding the frequency response of the filter that definition revises to increases from passband to high frequency.Therefore Q not only _LStill further increase, and high frequency attenuation also strengthens.At last, C _S1431 and C _S2433 can also improve the low frequency performance of filter.

Figure 11 shows the realization of band-pass circuit that the topology of using Figure 10 a (microstrip transmission line that comprises first embodiment) has the centre frequency of 70MHz.In Figure 12 a and Figure 12 b, illustrate the analog response of the filter of Figure 11.The Q of this circuit _LValue is approximately 21; The fraction bandwidth is about 4.8%.

Figure 13 shows the realization of band-pass circuit that the topology of using Figure 10 a (microstrip transmission line that comprises first embodiment) has the centre frequency of 400MHz.In Figure 14 a and Figure 14 b, illustrate the analog response of the filter of Figure 13.The Q of this circuit _LValue is approximately 21; The fraction bandwidth is about 4.8%.

Figure 15 shows the realization of band-pass circuit that the topology of using Figure 10 a (microstrip transmission line that comprises first embodiment) has the centre frequency of 800MHz.In Figure 16 a and Figure 16 b, illustrate the analog response of the filter of Figure 15.The Q of this circuit _LValue is approximately 15; The fraction bandwidth is about 6.6%.

Figure 17 shows the realization of band-pass circuit that the topology of using Figure 10 a microstrip transmission line of the parallel connection of multiple bar chart 8a and 8b (but comprise) has the centre frequency of 400MHz.In Figure 18 a and Figure 18 b, illustrate the analog response of the filter of Figure 17.The Q of this circuit _LValue is approximately 34; The fraction bandwidth is about 2.9%.

Figure 19 shows the realization of band-pass circuit that the topology of using Figure 10 a (microstrip transmission line that comprises first embodiment) has the centre frequency of 400MHz, and wherein additional resonance device 1900 is coupling between the input and output resonator 432,434.Resonator 1900 has identical topology with resonator 432,434, has the capacitor C in parallel with little charged sensing unit 1904 _P1902.In Figure 20 a and Figure 20 b, illustrate the analog response of the filter of Figure 19.The Q of this circuit _LValue is approximately 19.5; The fraction bandwidth is about 5%.

Figure 21 shows the realization of band-pass circuit that the topology of using Figure 10 a (microstrip transmission line that comprises first embodiment) has the centre frequency of 400MHz.This circuit comprises to the input of input resonator 432 balances and to output resonator 434 unbalanced outputs (vice versa).This circuit is used as signal synthesizer or demultiplexer in passable frequency range.In Figure 22 a and Figure 22 b, illustrate the analog response of the filter of Figure 21.The Q of this circuit _LValue is approximately 2.4; The fraction bandwidth is about 42%.

Figure 23 illustrates the 3rd preferred embodiment of the present invention, wherein adds building-out condenser in parallel with the series tuning input 320 of the prior art topology of Fig. 3 and output 340 resonators and (is respectively C _P1350 and C _P2370).Series capacitor C with respect to Fig. 1 a _S111 and C _S213, C _P1350 and C _P2370 is very big.And the interpolation of such shunt capacitor may not be intuitively to one skilled in the art, C _P1350 and C _P2370 interpolation has improved the response of band pass filter of the prior art topology of Fig. 1 a and 3 in fact significantly.The interpolation of two elements that are dirt cheap has changed over six grades of filters with this band pass filter from the level Four filter, and it is the same among addition manner and Fig. 1 b and Fig. 5 series capacitance to be added to the mode of tuning circuit in parallel.It is substantially the same with the transfer function of top disclosed Figure 10 a to have the transfer function that the example of the present invention of the improved topology of Figure 23 derives.Reason be they in theory in correspondence with each other.

Figure 24 shows and uses Figure 23 (to utilize air windings to obtain high Q to inductor _LBe worth needed higher inductance value) the realization of band-pass circuit of topology with centre frequency of 70MHz.In Figure 25 a and Figure 25 b, illustrate the analog response of the filter of Figure 24.The Q of this circuit _LValue is approximately 46; The fraction bandwidth is about 2.2%.

Figure 26 shows and uses Figure 23 (to utilize air windings to obtain high Q to inductor _LBe worth needed higher inductance value) the realization of band-pass circuit of topology with centre frequency of 400MHz.In Figure 27 a and Figure 27 b, illustrate the analog response of the filter of Figure 26.The Q of this circuit _LValue is approximately 33.33; The fraction bandwidth is about 3%.

Figure 28 shows and uses Figure 23 (to utilize air windings to obtain high Q to inductor _LBe worth needed higher inductance value) the realization of band-pass circuit of topology with centre frequency of 70MHz.In Figure 29 a and Figure 29 b, illustrate the analog response of the filter of Figure 28.The Q of this circuit _LValue is approximately 34.8; The fraction bandwidth is about 2.9%.

Figure 30 is the table of value of various realizations that is applicable to the double tunning topology in parallel of circuit inductance device element microstrip line about employing, comprises size and other for information about.

When the frequency increase surpassed about 1GHz, inductance coupling high increased by such point, put the interval of passing through directly to increase between this resonator at this, and the minimizing of mutual inductance M can be actually used in the compensating inductance coupling to be increased with the maintenance Best Coupling.In addition, the increase of frequency reduces Q _LBe worth under such point, can directly shorten the length of metal tape so that reduce the value of the effective inductance L of each (the in parallel tuning realization of Fig. 7 or 10a) resonator at this point.Under the situation according to the preferred embodiment that utilizes the standard printed circuit board manufacturing tolerance to make, minimum length approximately is 5mm.Because the length of little band has become the tolerance control by manufacture process, the inaccuracy that therefore is reflected at the filter response aspect becomes the necessary given fraction bandwidth of unacceptable interested application.In addition, as previously mentioned, restricted so that reduce the number of elements of effective inductance of each resonator to can directly installing in parallel.

Figure 32 a discloses ground of the present invention four embodiment, and in this embodiment kind, each resonator of original topology (Fig. 7 and 10a) has self mirror image of the holding wire shown in being coupled to.This topology provides two kinds of very important features, and these features allow at from about 500MHz and the frequency range applications more than 2GHz.At first, it allows the effective inductance value minimizing of each resonator even is lower than such restriction, can shorten based on manufacturing tolerance this restriction metal tape.The inductance component L of input resonator _1a508 and L _1b509 and the inductance component L of output resonator _2a510 and L _2b512 is in parallel mutually respectively, and the effective inductance that therefore reduces the input and output resonator surpasses 50%.

Compensate Q owing to increase frequency _LThe reduction of value, the ability that therefore further reduces inductance value allows to increase shunt capacitor C _P1a504, C _P1b506 and C _P2a514, C _P2b516.In addition, even can be by with L _1a508, L _1b509, L _2a510 and L _2b512 as the further effective inductance that reduces each resonator of parallel connection combination that combines Fig. 8 a and 30 little bands (being respectively 606,608,610 and 612 among Figure 10 d) of describing with the front.As previously mentioned, there is one to the restriction of the quantity of little band of installation in parallel in this manner.In Figure 32 b graphic realization in addition produce than directly with compound mode in parallel install little band the littler inductance value of the inductance that obtained, such inductor comprises 606,608,610 and 612.

Second principal character that makes this topology be suitable for the extension frequency of 1 to 2GHz scope is: the inverse parallel naturally of this topology.Because it is opposite that the electric current in the inductance element is a direction, thus often can offset the mutual coupling between the resonator, thereby reduced the mutual inductance M (with whole inductance coupling high) between the resonator in fact.Therefore, even 1 to the frequency place more than the 2GHz, also can by in circuit with the variation of M approximate function as resonator, easily this coupling is remained in the optimum range.

Describe a kind of like this mode below with reference to Figure 33 a-d, in this mode, inverse parallel topology of the present invention has been offset the mutual inductance between the resonator.This analysis is to have under the zero width at the hypothesis inductor, carries out in series of steps.In the first step, at first shown in Figure 33 a, consider inductance component L _1a710 and L _2aMutual inductance between 712.Inductance between these two inductance elements is by formula $M_{1a,2b}=-\frac{{\mathrm{\&mu;}}_{0}b}{2\mathrm{\&pi;}}\{\ln \&lsqb;\frac{b}{a}+\sqrt{{\left(\frac{b}{d}\right)}^2+1}\&rsqb;+\frac{d}{b}-\sqrt{{\left(\frac{d}{b}\right)}^2+1}\}$ Provide.In second step, inductance component L _1a710 and L _2bMutual inductance between 714 is by formula $M_{1a,2b}=-\frac{{\mathrm{\&mu;}}_{0}b}{2\mathrm{\&pi;}}\{\ln \&lsqb;-\frac{b}{d}+\sqrt{{\left(\frac{b}{d}\right)}^2+1}\&rsqb;-\frac{d}{b}-\sqrt{{\left(\frac{d}{b}\right)}^2+1}\}$ Provide.In the 3rd step, the circuit of Figure 33 a and 33b superposes mutually to produce the circuit shown in Figure 33 c.Can simplify the formula of describing synthetic mutual inductance then: $M_{1a,2a}+M_{1a,2b}=-\frac{{\mathrm{\&mu;}}_{0}b}{2\mathrm{\&pi;}}\{\ln \&lsqb;(\frac{b}{d}+\sqrt{{\left(\frac{b}{d}\right)}^2+1})(-\frac{b}{d}+\sqrt{{\left(\frac{b}{d}\right)}^2+1})\&rsqb;+2\sqrt{{\left(\frac{d}{b}\right)}^2+1}\},$ Can also further be reduced to: $M_{1a,2a}+M_{1a,2b}=-\frac{{\mathrm{\&mu;}}_{0}b}{2\mathrm{\&pi;}}\{\ln \&lsqb;{\left(\frac{b}{d}\right)}^2+1-{\left(\frac{b}{d}\right)}^2\&rsqb;+2\sqrt{{\left(\frac{d}{b}\right)}^2+1}\}$ This formula can also further be reduced to Therefore, as can be seen, L _1aWith by L _2aAnd L _2bMutual inductance between the dipole that constitutes in fact and the gap between the inductor irrelevant.The final step that is used to analyze the mutual inductance between the mirror image resonator of the present invention will be determined L _1aWith by L _2aAnd L _2bMutual inductance (M between the dipole that constitutes _{1b, 2a, 2b}).Because pass through L _1bOpposite current in L _1aDirection flow, so except opposite in sign, in fact this mutual inductance is by about L _1aWith L _2aAnd L _2bThe same formula of the mutual inductance of the dipole that constitutes is given:

With L _1bBe overlapped into the structure shown in Figure 33 c with this dipole, cause as Figure 33 d the resonator topology of graphic mirror image of the present invention.Therefore, the mutual inductance between the resonator of these mirror images is $M_{1a,1b,2a,2b}=\frac{{\mathrm{\&mu;}}_{0}b}{\mathrm{\&pi;}}-\frac{{\mathrm{\&mu;}}_{0}b}{\mathrm{\&pi;}}=0$ 。Therefore the mutual inductance between the resonator of mirror image of the present invention for have with resonator between the interval relative long inductance element of comparing be zero.

As previously mentioned, the analysis of front hypothesis inductance element has zero width.This element width provides enough mutual induction amount, so that the resonator structure of mirror image obtains Best Coupling.But the induced current that increases with frequency is in a large number cancelled each other, to produce useful electric current.Be noted that between the resonator mutual inductance can also by inductance element whether mutually the degree of (parallel) in parallel control.Because a rotation about other in the inductance element of resonator is so can reduce the degree of counteracting.

Figure 32 b illustrates a preferred embodiment of the resonator topology of mirror image of the present invention.Owing to added the 3rd resonator 602, so about the exponent number height of the circuit of transfer function ratio Figure 32 a of the circuit of Figure 32 b.Resonator 602 has the structure about resonator 600 and 604 paraphase, but the equivalence in operation of this structure.Therefore resonator 600 and 604 also can carry out paraphase in such a way, as Figure 32 c graphic resonator 600i of institute and 604i.This symmetry operation provides about the extra degree of freedom of the physical layout of circuit.The embodiment of Figure 32 b and 32c also illustrates inductor element L _1a606, L _1b608 L _2a610, L _2b612, L _3a614 and L _3b616 realizations as little band of three parallel connections, wherein every little band provides about 1/3 effective inductance of the inductance of one of little band in parallel.All reduce about each the total effective inductance in three resonators and to surpass additionally 50%, therefore be less than the L of each independent little band of 1/6.

Should be noted that for the embodiment of Figure 32 a-c, the by-pass capacitor of each resonator is (such as the C of resonator 600 _P1a618 and C _P2620) all be in parallel, and the device value is added up to obtain total effective bypass electric capacity of each resonator.Realize each by-pass capacitor according to 2 or more each shunt capacitors,, be used for reducing dead resistance and inductance significantly, thereby improve the performance of filter circuit for each shunt capacitor provides the additional benefits of disposing dead resistance and inductance.

Figure 34 a shows actual a realization of the resonator topology of mirror image.It with have identical circuit in conjunction with the disclosed topology of Figure 32 b.In two width of cloth figure, components identical is used same labelled notation.The effective inductance of each in the resonator 600,602 and 604 all is 1.5nH.Centre frequency is the 1015.75MHz with 30MHz passband.Disclose the actual value of electric capacity and inductance element, comprised length, width and the gap metric of inductance element, comprised the interval S652 between length L 656 resonator of the clearance G 650 between little band in parallel, the width W 654 of little band, little band.Can use the resonator of mirror image of the present invention to obtain to have resonator than the much lower effective inductance of 0.5nH.Certainly, because the resolution of manufacture process is more and more meticulousr, so can reduce the minimum effective inductance of resonator.

Figure 34 b, 34c and 34d illustrate actual measurement, about the transfer function of the circuit of Figure 34 a.The frequency that 3dB is ordered is respectively 1000MHz and 1030MHz, so the Q of this circuit _LValue is 34 concerning 3% fraction bandwidth.Figure 34 e shows the return loss of circuit that measure, Figure 34 a.

Except filtering application, the present invention can provide in the FREQUENCY CONTROL ability that its uniqueness characteristic promptly combines with low insertion loss in the different application of clear superiority and utilize.An application example like this is to use the present invention on the feedback path of oscillator, as shown in figure 31.The input/output end port of resonance filter 400 of coupling is connected to RF amplifier 3100 input/output end ports feedback path from the output port of amplifier 3100 to its input port can be provided, closed hoop is around the loop of amplifier 3100 effectively.In hypothesis loop gain during, be to vibrate under the frequency situation of 0 degree (or multiples of 360 degree) around the phase shift of this ring greater than 1 (being of the insertion loss of the yield value of amplifier 3100) greater than feedback path.The phase shift of the resonator structure 400 of coupling is 180 degree on centre frequency, and uses inverting amplifier (having phase shift in 180 degree) that total phase shift of 360 degree can be provided, and therefore, satisfies the required necessary condition of vibration.Use have 0 degree phase shift (for example, by little band with respect to other, will import 72 or export 74 little band Rotate 180 degree) the resonator of coupling, and follow the circuit of the noninverting amplifier that is applicable to amplifier 3100, also satisfy the required condition of vibrating.

The narrow bandwidth of magnetic-coupled resonator (is high Q _LValue) with centre frequency near precipitous phase slope be associated.Precipitous phase slope in the feedback control loop can improve the phase noise performance of the oscillator of Figure 31.

Claims (15)

1. circuit comprises:

Magnetic coupling is to first resonator of second resonator, and described first and second resonators comprise separately:

First inductance element that has first capacitor of first electric capacity and have first inductance is coupling between a holding wire and first earth point;

Second inductance element that has second capacitor of second electric capacity and have second inductance is coupling between the described holding wire and second earth point, so that it is opposite to flow through the actual direction of electric current of described first and second inductance elements; And

The product of wherein said first electric capacity and described first inductance is substantially equal to the product of described second electric capacity and described second inductance.

2. according to the circuit of claim 1, the described holding wire of wherein said first resonator is used for input signal is sent to described first resonator, and the described holding wire of described second resonator is used for the output signal from described circuit is sent to load, by first coupling capacitor of connecting described input signal is coupled to described first resonator, and described output signal is coupled to described load by second coupling capacitor of connecting with described second resonator with described first resonator.

3. according to the circuit of claim 1, wherein said first and second inductance and described first and second electric capacity are respectively inductance and the electric capacity that equates.

4. according to the circuit of claim 1, the Best Coupling between wherein said first and second resonators remains within the frequency range by the physical proximity between the inductance element that changes first and second resonators.

5. according to the circuit of claim 1, wherein one or more described first and second inductance elements of each described first and second resonator comprise the block inductance that is formed by metal remained line on the non-conductive in itself surface.

6. according to the circuit of claim 5, wherein said one or more described inductance elements are by mutual two of being coupled or many metal line form in parallel.

7. according to the circuit of claim 1, wherein one or more described first and second capacitors are formed by the capacitor of two or more parallel connections, so that reduce the ghost effect that is associated with described first and second capacitors.

8. circuit comprises:

In series magnetic-coupled mutually two or more resonators, described two or more resonators comprise separately:

First inductance element that has first capacitor of first electric capacity and have first inductance is coupling between a holding wire and first earth point;

Second inductance element that has second capacitor of second electric capacity and have second inductance is coupling between the described holding wire and second earth point, so that it is opposite to flow through the actual direction of electric current of described first and second inductance elements.

9. circuit according to Claim 8, the described holding wire of first resonator in wherein said two or more resonator is used for input signal is sent to described first resonator, and the described holding wire of second resonator in described two or more resonator is used for the output signal from described circuit is sent to load, by first coupling capacitor of connecting described input signal is coupled to described first resonator, and described output signal is coupled to described load by second coupling capacitor of connecting with described second resonator with described first resonator.

10. according to the circuit of claim 1, wherein described first and second inductance in each described two or more resonator and described first and second electric capacity are respectively inductance and the electric capacity that equates.

11. according to the circuit of claim 1, the Best Coupling between wherein said two or more resonators remains within the frequency range by the physical proximity between the inductance element that changes described two or more resonators.

12. according to the circuit of claim 1, wherein one or more described first and second inductance elements in each described two or more resonator comprise the block inductance that is formed by metal remained line on the non-conductive in itself surface.

13. according to the circuit of claim 5, wherein said one or more described inductance elements are by mutual two of being coupled or many metal line form in parallel.

14. according to the circuit of claim 1, wherein one or more described first and second capacitors are formed by the capacitor of two or more parallel connections, so that reduce the ghost effect that is associated with described first and second capacitors.

15. method that keeps high capacity Q and carry out Best Coupling about the in parallel tuning series resonant circuit on extended frequency range, this circuit has the magnetic coupling tuned resonator of two or more mutual series connection, each resonator comprises that is coupling in an inductance element that has inductance L between holding wire and the earth point, and be coupling in the capacity cell that has capacitor C between holding wire and the earth point, described method comprises step:

With inductance element as being realized by the block inductance that forms on the nonconducting in itself surface of metal wire;

Basically offset the whole mutual inductance electric currents between two or more resonators;

When frequency increases, reduce the L value and increase the C value; And

Control coupling between these two or more resonators by changing physical distance between two or more resonators.

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