CN117691909A - Linear induction motor three-level double-vector model prediction thrust control method and system - Google Patents
Linear induction motor three-level double-vector model prediction thrust control method and system Download PDFInfo
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/483—Converters with outputs that each can have more than two voltages levels
- H02M7/487—Neutral point clamped inverters
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/13—Observer control, e.g. using Luenberger observers or Kalman filters
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
- H02P21/18—Estimation of position or speed
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
- H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
- H02P25/06—Linear motors
- H02P25/062—Linear motors of the induction type
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
- H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
- H02P27/12—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation pulsing by guiding the flux vector, current vector or voltage vector on a circle or a closed curve, e.g. for direct torque control
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Abstract
The invention discloses a method and a system for controlling a three-level double-vector model predictive thrust of a linear induction motor, which belong to the technical field of linear motor control, and comprise the following steps: collecting system state parameters, and estimating thrust and primary flux linkage by combining a motor model; inputting the thrust reference value, the estimated thrust, the primary flux linkage and the like into a reference primary flux linkage vector generator, and calculating and outputting a reference primary flux linkage vector; solving a reference voltage vector according to the reference primary flux linkage vector, determining candidate voltage vector combinations according to the reference voltage vector position, and calculating respective corresponding duty ratios; substituting each candidate voltage vector combination and the duty ratio thereof into the defined cost function to determine the optimal voltage vector combination; and generating switching pulse according to the optimal voltage vector combination and the duty ratio thereof, and realizing the control of the linear induction motor. The invention can effectively reduce the implementation complexity of the double-vector model predictive thrust control in the linear induction motor system driven by three-level frequency conversion, and remarkably improves the steady-state performance of the system.
Description
Technical Field
The invention belongs to the technical field of linear motor control, and particularly relates to a three-level double-vector model predictive thrust control method and system for a linear induction motor.
Background
Compared with a track traffic system driven by a rotary induction motor, the linear induction motor can directly generate linear motion without a transmission mechanism such as a gear box, has the advantages of stronger climbing capacity, smaller turning radius, smaller sectional area and the like, and is highly matched with the development requirement of a complicated track traffic network. Meanwhile, along with the continuous increase of the passenger flow of the rail transit, a new line mostly adopts a direct current 1500V power supply mode so as to meet the requirements of traction motors with higher power levels. Compared with the two-level topology, the three-level inverter topology (the NPC three-level inverter topology is most widely applied) has the advantages of high output voltage sine degree, small voltage stress of a single power device, small output current distortion rate and the like. Therefore, the linear induction motor traction system driven by three-level frequency conversion has wide application prospect in rail transit.
However, the linear induction motor suffers from serious end effects due to the breaking of the primary core, so that the mutual inductance in the operation of the motor varies severely and causes serious thrust decay in the high-speed operation. The model predictive thrust control combines direct thrust control and model predictive control, takes thrust and primary flux linkage as control targets, adopts an on-line optimizing mode to determine the acting voltage vector, is easy to process multi-target constraint conditions, can effectively relieve thrust attenuation caused by end effect, simultaneously maintains quick dynamic response capability of the direct thrust control, and has great development potential in a linear induction motor driving system.
Because only one voltage vector acts in the whole control period, the traditional single vector model predicts that the thrust control has larger thrust fluctuation and current harmonic wave. In order to further improve steady-state performance, a mode of applying two voltage vectors in each control period, that is, a dual-vector model predictive thrust control is generally adopted, so that thrust fluctuation is obviously reduced under the condition of slightly increasing switching frequency, and related methods have been applied to linear induction motor systems driven by two-level inverters. However, the method is directly applied to the three-level inverter and still has certain difficulty. First, the significantly increasing discrete voltage vectors of the three-level inverter causes the number of voltage vector combinations to increase dramatically. Without considering simplification, there are only 49 kinds of double vector combinations for the two-level inverter, but 729 kinds of double vector combinations for the three-level inverter, and it is difficult to determine the optimal voltage vector combination using the existing method. Second, existing approaches have difficulty in considering inherent control requirements imposed by three-level inverter topologies, such as neutral voltage balancing and smooth switching transitions. Therefore, although the core idea of the dual vector model predictive thrust control is general, the existing method is difficult to effectively expand to a three-level variable frequency driven linear induction motor system to achieve the improvement of steady state performance.
Disclosure of Invention
Aiming at the defects of the prior art, the invention aims to provide a linear induction motor three-level double-vector model prediction thrust control method and system considering voltage vector optimization, and aims to solve the problems that the existing double-vector model prediction thrust control is complex in algorithm, inherent control requirements of a three-level inverter are difficult to consider effectively and the like when the existing double-vector model prediction thrust control is applied to a linear induction motor system driven by three-level frequency conversion.
In order to achieve the above purpose, the invention provides a linear induction motor three-level double-vector model prediction thrust control method, which comprises the following steps:
s1: collecting state parameters of a linear induction motor control system driven by an Neutral-Point-Clamped (NPC) NPC type three-level inverter in real time, wherein the state parameters comprise motor phase current, speed and inverter Neutral Point voltage; calculating the motor thrust and the primary flux linkage at the current moment by using a thrust and flux linkage observer according to the sampling result;
s2: inputting a thrust reference value, a primary flux linkage amplitude reference value, motor thrust at the current moment, motor speed and primary flux linkage generated by a speed controller into a reference primary flux linkage vector generator, and outputting a reference primary flux linkage vector through calculation of the reference primary flux linkage vector generator;
s3: solving a reference voltage vector according to the reference primary flux linkage vector, determining candidate voltage vector combinations according to the reference voltage vector positions, and calculating the duty ratio corresponding to each candidate voltage vector combination; substituting each candidate voltage vector combination and the duty ratio thereof into the defined cost function to determine the optimal voltage vector combination;
s4: and generating switching pulses of each bridge arm according to the optimal voltage vector combination and the duty ratio thereof, and acting on the three-level inverter to realize the control of the linear induction motor.
Further preferred is a method of determining a candidate voltage vector combination from a reference voltage vector position comprising the steps of:
dividing all discrete voltage vectors which can be generated by the NPC three-level inverter into a large vector, a medium vector, a small vector and a zero vector according to the magnitude; wherein the large vector magnitude is u 2/3 times of u dc The method comprises the steps of carrying out a first treatment on the surface of the The magnitude of the middle vector isU times dc The method comprises the steps of carrying out a first treatment on the surface of the U with small vector amplitude 1/3 times dc ；u dc Is the voltage of a direct current bus;
dividing an inverter output voltage plane into 6 large sectors I-VI by taking 6 middle vectors as dividing lines; wherein the angular bisector direction of the sector I is defined as an alpha-axis direction, and the boundary direction of the sector II and the sector III is defined as a beta-axis direction;
taking sector I as an example, take u 1α ＝u dc 3 and u 1β =0 is a boundary, dividing the I-th sector into 4 small areas: r1 to R4; determining candidate voltage vector combinations according to the specific sizes of the reference voltage vectors in combination with the table 1, and calculating the duty ratio corresponding to each candidate voltage vector combination; for the condition that the reference voltage vector is in other sectors, the reference voltage vector is rotated to the I sector through coordinate transformation, and then the condition in the I sector is adopted to judge the area, so that the determination is further madeDetermining the corresponding candidate voltage vector combinations;
TABLE 1
Substituting each candidate voltage vector combination and the duty ratio thereof into the defined cost function to determine the optimal voltage vector combination;
further preferably, for the candidate voltage vector combination (u i ,u j ) The control targets of thrust, primary flux linkage, midpoint voltage and switching state switching constraint are synthesized, and a designed cost function form is as follows:
wherein k is u As the weight coefficient of the light-emitting diode,d for taking into account the predicted value of the midpoint voltage after delay compensation opt For candidate voltage vector combinations (u i ,u j ) The corresponding duty cycle, expressed as:
H m in order to avoid the constraint condition of direct switching of P, N states according to the switch states, the expression is:
wherein S is x (x will be a, b and c, respectively, to denote the switching states of the x-phase, with the numbers "1", "0" and "-1" representing the switching states P, O and N, respectively; k and k+1 represent times k and k+1, respectively.
In another aspect, the present invention provides a linear induction motor three-level bi-vector model predictive thrust control system, comprising:
the thrust and primary flux linkage observation module is used for calculating the current moment thrust and primary flux linkage of the motor by using a thrust and flux linkage observer according to the phase current and speed of the motor and the primary flux linkage calculated at the previous moment;
the reference quantity conversion module is used for inputting the thrust reference value, the primary flux linkage amplitude reference value, the motor thrust at the current moment, the motor speed and the primary flux linkage generated by the speed controller into the reference primary flux linkage vector generator, outputting a reference primary flux linkage vector through operation, and further solving a reference voltage vector according to the optimal reference flux linkage vector;
the optimal voltage vector combination selection module is used for determining candidate voltage vector combinations according to the positions of the reference voltage vectors, calculating the duty ratios corresponding to the combinations, substituting the candidate voltage vector combinations and the duty ratios thereof into defined cost functions respectively, and selecting the candidate voltage vector combination with the smallest cost function as the optimal voltage vector combination;
and the pulse sequence control module is used for generating three-phase bridge arm pulses according to the optimal voltage vector combination and the duty ratio thereof and acting on the corresponding bridge arm of the NPC type three-level inverter to realize the control of the linear induction motor.
In general, the above technical solutions conceived by the present invention have the following beneficial effects compared with the prior art:
(1) According to the linear induction motor three-level double-vector model prediction thrust control method and system, a new voltage vector plane sector division method and a corresponding simplified search method are provided for a three-level inverter, so that the number of voltage vector combinations required to be evaluated can be greatly reduced, and the implementation complexity of the double-vector model prediction thrust control in a three-level variable frequency driven linear induction motor system is effectively reduced;
(2) According to the linear induction motor three-level double-vector model prediction thrust control method and system, switching times under different voltage vector combinations are analyzed, the voltage vector combinations to be evaluated are optimized by utilizing the redundancy small vector characteristics, and the switching frequency is effectively reduced; and inherent control requirements of the three-level inverter such as neutral point voltage balance, voltage vector smooth switching and the like are contained in a cost function through analysis and mathematical deduction, so that the control performance of the system is ensured.
Drawings
FIG. 1 is a flow chart of a method for controlling the predicted thrust of a three-level double-vector model of a linear induction motor;
fig. 2 is a T-type equivalent circuit diagram of the linear induction motor provided by the invention;
FIG. 3 is a block diagram of a reference primary flux linkage vector generator provided by the present invention;
FIG. 4 is a graph of the NPC type three level inverter topology and its voltage vector distribution provided by the present invention, (a) NPC type three level inverter topology, (b) NPC type three level inverter discrete voltage vector distribution;
FIG. 5 is a schematic diagram of voltage vector sector division provided by the present invention, (a) R1 region specific location, (b) R2 region specific location;
FIG. 6 is a diagram of a switching pulse diagram of a combination of different voltage vectors provided by the present invention, (a) (u) 13 ,u 25 ) Voltage vector combined switching pulse diagram, (b) (u) 14 ,u 25 ) Voltage vector combination switch pulse diagram;
FIG. 7 is a schematic diagram of a linear induction motor three-level bi-vector model predictive thrust control system according to the present invention;
fig. 8 is a comparison chart of effects of whether to consider P, N state direct switching provided by the embodiment of the present invention, (a) does not consider P, N state direct switching effects, (b) considers P, N state direct switching effects;
fig. 9 is a voltage-current waveform and harmonic analysis chart of each model predictive thrust control method under the same switching frequency provided by the embodiment of the invention, (a) conventional single-vector model predictive thrust control, (b) double-vector model predictive thrust control without voltage vector optimization, and (c) the method of the invention.
Detailed Description
The present invention will be described in further detail with reference to the drawings and examples, in order to make the objects, technical solutions and advantages of the present invention more apparent. It should be understood that the specific embodiments described herein are for purposes of illustration only and are not intended to limit the scope of the invention. In addition, the technical features of the embodiments of the present invention described below may be combined with each other as long as they do not interfere with each other.
The flow chart of the linear induction motor three-level double-vector model prediction thrust control method provided by the invention, as shown in fig. 1, specifically comprises the following steps:
s1: respectively sampling motor phase current and speed signals by using a current sensor and a speed sensor; respectively calculating the motor thrust and the primary flux linkage at the current moment by using a thrust observer and a flux linkage observer according to the sampling result;
specifically, as shown in fig. 2, compared with a rotary induction motor, due to the primary core breaking structure, an edge effect is generated, so that excitation inductance changes in motor operation, and in order to quantitatively describe the mutual inductance change, a function f (Q) is defined:
wherein: q=lr 2 /v 2 (L m0 +L l2 ) The method comprises the steps of carrying out a first treatment on the surface of the l is the initial length of the motor; v 2 Is the linear speed of the motor; r is R 2 The secondary resistor is a motor; l (L) l2 The secondary leakage inductance of the motor; l (L) m0 Exciting inductance when the motor is stationary;
according to the equivalent circuit shown in fig. 2, the linear induction motor mathematical model can be expressed as:
wherein u is 1 ＝u 1α +ju 1β And u 2 ＝u 2α +ju 2β Primary and secondary voltage vectors, respectively; i.e 1 ＝i 1α +ji 1β And i 2 ＝i 2α +ji 2β Is the primary sumA secondary current vector; psi phi type 1 ＝ψ 1α +jψ 1β Sum phi 2 ＝ψ 2α +jψ 2β Is a primary and secondary flux linkage vector; l (L) 1 ＝L meq +L l1 And L 2 ＝L meq +L l2 Primary and secondary inductances; r is R 1 And R is 2 Primary and secondary resistances for the motor; p is a differential operator, ω 2 For the secondary angular velocity omega 2 ＝v 2 Pi/tau, tau being the motor pole pitch; l (L) meq To consider the equivalent excitation inductance after the side effect, it can be expressed as:
L meq ＝L m0 (1-f(Q)) (3)
more specifically, the thrust and flux observer can be built based on the mathematical model in equation (2) as follows:
wherein k and k-1 represent motor state variables at the times k and k-1, respectively, superscript "] represents observed quantity,as leakage inductance, F e Is electromagnetic thrust, T s For the control period.
S2: inputting a thrust reference value, a primary flux linkage amplitude reference value, motor thrust at the current moment, motor speed and primary flux linkage generated by a speed controller into a reference primary flux linkage vector generator, and outputting a reference primary flux linkage vector through calculation of the reference primary flux linkage vector generator;
the structure of the reference primary flux linkage generator is shown in fig. 3, and in order to clearly explain the operation principle of the reference primary flux linkage generator, the thrust expression of the linear induction motor is first simplified to:
wherein omega 1 For the rotational angular velocity of the primary flux linkage, i.e. the synchronous angular velocity,τ c ＝σL 2 /R 2 . The laplace transform of formula (6) can be obtained:
wherein omega s Is slip angular velocity.
As can be seen from equation (7), the thrust transfer function with the slip angular velocity as an input is equivalent to a first order system. Thus, a PI controller may be employed to achieve good thrust tracking performance and thereby convert thrust error to equivalent slip angular velocity ω s . Angle of change in position of primary flux linkage delta theta 1 Calculated from the equivalent slip angular velocity and the secondary angular velocity. The phase angle of the reference primary flux linkage vector may be further based on the primary flux linkage phase angle θ at the current time 1 And a position change angle delta theta 1 And (5) calculating. After the primary flux linkage vector of the current moment is estimated in the formula (5), the phase angle theta 1 Calculated from its alpha-beta component:
finally, the primary flux linkage amplitude reference value and the phase angle reference value are used for referencing the primary flux linkage vector, and the primary flux linkage vector is used as a new control target. Wherein the thrust control target is embodied in its phase angle and the flux linkage control target is embodied in its amplitude. By introducing a reference primary flux linkage generator, the use of two different dimensional reference values (thrust and flux linkage) can be avoided to calculate the reference voltage vector.
S3: solving a reference voltage vector according to the reference primary flux linkage vector, determining candidate voltage vector combinations according to the reference voltage vector positions, and calculating the duty ratio corresponding to each candidate voltage vector combination; substituting each candidate voltage vector combination and the duty ratio thereof into the defined cost function to determine the optimal voltage vector combination;
specifically, for the delay caused by the calculation time of the actual control system, the influence of the delay needs to be compensated by further combining with the prediction of the motor mathematical model, so that the control precision of the controller is improved. And predicting the k+1 moment by sampling and observing values at the current k moment, wherein the prediction expression is as follows:
wherein the superscript k+1 represents the motor state variable at time k+1,u opt is the equivalent primary voltage applied to the inverter at the previous time.
Based on the dead beat principle, the reference voltage vector can be calculated by letting the primary flux linkage prediction value at time k+2 be equal to its reference value:
fig. 4 (a) is a topology diagram of an NPC type three-level inverter, in which each phase of the motor can be connected to the positive pole (P), the negative pole (N) or the neutral point (O) of the dc bus under different switching states; thus, three phases can be combined with each other to generate 27 voltage vectors to be evaluated, which are classified into four types according to their magnitudes: large vector { u } 1 ,u 3 ,...u 11 Intermediate vector { u } 2 ,u 4 ,...u 12 Small vector { u } 13 ,u 14 ,...u 24 Zero vector { u }, zero vector 25 ,u 26 ,u 27 The specific distribution of each voltage vector is shown in fig. 4 (b). In fig. 4 (b), points at which the double vectors can be actually synthesized are indicated by solid lines in the figure, while being distant fromThe point closest to the reference voltage vector will minimize the value of the cost function, thereby producing an optimal control effect; therefore, the selection of the optimal voltage vector combination is guided based on the reference voltage vector, each voltage vector combination can be effectively prevented from being substituted into the cost function to be repeatedly calculated and compared, and the algorithm complexity is effectively reduced.
Further, in order to facilitate the removal of as many voltage vector combinations as possible from the reference voltage vector, the inverter output voltage range is divided into 6 large sectors I to VI with 6 middle vectors as dividing lines. Taking sector I as an example, each boundary line corresponds to 1 to 2 voltage vector combinations: (u) 13 ,u 25 )，(u 14 ,u 25 )，(u 14 ,u 15 )，(u 13 ,u 16 )，(u 1 ,u 13 )，(u 1 ,u 14 )，(u 2 ,u 13 )，(u 2 ,u 14 )，(u 1 ,u 2 )，(u 13 ,u 24 )，(u 14 ,u 23 )，(u 1 ,u 12 )，(u 12 ,u 13 )，(u 12 ,u 14 ). To further reduce the number of voltage vector combinations that need to be evaluated, u 1α ＝u dc 3 and u 1β =0 is a boundary, dividing the I-th sector into 4 small areas: r1 to R4 are as shown in fig. 5, and the voltage vector combination is optimized in consideration of the number of switching operations. For example, (u) 13 ,u 25 ) Sum (u) 14 ,u 25 ) The same effect is achieved for both thrust and flux linkage control, however the former produces 2 more switching actions than the latter, as shown in FIG. 6, so that only (u) 14 ,u 25 ) As a candidate voltage vector combination.
When the reference voltage vector is in the I-th sector, according to the specific region in which the reference voltage vector is located and considering the optimization of the switching times, the candidate voltage vector combinations under different conditions can be obtained as shown in the table 1, whereinAnd->The alpha and beta components of the reference voltage vector, respectively.
TABLE 1
When the reference voltage is in the other sector n, it can be first rotated to the I sector through coordinate transformation, and then a specific area thereof is determined with reference to table 1, thereby determining respective corresponding candidate voltage vector combinations. The coordinate transformation formula is:
further, after determining the candidate voltage vector combinations, it is necessary to further determine the respective time of action, i.e. the duty cycle, of the two voltage vectors. When the reference voltage vector isFor the voltage vector combination (u i ,u j )，u i Duty cycle d of (2) opt Can be expressed as:
wherein, ||u, represents the dot product of two vectors i -u j The expression vector (u) i -u j ) Is a function of the magnitude of (a). Correspondingly, u j The duty cycle of (1-d) opt ). The duty cycle calculated by equation (12) is further constrained to [0,1 ]]Within the interval.
On the other hand, compared with a two-level inverter, because neutral points are led out, the neutral point voltage of the NPC type three-level inverter can change along with the change of the current flowing through the neutral points in the running process of the motor, and the larger neutral point voltage deviation of the inverter can cause the fluctuation of current and thrust, so that the neutral point voltage needs to be controlled in order to ensure the control performance of a motor driving system, and the neutral point voltage of the inverter is defined as:
ΔU c12 ＝U dc1 -U dc2 (13)
wherein U is dc1 And U dc2 Respectively the capacitance C 1 And C 2 And a voltage on the same. Based on kirchhoff's current law and the circuit topology shown in fig. 4, the midpoint voltage at time k+1 can be predicted according to the current motor phase current, the capacitor voltage and the switch state to be acted:
wherein,for the voltage vector u i The switch state matrix of each phase bridge arm at k moment is respectively represented by numerals of ' 1 ', ' 0 ' and ' -1 ' to represent switch states P, O and N ' -/->For a three-phase current matrix, c=c 1 ＝C 2 。
When there is a voltage vector combination (u based on equation (14) i ,u j ) When acting on a control period at the same time, the midpoint voltage prediction expression at the moment k+1 is as follows:
meanwhile, when the switching state of a certain phase is directly switched between P and N, high voltage jump occurs to the phase voltage, so that the motor current distortion even causes IGBT through in the state switching process, and measures are needed to avoid direct switching between P, N states. It should be noted that in the switching pulse mode shown in fig. 6, only two cases (regardless of the sequencing) can occur for direct switching of P and N, as shown in table 2.
TABLE 2
According to the switch state relationship given in Table 2, constraint H is defined m To avoid direct switching between P, N states, specifically:
where x will be a, b and c, respectively. The control targets of the thrust, the primary flux linkage, the midpoint voltage and the switching state switching constraint are synthesized, and the designed cost function is as follows:
wherein,to further consider the midpoint voltage of the delay compensation according to equation (15), k u Is a weight coefficient. And (3) substituting each candidate voltage vector combination and the duty ratio thereof into the formula (17) in sequence to calculate a cost function, and selecting the candidate voltage vector combination with the minimum cost function value as the optimal voltage vector combination.
S4: and generating switching pulses of each bridge arm according to the optimal voltage vector combination and the duty ratio thereof, and acting on the three-level inverter to realize the control of the linear induction motor.
Specifically, after the optimal voltage vector combination is determined, switching pulses of all bridge arms can be synthesized according to the switching states and the duty ratios of the two voltage vectors in the combination, and the switching pulses directly act on 12 IGBTs of the NPC three-level inverter to complete the control of the linear induction motor.
Fig. 7 is a schematic diagram of a linear induction motor three-level dual-vector model predictive thrust control system according to the present invention, which includes a thrust and primary flux linkage observation module 100, a reference amount conversion module 200, an optimal voltage vector combination selection module 300, and a pulse sequence control module 400, wherein,
the thrust and primary flux linkage observation module 100 is configured to calculate a current moment thrust and primary flux linkage of the motor according to the motor phase current, the motor speed and the primary flux linkage calculated at the previous moment by using a thrust and flux linkage observer;
the reference quantity conversion module 200 is configured to input the thrust reference value, the primary flux linkage amplitude reference value, the motor thrust at the current moment, the motor speed and the primary flux linkage generated by the speed controller into the reference primary flux linkage vector generator, output a reference primary flux linkage vector through operation, and further solve a reference voltage vector according to the optimal reference flux linkage vector;
the optimal voltage vector combination selection module 300 is configured to determine candidate voltage vector combinations according to positions of reference voltage vectors, calculate duty ratios corresponding to the combinations, respectively substitute the candidate voltage vector combinations and the duty ratios thereof into defined cost functions, and select the candidate voltage vector combination with the smallest cost function as the optimal voltage vector combination;
the pulse sequence control module 400 is configured to generate three-phase bridge arm pulses according to the optimal voltage vector combination and the duty ratio thereof, and act on corresponding bridge arms of the NPC-type three-level inverter, so as to control the linear induction motor.
The experimental effects of the double-vector model predictive thrust control method and the traditional double-vector model predictive thrust control without voltage vector optimization are shown in fig. 8, wherein the motor speed is 11m/s, the load is 50N, the State represents whether direct switching exists between P, N, state=1 represents that a certain phase has direct switching, and state=0 represents that no direct switching exists. From the implementation effect, the conventional method sometimes causes direct switching of P, N state, which causes large distortion of line voltage and phase current, thereby affecting the control performance of the motor. The proposed method, analyzed by table 2 and introducing constraints in the cost function, effectively prevents direct switching between P, N states.
In order to further verify the effect of the proposed control method on improving the steady state performance of the linear induction motor driving system, the steady state performance of the model predictive thrust control method in the two existing documents is respectively compared with that of the model predictive thrust control method in the same switching frequency (1.23 kHz), the model predictive thrust control method in the invention is shown in the following description (a) in fig. 9, the model predictive thrust control method in the prior art is shown in the following description, and the model predictive thrust control method in the prior art is shown in the following description. Fig. 9 shows line voltage, phase current and harmonic analysis of the same at motor speed 11m/s, load 180N for three methods. It can be seen that the proposed method shows better steady state performance at the same switching frequency by introducing dual vector control and taking into account voltage vector optimization.
It will be readily appreciated by those skilled in the art that the foregoing description is merely a preferred embodiment of the invention and is not intended to limit the invention, but any modifications, equivalents, improvements or alternatives falling within the spirit and principles of the invention are intended to be included within the scope of the invention.
Claims (8)
1. The linear induction motor three-level double-vector model prediction thrust control method is characterized by comprising the following steps of:
collecting motor phase current, motor speed and inverter neutral point voltage in real time; calculating the motor thrust and the primary flux linkage at the current moment according to the sampling result;
taking a thrust reference value, a primary flux linkage amplitude reference value, motor thrust at the current moment, motor speed and a primary flux linkage as inputs, and outputting a reference primary flux linkage vector through vector operation;
solving a reference voltage vector based on a dead beat principle according to a reference primary flux linkage vector, determining candidate voltage vector combinations according to the positions of the reference voltage vectors, and calculating the duty ratio corresponding to each candidate voltage vector combination; substituting each candidate voltage vector combination and the duty ratio thereof into the defined cost function, and selecting the candidate voltage vector combination with the minimum cost function as the optimal voltage vector combination;
and generating switching pulses of each bridge arm according to the optimal voltage vector combination and the duty ratio thereof, and acting on the three-level inverter to realize the control of the linear induction motor.
2. The control method according to claim 1, characterized in that the method of determining candidate voltage vector combinations from reference voltage vector positions comprises the steps of:
dividing all discrete voltage vectors which can be generated by the three-level inverter into 6 large vectors { u } according to magnitude 1 ,u 3 ,...u 11 } 6 middle vectors { u } 2 ,u 4 ,...u 12 } 12 small vectors { u } 13 ,u 14 ,...u 24 Sum of 3 zero vectors u 25 ,u 26 ,u 27 -a }; wherein the large vector magnitude is u 2/3 times of u dc The method comprises the steps of carrying out a first treatment on the surface of the The magnitude of the middle vector isU times dc The method comprises the steps of carrying out a first treatment on the surface of the U with small vector amplitude 1/3 times dc ；u dc Is the voltage of a direct current bus;
dividing an inverter output voltage plane into 6 large sectors I-VI by taking 6 middle vectors as dividing lines; wherein the angular bisector direction of the sector I is defined as an alpha-axis direction, and the boundary direction of the sector II and the sector III is defined as a beta-axis direction;
for sector I, let u 1α ＝u dc 3 and u 1β =0 is a boundary, dividing the I-th sector into 4 small areas: r1 to R4; determining candidate voltage vector combinations according to specific positions of the reference voltage vectors, and calculating duty ratios corresponding to the candidate voltage vector combinations; for the case that the reference voltage vector is in other sectors, the reference voltage vector can be rotated to the I sector through coordinate transformation, then the condition in the I sector is adopted to judge the area, and further the corresponding candidate voltage vector combination is determined.
3. The control method according to claim 2, wherein the reference voltage vector positionDetermining candidate voltage vector combinations includes: for candidate voltage vector combinations (u i ,u j ) The control targets of thrust, primary flux linkage, midpoint voltage and switching state switching constraint are synthesized, and a designed cost function form is as follows:
wherein k is u As the weight coefficient of the light-emitting diode,d for taking into account the predicted value of the midpoint voltage after delay compensation opt For candidate voltage vector combinations (u i ,u j ) The corresponding duty cycle, expressed as:
wherein,for the reference voltage vector, 1 is less than or equal to i, and j is less than or equal to 27.
4. A control method according to claim 3, wherein H m For the constraint condition designed according to the switch state and avoiding the direct switching of P, N states, the expression is as follows:
wherein S is x The switching state of the x phase is represented, x=a, b, c, and the numbers "1", "0" and "-1" are used to represent the switching states P, O and N, respectively; k and k+1 represent times k and k+1, respectively.
5. The utility model provides a linear induction motor three-level bi-vector model predicts thrust control system which characterized in that includes:
the thrust and primary flux linkage observation module is used for calculating the motor thrust and primary flux linkage at the current moment according to the motor phase current, the motor speed and the midpoint voltage of the inverter which are acquired in real time;
the reference quantity conversion module is used for taking a thrust reference value, a primary flux linkage amplitude reference value, motor thrust at the current moment, motor speed and a primary flux linkage as inputs, outputting a reference primary flux linkage through vector operation, and further solving a reference voltage vector based on a dead beat principle according to a reference primary flux linkage vector;
the optimal voltage vector combination selection module is used for determining candidate voltage vector combinations according to the positions of the reference voltage vectors, calculating the duty ratios corresponding to the candidate voltage vector combinations, substituting the candidate voltage vector combinations and the duty ratios into defined cost functions respectively, and selecting the candidate voltage vector combination with the smallest cost function as the optimal voltage vector combination;
and the pulse sequence control module is used for generating each bridge arm switch pulse according to the optimal voltage vector combination and the duty ratio thereof, acting on the three-level inverter and realizing the control of the linear induction motor.
6. The control system of claim 5, wherein the method of determining candidate voltage vector combinations based on reference voltage vector positions comprises:
dividing all discrete voltage vectors which can be generated by the three-level inverter into 6 large vectors, 6 medium vectors, 12 small vectors and 3 zero vectors according to the magnitude; wherein the large vector magnitude is u 2/3 times of u dc The method comprises the steps of carrying out a first treatment on the surface of the The magnitude of the middle vector isU times dc The method comprises the steps of carrying out a first treatment on the surface of the U with small vector amplitude 1/3 times dc ；u dc Is the voltage of a direct current bus;
dividing an inverter output voltage plane into 6 large sectors I-VI by taking 6 middle vectors as dividing lines; wherein the angular bisector direction of the sector I is defined as an alpha-axis direction, and the boundary direction of the sector II and the sector III is defined as a beta-axis direction;
for sector I, let u 1α ＝u dc 3 and u 1β =0 is a boundary, dividing the I-th sector into 4 small areas: r1 to R4; determining candidate voltage vector combinations according to specific positions of the reference voltage vectors, and calculating duty ratios corresponding to the candidate voltage vector combinations; for the case that the reference voltage vector is in other sectors, the reference voltage vector can be rotated to the I sector through coordinate transformation, then the condition in the I sector is adopted to judge the area, and further the corresponding candidate voltage vector combination is determined.
7. The control system of claim 6, wherein said determining a candidate voltage vector combination based on a reference voltage vector position comprises: for candidate voltage vector combinations (u i ,u j ) The control targets of thrust, primary flux linkage, midpoint voltage and switching state switching constraint are synthesized, and a designed cost function form is as follows:
wherein k is u As the weight coefficient of the light-emitting diode,d for taking into account the predicted value of the midpoint voltage after delay compensation opt For candidate voltage vector combinations (u i ,u j ) The corresponding duty cycle, expressed as:
wherein,for the reference voltage vector, 1 is less than or equal to i, and j is less than or equal to 27.
8. The control system of claim 7, wherein H m For the constraint condition designed according to the switch state and avoiding the direct switching of P, N states, the expression is as follows:
wherein S is x The switching state of the x phase is represented, x=a, b, c, and the numbers "1", "0" and "-1" are used to represent the switching states P, O and N, respectively; k and k+1 represent times k and k+1, respectively.
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