CN107623479B - Fault-tolerant fault control method and device for motor - Google Patents
Fault-tolerant fault control method and device for motor Download PDFInfo
- Publication number
- CN107623479B CN107623479B CN201710820605.9A CN201710820605A CN107623479B CN 107623479 B CN107623479 B CN 107623479B CN 201710820605 A CN201710820605 A CN 201710820605A CN 107623479 B CN107623479 B CN 107623479B
- Authority
- CN
- China
- Prior art keywords
- phase
- current
- motor
- voltage
- coordinate system
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired - Fee Related
Links
Images
Landscapes
- External Artificial Organs (AREA)
- Control Of Ac Motors In General (AREA)
Abstract
The invention discloses a motor fault-tolerant fault control method and a device, comprising the following steps: collecting non-fault 5-phase current in a six-phase motor in the current sampling period, and converting the 5-phase current into 5 current components in a decoupling coordinate system; predicting the phase current value of each phase of the motor in the next sampling period by using a mathematical model of the motor and 5 current components, and predicting the optimal voltage vector output by the motor controller in the next sampling period by using the instruction value of each phase current; converting the optimal voltage vector into 5 voltage components under a decoupling coordinate system, and limiting the voltage vector synthesized by the orthogonal and direct axis voltage components; and converting the amplitude-limited alternating-axis voltage component and direct-axis voltage component back to a phase coordinate system, converting the final optimal voltage vector into each phase duty ratio, and adjusting a driving signal of each phase voltage according to each phase duty ratio to drive the motor. The invention obviously reduces the phase current harmonic wave under the open-circuit fault tolerance working condition of the one-phase winding of the six-phase motor without influencing the dynamic performance of the motor.
Description
Technical Field
The invention belongs to the technical field of electromechanics, and particularly relates to a fault-tolerant fault control method and device for a motor.
Background
At present, a multi-phase motor is widely applied to the fields of high power density and high reliability, and has wide application prospects in the fields of wind power plants, electric automobiles, ship propulsion, aerospace and the like. Compared with a three-phase motor, the multi-phase motor has more control degrees of freedom, so that the control of the multi-phase motor is more flexible, when one-phase open circuit fault occurs in the multi-phase motor, the amplitude and the phase of each phase of current can still be redistributed through a control algorithm, the motor can still normally run, and the rated power of a system is reduced slightly.
In the six-phase motor with the fault-tolerant working condition, the spatial distribution of each phase winding is not symmetrical any more, the traditional space vector modulation method is not applicable any more, and researchers provide a model predictive control algorithm for solving the problem. The specific contents of the model predictive control algorithm are as follows: the frequency converter of the motor only has a limited number of switching states, the influence of each switching state on the working condition of the motor is different, and the controller calculates and predicts the influence of each state on the performance of the motor in each sampling period to obtain the switching state which enables the motor to track the operation instruction most quickly, namely the optimal switching state, and acts on the next sampling period. In a traditional model predictive control algorithm, the optimal switching vector acts on the whole sampling period, namely the duty ratio of each bridge arm power device is kept to be 1 or 0 all the time. The traditional model prediction method is not suitable for general modulation methods, such as sine wave pulse width modulation and space vector pulse width modulation, the modulation is based on the calculation, adjustment and distribution of the duty ratio of a power device, and when the duty ratio is always 1 or 0, the modulation method cannot work.
In conclusion, the traditional model prediction method has the advantage of high dynamic performance, but meanwhile, the algorithm is based on the principle of exhaustion of the switching state of the frequency converter, so that the method has the defects of large calculation amount, high current harmonic content, poor steady-state performance and high copper consumption, and the requirement of long-time fault-tolerant working condition operation is difficult to meet.
Disclosure of Invention
Aiming at the defects of the prior art, the invention aims to solve the technical problems that the traditional model prediction method has the defects of large calculated amount, high current harmonic content, poor steady-state performance and high copper consumption and is difficult to meet the requirement of long-time fault-tolerant working condition operation due to the principle that the algorithm is based on the on-off state of the exhaustive frequency converter.
In order to achieve the above object, in one aspect, the present invention provides a fault-tolerant control method for a motor, including the following steps:
(1) collecting non-fault 5-phase current in a six-phase motor in a current sampling period, and converting the 5-phase current into 5 current components in a decoupling coordinate system, wherein the 5 current components comprise alternating-axis and direct-axis current components related to the generation of electromagnetic torque of the motor and x-axis, y-axis and 0-axis current components unrelated to the generation of the electromagnetic torque, and one-phase current fault exists in the six-phase motor;
(2) predicting the phase current value of each phase of the motor in the next sampling period by using a mathematical model of the motor and 5 current components in the decoupling coordinate system obtained in the step (1), and predicting the optimal voltage vector output by the motor controller in the next sampling period according to the instruction value of each phase current and the predicted phase current of each phase of the motor in the next sampling period;
(3) converting the optimal voltage vector into 5 voltage components under a decoupling coordinate system, wherein the 5 voltage components comprise alternating-axis voltage components and direct-axis voltage components related to the generation of the electromagnetic torque of the motor, and when the amplitude of the voltage vector synthesized by the alternating-axis voltage components and the direct-axis voltage components exceeds an amplitude limit value, limiting the amplitude of the voltage vector to the amplitude limit value, and keeping the vector direction of the voltage vector unchanged to obtain final alternating-axis voltage components and direct-axis voltage components under the decoupling coordinate system;
(4) and (4) converting the final alternating-axis voltage component and the final direct-axis voltage component under the decoupling coordinate system obtained in the step (3) back to the phase coordinate system, determining a final optimal voltage vector, converting the final optimal voltage vector into each phase duty ratio, and adjusting a driving signal of each phase voltage according to each phase duty ratio to drive the motor.
The method comprises the steps of (1) calculating a required ideal voltage vector from a sampling current to a reference current by using a six-phase motor mathematical model in the step (2) through a discretized voltage equation in the step (2), converting the voltage vector under a rotating coordinate system into a phase coordinate system through a park transformation array and an inverse matrix of the park transformation array in the step (3), and in the step (4), calculating the duty ratio of each phase bridge arm under the phase coordinate system and combining a space vector modulation method to realize the drive control of the multi-phase motor under the fault-tolerant working condition.
The method solves the problem of large calculation amount caused by the fact that a model prediction control algorithm needs to enumerate various states of a frequency converter under the fault-tolerant working condition by predicting the current value and partial differential at the next moment to obtain the minimum difference value between the instruction value and the actual value, and obviously reduces the current harmonic content, improves the control precision of the motor and improves the current quality of the motor under the fault-tolerant working condition by combining the amplitude limiting value and the optimal voltage vector duty ratio modulation process.
Optionally, the step (1) comprises the steps of:
(1.1) measuring the electrical angle theta (k) of the motor and the phase current i of the remaining non-faulted phase of the motorabcuv(k) Setting the current sampling period as k time, phase current iabcuv(k) Component is Ia,Ib,Ic,Iu,IvWherein a, b, c, u and v respectively represent the windings of each non-fault phase;
(1.2) calculating the angular velocity ω from the electrical angle θ (k)e(k) Wherein e represents the rotation speed as an electrical angular speed;
(1.3) with the measured phase current iabcuv(k) Obtaining the current i under a decoupling coordinate system through park transformation and Clark transformationdqxy0(k) The decoupled current component is Id,Iq,Ix,Iy,I0In which Id,IqFor the direct and quadrature current components associated with the generation of electromagnetic torque, Ix,Iy,I0Is a current component that is not related to the generation of electromagnetic torque;
clark transformation matrix as TcAnd the park transformation matrix is TpWhere the index c represents a Clark transformation matrix, p represents a park transformation matrix, TcAnd TpRespectively, as follows:
wherein, I4An identity diagonal matrix representing 4x 4; the transformation from phase current to decoupled current is as follows:
[Id,Iq,Ix,Iy,I0]T＝Tp·Tc·[Ia,Ib,Ic,Iu,Iv]Twhere the superscript T represents the transpose of the matrix.
Optionally, the step (2) comprises the steps of:
(2.1) Current value i at the time of passing kdqxy0(k)、ωe(k) And a voltage equation in a mathematical model of the motor, predicting the current value i at the moment k +1dqxy0(k +1), setting the next sampling period as the moment of k + 1;
the voltage equation used to predict the current in the mathematical model of the motor is:
wherein v isd,vq,vx,vy,v0Are respectively voltage components in a decoupled coordinate system Id,Iq,Ix,Iy,I0Are each a component of the current, R, in the decoupled coordinate systemsThe value of the phase resistance is the value of the phase resistance,denotes the derivation of time t, Ld,Lq,Lx,Ly,L0Respectively, inductance values of the phases, psi, in the decoupled coordinate systemmIs a permanent magnet flux linkage of the motor;
discretizing the voltage equation by using a forward Euler methodWhere x is any variable to be discretized, TsFor the sampling period, x (k) represents the value of the variable at the k-th time, x (k +1) represents the value of the variable at the k +1 time, and the discretized equation is expressed as:
idqxy0(k+1)＝Aidqxy0(k)+BU(k)+C
wherein idqxy0(k+1)＝[Id(k+1),Iq(k+1),Ix(k+1),Iy(k+1),I0(k+1)]T， idqxy0(k)＝[Id(k),Iq(k),Ix(k),Iy(k),I0(k)]T，U(k)＝[vd,vq,vx,vy,v0]T(ii) a A. B, C are all coefficients of an equation, specifically:
i obtained by the above calculationdqxy0The (k +1) is the predicted current value at the time k + 1.
(2.2) generating a command value of a phase current from a rotational speed command value and a rotational speed measurement value by a Proportional Integral controller (PI) loop of a motor rotational speed
(2.3) applying the current i at the next momentdqxy0(k +1) and instruction valueThe difference value between them is used as a scalar function g (k +1), and the partial differentiation is made for each current variable of said function, and the minimum value of scalar function is calculated, so that the obtained solution vo(k +1) is the optimum powerPressing the vector;
wherein g represents a scalar function, and g (k +1) represents the value of this scalar function at time k + 1;
the process of solving the partial derivative of the voltage component under each decoupling coordinate system at the moment k +1 for the scalar function g is as follows:
the solution to this equation is expressed as:
vo(k+1)＝[vod(k+1),voq(k+1),vox(k+1),voy(k+1),vo0(k+1)]T，
wherein the subscript o indicates that the calculated voltage vector is the optimum voltage vector, vod(k+1),voq(k+1),vox(k+1),voy(k+1),vo0And (k +1) are respectively the optimal voltage components under the decoupling coordinate system at the moment of k + 1.
Optionally, the step (3) comprises the steps of:
(3.1) by pairing voThe voltage of each phase (k +1) is separated under a decoupling coordinate system through park transformation and Clark coordinate transformation to obtain a component v participating in the generation of electromagnetic torqueodq(k+1)：
vodq(k+1)＝[vod(k+1),voq(k+1)]T；
(3.2) setting the limit value v of the output capacity of the frequency convertermaxTo v is to vodqThe magnitude of (k +1) is limited.
The output capacity of the frequency converter is expressed as vmaxV is this vmaxIs generally set to the magnitude of the maximum voltage vector that can be output when the six-phase motor is normally operated,
wherein v isdcIndicating the voltage of the DC bus connected with the frequency converter；
The voltage component amplitude limiting process associated with electromagnetic torque generation is as follows:
optionally, the step (4) comprises the steps of:
(4.1) converting the processed optimal voltage vector back to a phase coordinate system, determining a final optimal voltage vector, and determining the duty ratio D (k +1) of each phase bridge arm power device through the following formula;
And (4.2) obtaining a driving signal of the motor through modulation, inputting the driving signal into the six-phase frequency converter, and driving the motor to operate.
In another aspect, the present invention provides a fault-tolerant control apparatus for a motor, including:
the phase current acquisition unit is used for acquiring non-fault 5-phase current in the six-phase motor in the current sampling period and converting the 5-phase current into 5 current components in a decoupling coordinate system, wherein the 5 current components comprise alternating-axis and direct-axis current components related to the generation of electromagnetic torque of the motor and x-axis, y-axis and 0-axis current components unrelated to the generation of the electromagnetic torque, and one-phase current fault exists in the six-phase motor;
the optimal voltage vector determining unit is used for predicting the phase current value of each phase of the motor in the next sampling period according to the mathematical model of the motor and 5 current components in the decoupling coordinate system acquired by the phase current acquiring unit, and predicting the optimal voltage vector output by the motor controller in the next sampling period according to the instruction value of each phase current and the predicted phase current of each phase of the motor in the next sampling period;
the voltage vector amplitude limiting unit is used for converting the optimal voltage vector into 5 voltage components under a decoupling coordinate system, wherein the 5 voltage components comprise alternating-axis voltage components and direct-axis voltage components related to the generation of the electromagnetic torque of the motor, and when the amplitude of the voltage vector synthesized by the alternating-axis voltage components and the direct-axis voltage components exceeds an amplitude limit value, the amplitude of the voltage vector is limited to the amplitude limit value, and the vector direction of the voltage vector is kept unchanged, so that the final alternating-axis voltage component and the final direct-axis voltage component under the decoupling coordinate system are obtained;
and the duty ratio modulation module is used for converting the final alternating-axis voltage component and the final direct-axis voltage component under the decoupling coordinate system back to the phase coordinate system, determining a final optimal voltage vector, converting the final optimal voltage vector into each phase duty ratio, and adjusting the driving signal of each phase voltage according to each phase duty ratio to drive the motor.
Optionally, the phase current collecting unit is specifically configured to perform the following steps:
(1.1) measuring the electrical angle theta (k) of the motor and the phase current i of the remaining non-faulted phase of the motorabcuv(k) Setting the current sampling period as k time, phase current iabcuv(k) Component is Ia,Ib,Ic,Iu,IvWherein a, b, c, u and v respectively represent the windings of each non-fault phase;
(1.2) calculating the angular velocity ω from the electrical angle θ (k)e(k) Wherein e represents the rotation speed as an electrical angular speed;
(1.3) with the measured phase current iabcuv(k) Obtaining the current i under a decoupling coordinate system through park transformation and Clark transformationdqxy0(k) The decoupled current component is Id,Iq,Ix,Iy,I0In which Id,IqFor the direct and quadrature current components associated with the generation of electromagnetic torque, Ix,Iy,I0Is a current component that is not related to the generation of electromagnetic torque;
clark transformation matrix as TcAnd the park transformation matrix is TpWhere the index c represents a Clark transformation matrix, p represents a park transformation matrix, TcAnd TpRespectively, as follows:
wherein, I4An identity diagonal matrix representing 4x 4; the transformation from phase current to decoupled current is as follows:
[Id,Iq,Ix,Iy,I0]T＝Tp·Tc·[Ia,Ib,Ic,Iu,Iv]Twhere the superscript T represents the transpose of the matrix.
Optionally, the optimal voltage vector determining unit is specifically configured to perform the following steps:
(2.1) Current value i at the time of passing kdqxy0(k)、ωe(k) And a voltage equation in a mathematical model of the motor, predicting the current value i at the moment k +1dqxy0(k +1), setting the next sampling period as the moment of k + 1;
the voltage equation used to predict the current in the mathematical model of the motor is:
wherein v isd,vq,vx,vy,v0Are respectively voltage components in a decoupled coordinate system Id,Iq,Ix,Iy,I0Are each a component of the current, R, in the decoupled coordinate systemsThe value of the phase resistance is the value of the phase resistance,
denotes the derivation of time t, Ld,Lq,Lx,Ly,L0Respectively, inductance values of the phases, psi, in the decoupled coordinate systemmIs a permanent magnet flux linkage of the motor;
discretizing the voltage equation by using a forward Euler methodWhere x is any variable to be discretized, TsFor the sampling period, x (k) represents the value of the variable at the k-th time, x (k +1) represents the value of the variable at the k +1 time, and the discretized equation is expressed as:
idqxy0(k+1)＝Aidqxy0(k)+BU(k)+C
wherein idqxy0(k+1)＝[Id(k+1),Iq(k+1),Ix(k+1),Iy(k+1),I0(k+1)]T， idqxy0(k)＝[Id(k),Iq(k),Ix(k),Iy(k),I0(k)]T，U(k)＝[vd,vq,vx,vy,v0]T(ii) a A. B, C are all coefficients of an equation, specifically:
i obtained by the above calculationdqxy0The (k +1) is the predicted current value at the time k + 1.
(2.2) generating the instruction value of the phase current by the rotating speed instruction value and the rotating speed measurement value through a motor rotating speed proportional-integral controller loop
(2.3) applying the current i at the next momentdqxy0(k +1) and instruction value
The difference value between them is used as a scalar function g (k +1), and the partial differentiation is made for each current variable of said function, and the minimum value of scalar function is calculated, so that the obtained solution vo(k +1) is the optimal voltage vector;
wherein g represents a scalar function, and g (k +1) represents the value of this scalar function at time k + 1;
the process of solving the partial derivative of the voltage component under each decoupling coordinate system at the moment k +1 for the scalar function g is as follows:
the solution to this equation is expressed as:
vo(k+1)＝[vod(k+1),voq(k+1),vox(k+1),voy(k+1),vo0(k+1)]T，
wherein the subscript o indicates that the calculated voltage vector is the optimum voltage vector, vod(k+1),voq(k+1),vox(k+1),voy(k+1),vo0And (k +1) are respectively the optimal voltage components under the decoupling coordinate system at the moment of k + 1.
Optionally, the voltage vector clipping unit is specifically configured to perform the following steps:
(3.1) by pairing voThe voltage of each phase (k +1) is separated under a decoupling coordinate system through park transformation and Clark coordinate transformation to obtain a component v participating in the generation of electromagnetic torqueodq(k+1)：
vodq(k+1)＝[vod(k+1),voq(k+1)]T；
(3.2) setting the limit value v of the output capacity of the frequency convertermaxTo v is to vodqThe magnitude of (k +1) is limited.
The output capacity of the frequency converter is expressed as vmaxV is this vmaxIs generally set to the magnitude of the maximum voltage vector that can be output when the six-phase motor is normally operated,
wherein v isdcThe voltage of a direct current bus connected with the frequency converter is represented;
the voltage component amplitude limiting process associated with electromagnetic torque generation is as follows:
optionally, the duty cycle modulation module is specifically configured to perform the following steps:
(4.1) converting the processed optimal voltage vector back to a phase coordinate system, determining a final optimal voltage vector, and determining the duty ratio D (k +1) of each phase bridge arm power device through the following formula;
And (4.2) obtaining a driving signal of the motor through modulation, inputting the driving signal into the six-phase frequency converter, and driving the motor to operate.
Generally, compared with the prior art, the above technical solution conceived by the present invention has the following beneficial effects:
1. according to the method, a method for enumerating the on-off state of the frequency converter in the traditional model prediction method is not adopted, a method for predicting the current value at the next moment in the step (2.1) and obtaining the minimum difference value between the instruction value and the actual value by partial differentiation in the step (2.3) is adopted instead, the characteristic that the calculated amount of a model prediction control algorithm is large under the fault-tolerant working condition is overcome, meanwhile, the problem of large current harmonic content is realized by combining the modulation process in the step (4.2), and the performance of steady-state operation of the motor under the fault-tolerant working condition is improved. The method has compatibility and is suitable for improving the control performance of any symmetrical and asymmetrical six-phase motor.
2. According to the method, the dynamic performance of a motor control system is ensured, meanwhile, the fact that the number of times of enumeration calculation of a traditional model prediction method is reduced to 1 time of operation in the method is achieved, the calculated amount of a processor is obviously reduced, the current harmonic content is obviously reduced by adopting a modulation duty ratio process which is not adopted in the traditional model prediction method, the motor control precision is improved, and the current quality of the motor under the fault-tolerant working condition is improved.
Drawings
FIG. 1 is a schematic flow chart of a fault-tolerant motor fault control method according to the present invention;
fig. 2 is a topological diagram of a motor control main circuit provided by the present invention, wherein: vdcDenotes a DC voltage source, idcRepresenting bus current, S1～12Representing power devices and their antiparallel diodes, M representing a six-phase machine, iabcuvRepresents the current of each phase of abcuv;
fig. 3 is a schematic control block diagram of a motor provided in the present invention, wherein:
indicating a motor speed command value, ωeWhich is indicative of the measured rotational speed,indicates a reference current command value, Di(k +1) represents the calculated duty ratio of each phase bridge arm at the time of k +1, Si(k +1) represents the calculated switching signal of each phase arm at the time of k +1, Id,Iq,Ix,Iy,I0Respectively representing the respective current signals i after coordinate transformationabcuvThe phase currents are shown as being,represents a differentiator;
FIG. 4 is a phase current waveform provided by the present invention, wherein FIG. 4(a) is a phase current waveform of a fault-tolerant regime of a six-phase motor under a conventional model prediction method, and FIG. 4(b) is a phase current waveform of a fault-tolerant regime of a six-phase motor under a method according to the present invention;
FIG. 5 is a phase current harmonic distortion analysis provided by the present invention, wherein FIG. 5(a) is a one-phase current harmonic distortion analysis of a six-phase motor fault-tolerant regime under a conventional model prediction method, and FIG. 5(b) is a one-phase current harmonic distortion analysis of a six-phase motor fault-tolerant regime under a method of the present invention;
fig. 6 is each current waveform under the decoupling coordinate system provided by the present invention, wherein fig. 6(a) is each current waveform under the decoupling coordinate system of the six-phase motor fault-tolerant working condition under the system model prediction method, and fig. 6 (b) is each current waveform under the decoupling coordinate system of the six-phase motor fault-tolerant working condition adopting the method of the present invention;
fig. 7 is a schematic structural diagram of a fault-tolerant fault control device for a motor according to the present invention.
Detailed Description
In order to make the objects, technical solutions and advantages of the present invention more apparent, the present invention is described in further detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative of the invention and are not intended to limit the invention. In addition, the technical features involved in the embodiments of the present invention described below may be combined with each other as long as they do not conflict with each other.
Fig. 1 is a schematic flow chart of a fault-tolerant fault control method for a motor according to the present invention, as shown in fig. 1, including the following steps:
(1) the method comprises the steps of collecting non-fault 5-phase current in a six-phase motor in a current sampling period, converting the 5-phase current into 5 current components in a decoupling coordinate system, wherein the 5 current components comprise alternating-axis and direct-axis current components related to the generation of electromagnetic torque of the motor and x-axis, y-axis and 0-axis current components unrelated to the generation of the electromagnetic torque, and one-phase current fault exists in the six-phase motor.
Specifically, the electrical angle of the motor is measured, and the phase current of the remaining normal phase of the motor is measured. The rotating speed of the motor, the measured rotating speed and the rotating speed instruction value of the motor are obtained through electric angle calculation, and the instruction value of the current is generated through a proportional-integral controller. The current components under the decoupling coordinate system (namely the direct axis d, the quadrature axis q, the x axis, the y axis and the 0 axis) are obtained by carrying out park transformation and Clark coordinate transformation on the sampled phase currents (hereinafter, the currents after the transformation processing are also referred to as current measurement values). The current components include orthogonal and direct axis current components associated with electromagnetic torque generation, and x-axis, y-axis and 0-axis current components associated with electromagnetic torque generation. And the current instruction value and the current value obtained by the sampling and coordinate transformation participate in the prediction of the current value at the next moment in the fault-tolerant model.
(2) And (3) predicting the phase current value of each phase of the motor in the next sampling period by using a mathematical model of the motor and the 5 current components in the decoupling coordinate system obtained in the step (1), and predicting the optimal voltage vector output by the motor controller in the next sampling period according to the instruction value of each phase current and the predicted phase current of each phase of the motor in the next sampling period.
Specifically, the optimal voltage vector output by the controller in the next sampling period is predicted by a method of obtaining a minimum value by partial differentiation of a difference value between a command value of the current of the control system and a predicted value in the next sampling period, and the voltage vector obtained by the step enables the motor system to approach the set command value at the fastest speed. The specific method comprises the following steps: and generating the command value of each phase current in the next sampling period by the current rotating speed and the rotating speed command value through a rotating speed controller PI loop, predicting the current at the next moment by the current measurement value at the current moment through a motor model, taking the difference value between the current at the next moment and the command value as a scalar function, and performing partial differentiation on each current variable of the function to calculate the minimum value of the scalar function. So that the solution of the minimum value of the scalar function is the optimal voltage vector output by the controller.
(3) And converting the optimal voltage vector into 5 voltage components under a decoupling coordinate system, wherein the 5 voltage components comprise alternating-axis voltage components and direct-axis voltage components related to the generation of the electromagnetic torque of the motor, and when the amplitude of the voltage vector synthesized by the alternating-axis voltage components and the direct-axis voltage components exceeds an amplitude limit value, limiting the amplitude of the voltage vector to the amplitude limit value, and keeping the vector direction of the voltage vector unchanged to obtain the final alternating-axis voltage component and the final direct-axis voltage component under the decoupling coordinate system.
Specifically, when the difference between the measured value of the motor current and the command value is large, the voltage vector obtained by partial differentiation may exceed the range of the output capability of the inverter, and therefore, the amplitude of the voltage vector needs to be limited. The specific method comprises the following steps: and (3) solving the voltage vector obtained by calculation in the step (2) into a voltage vector under a decoupling coordinate system, and dividing the voltage vector into an orthogonal-axis voltage component voltage and a direct-axis voltage component voltage which are related to the electromagnetic torque, and an x-axis voltage component, a y-axis voltage component and a 0-axis voltage component which are not related to the electromagnetic torque. When the vector amplitude of the voltage synthesized by the alternating-axis voltage and the direct-axis voltage exceeds the amplitude limiting value, the vector amplitude is limited to the amplitude limiting value, and the vector direction is kept unchanged; otherwise, the voltage vector is not processed. The x-axis, y-axis and 0-axis voltage components are not processed because they do not participate in the electromechanical energy conversion.
(4) And (4) converting the final alternating-axis voltage component and the final direct-axis voltage component under the decoupling coordinate system obtained in the step (3) back to the phase coordinate system, determining a final optimal voltage vector, converting the final optimal voltage vector into each phase duty ratio, and adjusting a driving signal of each phase voltage according to each phase duty ratio to drive the motor.
Specifically, the processed voltage vector is converted back to a phase coordinate system through inverse park conversion and inverse clark conversion, the voltage vector is converted into a duty ratio of each phase through a conversion formula, the duty ratio is modulated through a space vector pulse width modulation module (or a module based on any other modulation principle), and a driving signal is output to a frequency converter for motor driving.
The invention uses a six-phase motor mathematical model to calculate the required ideal voltage vector from the sampling current to the reference current, converts the voltage vector under a rotating coordinate system into a phase coordinate system through a reduced-order coordinate transformation matrix, calculates the duty ratio of each phase bridge arm, and combines a space vector modulation method to realize the drive control of the multi-phase motor under the fault-tolerant working condition.
Optionally, the step (1) comprises the steps of:
(1.1) measuring the electrical angle theta (k) of the motor and the phase current i of the remaining non-faulted phase of the motorabcuv(k) Setting the current sampling period as k time, phase current iabcuv(k) Component is Ia,Ib,Ic,Iu,IvWherein a, b, c, u and v respectively represent the windings of each non-fault phase;
(1.2) calculating the angular velocity ω from the electrical angle θ (k)e(k) Wherein e represents the rotation speed as an electrical angular speed;
(1.3) with the measured phase current iabcuv(k) Obtaining the current i under a decoupling coordinate system through park transformation and Clark transformationdqxy0(k) The decoupled current component is Id,Iq,Ix,Iy,I0In which Id,IqFor the direct and quadrature current components associated with the generation of electromagnetic torque, Ix,Iy,I0Is a current component that is not related to the generation of electromagnetic torque;
clark transformation matrix as TcAnd the park transformation matrix is TpWhere the index c represents a Clark transformation matrix, p represents a park transformation matrix, TcAnd TpRespectively, as follows:
wherein, I4An identity diagonal matrix representing 4x 4; the transformation from phase current to decoupled current is as follows:
[Id,Iq,Ix,Iy,I0]T＝Tp·Tc·[Ia,Ib,Ic,Iu,Iv]Twhere the superscript T represents the transpose of the matrix.
Optionally, the step (2) comprises the steps of:
(2.1) Current value i at the time of passing kdqxy0(k)、ωe(k) And a voltage equation in a mathematical model of the motor, predicting the current value i at the moment k +1dqxy0(k +1), setting the next sampling period as the moment of k + 1;
the voltage equation used to predict the current in the mathematical model of the motor is:
wherein v isd,vq,vx,vy,v0Are respectively voltage components in a decoupled coordinate system Id,Iq,Ix,Iy,I0Are each a component of the current, R, in the decoupled coordinate systemsThe value of the phase resistance is the value of the phase resistance,denotes the derivation of time t, Ld,Lq,Lx,Ly,L0Respectively, inductance values of the phases, psi, in the decoupled coordinate systemmIs a permanent magnet flux linkage of the motor;
discretizing the voltage equation by using a forward Euler method
Where x is any variable to be discretized, TsFor the sampling period, x (k) represents the value of the variable at the k-th time, x (k +1) represents the value of the variable at the k +1 time, and the discretized equation is expressed as:
idqxy0(k+1)＝Aidqxy0(k)+BU(k)+C
wherein idqxy0(k+1)＝[Ia(k+1),Ib(k+1),Ic(k+1),Iu(k+1),Iv(k+1)]T， idqxy0(k)＝[Ia(k),Ib(k),Ic(k),Iu(k),Iv(k)]T，U(k)＝[vd,vq,vx,vy,v0]T(ii) a A. B, C are all coefficients of an equation, specifically:
i obtained by the above calculationdqxy0The (k +1) is the predicted current value at the time k + 1.
(2.2) generating the instruction value of the phase current by the rotating speed instruction value and the rotating speed measurement value through a motor rotating speed proportional-integral controller loop
(2.3) applying the current i at the next momentdqxy0(k +1) and instruction valueThe difference value between them is used as a scalar function g (k +1), and the partial differentiation is made for each current variable of said function, and the minimum value of scalar function is calculated, so that the obtained solution vo(k +1) is the optimal voltage vector;
wherein g represents a scalar function, and g (k +1) represents the value of this scalar function at time k + 1;
the process of solving the partial derivative of the voltage component under each decoupling coordinate system at the moment k +1 for the scalar function g is as follows:
the solution to this equation is expressed as:
vo(k+1)＝[vod(k+1),voq(k+1),vox(k+1),voy(k+1),vo0(k+1)]T，
wherein the subscript o indicates that the calculated voltage vector is the optimum voltage vector, vod(k+1),voq(k+1),vox(k+1),voy(k+1),vo0And (k +1) are respectively the optimal voltage components under the decoupling coordinate system at the moment of k + 1.
Optionally, the step (3) comprises the steps of:
(3.1) by pairing voThe voltage of each phase (k +1) is separated under a decoupling coordinate system through park transformation and Clark coordinate transformation to obtain a component v participating in the generation of electromagnetic torqueodq(k+1)：
vodq(k+1)＝[vod(k+1),voq(k+1)]T；
(3.2) setting the limit value v of the output capacity of the frequency convertermaxTo v is to vodqThe magnitude of (k +1) is limited.
The output capacity of the frequency converter is expressed as vmaxV is this vmaxIs generally set to the magnitude of the maximum voltage vector that can be output when the six-phase motor is normally operated,
wherein v isdcThe voltage of a direct current bus connected with the frequency converter is represented;
the voltage component amplitude limiting process associated with electromagnetic torque generation is as follows:
optionally, the step (4) comprises the steps of:
(4.1) converting the processed optimal voltage vector back to a phase coordinate system, determining a final optimal voltage vector, and determining the duty ratio D (k +1) of each phase bridge arm power device through the following formula;
And (4.2) obtaining a driving signal of the motor through modulation, inputting the driving signal into the six-phase frequency converter, and driving the motor to operate.
FIG. 2 is a topological diagram of a main circuit of the motor control provided by the present invention, VdcIs the bus voltage idcFor bus current, S1～12Is a power device of each phase bridge arm, C is a bus capacitor, ia,ib,ic,iu,iv,iwFor phase current of each phase arm, wherein iwThe phase current of the failed phase has a current value of 0, and the circle marked with M represents the motor.
FIG. 3 is a schematic diagram of a motor control scheme according to the present invention, showing a rotational speed command value
And actual value of rotation speed omegaeThe difference value of (a) is used for generating a current instruction value through the action of a proportional-integral controllerThe command value enters a fault-tolerant prediction model and passes through a motor mathematical model and a current sampling value iabcuv(k) Rotational speed omegae(k) And the rotation angle theta (k) predicts the optimal voltage vector at the next moment, and obtains a duty ratio vector D acting on each phase through links such as amplitude limiting, coordinate transformation, duty ratio calculation and the likei(k +1), (i ═ a, b, c, u, v), by the modulating action of the modulating module, most preferablySwitch drive signal S for lifetime power devicesi(k +1), (i ═ a, b, c, u, v), driving the six-phase motor to run. When the current of the motor is sampled, the current is directly sampled to obtain iabcuv(k) Then obtaining i by park transformation and Clark transformationdqxy0(k) The current value under the decoupling coordinate system, wherein the work flow of the whole device is as follows:
1. measuring the electrical angle theta (k) of the motor and the residual normal phase current i of the motorabcuv(k) Calculating the angular velocity ωe(k) In that respect Generating reference values of phase currents from the rotational speed command values and the rotational speed measurement values by means of a rotational speed controller PI loop
2. Using measured phase current iabcuv(k) Obtaining the current i under a decoupling coordinate system through park transformation and Clark transformationdqxy0(k) Predicting the current value i at the k +1 moment by a voltage equation in a mathematical model of the motordqxy0(k + 1). The current i at the next momentdqxy0(k +1) and instruction value
The difference value between them is used as a scalar function g (k +1), and the partial differentiation is made for each current variable of said function, and the minimum value of scalar function is calculated, so that the obtained solution vo(k +1) is the optimum voltage vector.
3. By pairing voV is obtained in (k +1) by participating in the separation of the electromagnetic torque generation componentodq(k +1), setting the limit amplitude v of the output capacity of the frequency convertermaxTo v is to vodqThe magnitude of (k +1) is limited.
4. And obtaining the duty ratio D (k +1) of each phase of bridge arm power device by the processed optimal voltage vector through a calculation formula. And then, a driving signal of the motor is obtained through space vector pulse width modulation (or modulation based on any other modulation principle), and is input into the six-phase frequency converter to drive the motor to operate.
The specific steps can refer to the method embodiment shown in fig. 1, and are not described herein again.
In a specific example, the embodiment of the present invention is implemented by using a surface-mounted permanent magnet synchronous motor, and compared with a conventional model predictive control algorithm, at a rated rotation speed point (100rpm), a current waveform is significantly improved, and a phase current waveform is compared as shown in fig. 4(a) and fig. 4(b), it can be observed that the sine degree of the phase current waveform obtained by the motor fault-tolerant fault control method or device provided by the embodiment of the present invention is significantly improved, and a curve is smoother.
Fig. 5 shows a Total Harmonic Distortion (THD) analysis of phase current, and compared with the conventional method, the fault-tolerant fault control method provided by the present invention reduces the Total Harmonic Distortion of current from 19.35% to 5.96%. Therefore, the invention obviously reduces the current harmonic content of the fault motor, improves the control precision of the motor and improves the current quality of the motor under the fault-tolerant working condition.
The current waveform under the decoupling coordinate system is shown in fig. 6, and compared with the traditional method, the method has the advantages that the improvement of the current waveform is very obvious, the burrs of the current waveform are obviously reduced, and the curve is smoother. Therefore, compared with the traditional method, the novel multiphase motor fault-tolerant working condition current harmonic suppression method based on the model predictive control algorithm provided by the invention has the advantages that the harmonic wave is obviously reduced, the control performance is obviously improved, and the method is an efficient and practical control method.
The invention provides a six-phase motor fault-tolerant working condition current harmonic suppression method based on a model predictive control algorithm, which is suitable for the condition that a six-phase motor control system has an open-circuit fault in a phase winding. The method is based on model prediction control, an optimal voltage vector adopted at each sampling moment is predicted through a mathematical model under the fault-tolerant working condition of the motor, and then the duty ratio of each bridge arm of the six-phase inverter is output through a pulse width modulation current control method. The invention obviously reduces the phase current harmonic under the fault-tolerant working condition of the open-circuit fault of the one-phase winding of the six-phase motor without influencing the dynamic performance of the motor, and has obvious effects of improving the output current quality of the motor, reducing the copper consumption of the motor and improving the control performance of a motor system.
Fig. 7 is a schematic structural diagram of a fault-tolerant fault control apparatus for a motor according to the present invention, as shown in fig. 7, including: the device comprises a phase current acquisition unit, an optimal voltage vector determination unit, a voltage vector amplitude limiting unit and a duty ratio modulation module.
The phase current acquisition unit is used for acquiring non-fault 5-phase current in the six-phase motor in the current sampling period and converting the 5-phase current into 5 current components in a decoupling coordinate system, wherein the 5 current components comprise alternating-axis and direct-axis current components related to the generation of electromagnetic torque of the motor and x-axis, y-axis and 0-axis current components unrelated to the generation of the electromagnetic torque, and one-phase current fault exists in the six-phase motor.
And the optimal voltage vector determining unit is used for predicting the phase current value of each phase of the motor in the next sampling period according to the mathematical model of the motor and the 5 current components of the decoupling coordinate system acquired by the phase current acquiring unit, and predicting the optimal voltage vector output by the motor controller in the next sampling period according to the instruction value of each phase current and the predicted phase current of each phase of the motor in the next sampling period.
And the voltage vector amplitude limiting unit is used for converting the optimal voltage vector into 5 voltage components under a decoupling coordinate system, wherein the 5 voltage components comprise alternating-axis voltage components and direct-axis voltage components related to the generation of the electromagnetic torque of the motor, and when the amplitude of the voltage vector synthesized by the alternating-axis voltage components and the direct-axis voltage components exceeds an amplitude limit value, the amplitude of the voltage vector is limited to the amplitude limit value, and the vector direction of the voltage vector is kept unchanged, so that the final alternating-axis voltage component and the final direct-axis voltage component under the decoupling coordinate system are obtained.
And the duty ratio modulation module is used for converting the final alternating-axis voltage component and the final direct-axis voltage component under the decoupling coordinate system back to the phase coordinate system, determining a final optimal voltage vector, converting the final optimal voltage vector into each phase duty ratio, and adjusting the driving signal of each phase voltage according to each phase duty ratio to drive the motor.
It is understood that the apparatus may further include more or less components, and the functions of the components may refer to the detailed description of the embodiment of the method shown in fig. 1, which is not repeated herein.
It will be understood by those skilled in the art that the foregoing is only a preferred embodiment of the present invention, and is not intended to limit the invention, and that any modification, equivalent replacement, or improvement made within the spirit and principle of the present invention should be included in the scope of the present invention.
Claims (8)
1. A fault-tolerant fault control method for a motor is characterized by comprising the following steps:
(1) acquiring non-fault 5-phase current in a six-phase motor in a current sampling period, and converting the 5-phase current into 5 current components in a decoupling coordinate system, wherein the 5 current components comprise alternating-axis and direct-axis current components related to the generation of electromagnetic torque of the motor and x-axis, y-axis and 0-axis current components unrelated to the generation of the electromagnetic torque, and one-phase current fault exists in the six-phase motor;
(2) predicting the phase current value of each phase of the motor in the next sampling period by using a mathematical model of the motor and 5 current components in the decoupling coordinate system obtained in the step (1), and predicting the optimal voltage vector output by the motor controller in the next sampling period according to the instruction value of each phase current and the predicted phase current of each phase of the motor in the next sampling period;
the step (2) comprises the following steps:
(2.1) Current value i at the time of passing kdqxy0(k)、ωe(k) And a voltage equation in a mathematical model of the motor, predicting the current value i at the moment k +1dqxy0(k +1), setting the next sampling period as the moment of k + 1;
the voltage equation used to predict the current in the mathematical model of the motor is:
wherein v isd,vq,vx,vy,v0Are respectively voltage components in a decoupled coordinate system Id,Iq,Ix,Iy,I0Are each a component of the current, R, in the decoupled coordinate systemsThe value of the phase resistance is the value of the phase resistance,
denotes the derivation of time t, Ld,Lq,Lx,Ly,L0Respectively, inductance values of the phases, psi, in the decoupled coordinate systemmIs a permanent magnet flux linkage of the motor;
discretizing the voltage equation by using a forward Euler method
Where x is any variable to be discretized, TsFor the sampling period, x (k) represents the value of the variable at the k-th time, x (k +1) represents the value of the variable at the k +1 time, and the discretized equation is expressed as:
idqxy0(k+1)＝Aidqxy0(k)+BU(k)+C
wherein idqxy0(k+1)＝[Id(k+1),Iq(k+1),Ix(k+1),Iy(k+1),I0(k+1)]T，idqxy0(k)＝[Id(k),Iq(k),Ix(k),Iy(k),I0(k)]T，U(k)＝[vd,vq,vx,vy,v0]T(ii) a A. B, C are all coefficients of an equation, specifically:
i obtained by the above calculationdqxy0(k +1) is the predicted current value at the time of k + 1;
(2.2) generating the instruction value of the phase current by the rotating speed instruction value and the rotating speed measurement value through a motor rotating speed proportional-integral controller loop
(2.3) applying the current i at the next momentdqxy0(k +1) and instruction value
The difference between them is used as a scalar function, partial differentiation is carried out on each current variable of the function, the minimum value of the scalar function is calculated, and the solution v is obtainedo(k +1) is the optimal voltage vector;
wherein g represents a scalar function, and g (k +1) represents the value of this scalar function at time k + 1;
the process of solving the partial derivative of the voltage component under each decoupling coordinate system at the moment k +1 for the scalar function g is as follows:
the solution to this equation is expressed as:
vo(k+1)＝[vod(k+1),voq(k+1),vox(k+1),voy(k+1),vo0(k+1)]T，
wherein the subscript o indicates that the calculated voltage vector is the optimum voltage vector, vod(k+1),voq(k+1),vox(k+1),voy(k+1),vo0(k +1) are each under the decoupling coordinate system at the moment of k +1An optimum voltage component;
(3) converting the optimal voltage vector into 5 voltage components under a decoupling coordinate system, wherein the 5 voltage components comprise alternating-axis voltage components and direct-axis voltage components related to the generation of the electromagnetic torque of the motor, and when the amplitude of the voltage vector synthesized by the alternating-axis voltage components and the direct-axis voltage components exceeds an amplitude limit value, limiting the amplitude of the voltage vector to the amplitude limit value, and keeping the vector direction of the voltage vector unchanged to obtain final alternating-axis voltage components and direct-axis voltage components under the decoupling coordinate system;
(4) and (4) converting the final alternating-axis voltage component and the final direct-axis voltage component under the decoupling coordinate system obtained in the step (3) back to the phase coordinate system, determining a final optimal voltage vector, converting the final optimal voltage vector into each phase duty ratio, and adjusting a driving signal of each phase voltage according to each phase duty ratio to drive the motor.
2. A method of controlling a fault-tolerant motor according to claim 1, wherein said step (1) comprises the steps of:
(1.1) measuring the electrical angle theta (k) of the motor and the phase current i of the remaining non-faulted phase of the motorabcuv(k) Setting the current sampling period as k time, phase current iabcuv(k) Component is Ia,Ib,Ic,Iu,IvWherein a, b, c, u and v respectively represent the windings of each non-fault phase;
(1.2) calculating the angular velocity ω from the electrical angle θ (k)e(k) Wherein e represents the rotation speed as an electrical angular speed;
(1.3) with the measured phase current iabcuv(k) Obtaining the current i under a decoupling coordinate system through park transformation and Clark transformationdqxy0(k) The decoupled current component is Id,Iq,Ix,Iy,I0In which Id,IqFor the direct and quadrature current components associated with the generation of electromagnetic torque, Ix,Iy,I0Is a current component that is not related to the generation of electromagnetic torque;
clark transformation matrix as TcAnd the park transformation matrix is TpWherein the subscript c represents the Clark transformation matrix and p represents the park transformationArray, TcAnd TpRespectively, as follows:
wherein, I4An identity diagonal matrix representing 4x 4; the transformation from phase current to decoupled current is as follows:
[Id,Iq,Ix,Iy,I0]T＝Tp·Tc·[Ia,Ib,Ic,Iu,Iv]Twhere the superscript T represents the transpose of the matrix.
3. A method of controlling a fault-tolerant motor according to claim 2, wherein said step (3) comprises the steps of:
(3.1) by pairing voThe voltage of each phase (k +1) is separated under a decoupling coordinate system through park transformation and Clark coordinate transformation to obtain a component v participating in the generation of electromagnetic torqueodq(k+1)：
vodq(k+1)＝[vod(k+1),voq(k+1)]T；
(3.2) setting the limit value v of the output capacity of the frequency convertermaxTo v is to vodqThe amplitude of (k +1) is limited;
the output capacity of the frequency converter is expressed as vmaxV is this vmaxThe magnitude of the maximum voltage vector that can be output when the six-phase motor is normally operated is set,
wherein v isdcThe voltage of a direct current bus connected with the frequency converter is represented;
the voltage component amplitude limiting process associated with electromagnetic torque generation is as follows:
4. a method of fault-tolerant fault control of an electric motor according to claim 3, characterized in that said step (4) comprises the steps of:
(4.1) converting the processed optimal voltage vector back to a phase coordinate system, determining a final optimal voltage vector, and determining the duty ratio D (k +1) of each phase bridge arm power device through the following formula;
And (4.2) obtaining a driving signal of the motor through modulation, inputting the driving signal into the six-phase frequency converter, and driving the motor to operate.
5. A fault-tolerant fault control apparatus for an electric motor, comprising:
the phase current acquisition unit is used for acquiring non-fault 5-phase current in the six-phase motor in the current sampling period and converting the 5-phase current into 5 current components in a decoupling coordinate system, wherein the 5 current components comprise alternating-axis and direct-axis current components related to the generation of electromagnetic torque of the motor and x-axis, y-axis and 0-axis current components unrelated to the generation of the electromagnetic torque, and one-phase current fault exists in the six-phase motor;
the optimal voltage vector determining unit is used for predicting the phase current value of each phase of the motor in the next sampling period according to the mathematical model of the motor and 5 current components in the decoupling coordinate system acquired by the phase current acquiring unit, and predicting the optimal voltage vector output by the motor controller in the next sampling period according to the instruction value of each phase current and the predicted phase current of each phase of the motor in the next sampling period; the method is specifically used for executing the following steps: (2.1) passing the current at time kValue idqxy0(k)、ωe(k) And a voltage equation in a mathematical model of the motor, predicting the current value i at the moment k +1dqxy0(k +1), setting the next sampling period as the moment of k + 1; the voltage equation used to predict the current in the mathematical model of the motor is:
wherein v isd,vq,vx,vy,v0Are respectively voltage components in a decoupled coordinate system Id,Iq,Ix,Iy,I0Are each a component of the current, R, in the decoupled coordinate systemsThe value of the phase resistance is the value of the phase resistance,
denotes the derivation of time t, Ld,Lq,Lx,Ly,L0Respectively, inductance values of the phases, psi, in the decoupled coordinate systemmIs a permanent magnet flux linkage of the motor; discretizing the voltage equation by using a forward Euler method
Where x is any variable to be discretized, TsFor the sampling period, x (k) represents the value of the variable at the k-th time, x (k +1) represents the value of the variable at the k +1 time, and the discretized equation is expressed as:
idqxy0(k+1)＝Aidqxy0(k)+BU(k)+C
wherein idqxy0(k+1)＝[Id(k+1),Iq(k+1),Ix(k+1),Iy(k+1),I0(k+1)]T，idqxy0(k)＝[Id(k),Iq(k),Ix(k),Iy(k),I0(k)]T，U(k)＝[vd,vq,vx,vy,v0]T(ii) a A. B, C are all coefficients of an equation, specifically:
i obtained by the above calculationdqxy0(k +1) is the predicted current value at the time of k + 1; (2.2) generating the instruction value of the phase current by the rotating speed instruction value and the rotating speed measurement value through a motor rotating speed proportional-integral controller loop
Wherein the number indicates that the value is a command value,
a matrix for representing the decoupled current composition; (2.3) applying the current i at the next momentdqxy0(k +1) and instruction value
The difference between them is used as a scalar function, partial differentiation is carried out on each current variable of the function, the minimum value of the scalar function is calculated, and the solution v is obtainedo(k +1) is the optimal voltage vector;
wherein g represents a scalar function, and g (k +1) represents the value of this scalar function at time k + 1; the process of solving the partial derivative of the voltage component under each decoupling coordinate system at the moment k +1 for the scalar function g is as follows:
the solution to this equation is expressed as:
vo(k+1)＝[vod(k+1),voq(k+1),vox(k+1),voy(k+1),vo0(k+1)]T，
wherein the subscript o indicates that the calculated voltage vector is the optimum voltage vector, vod(k+1),voq(k+1),vox(k+1),voy(k+1),vo0(k +1) are respectively the optimal voltage components under the decoupling coordinate system at the moment of k + 1;
the voltage vector amplitude limiting unit is used for converting the optimal voltage vector into 5 voltage components under a decoupling coordinate system, wherein the 5 voltage components comprise alternating-axis voltage components and direct-axis voltage components related to the generation of the electromagnetic torque of the motor, and when the amplitude of the voltage vector synthesized by the alternating-axis voltage components and the direct-axis voltage components exceeds an amplitude limit value, the amplitude of the voltage vector is limited to the amplitude limit value, and the vector direction of the voltage vector is kept unchanged, so that the final alternating-axis voltage component and the final direct-axis voltage component under the decoupling coordinate system are obtained;
and the duty ratio modulation module is used for converting the final alternating-axis voltage component and the final direct-axis voltage component under the decoupling coordinate system back to the phase coordinate system, determining a final optimal voltage vector, converting the final optimal voltage vector into each phase duty ratio, and adjusting the driving signal of each phase voltage according to each phase duty ratio to drive the motor.
6. The fault-tolerant fault control device of an electric motor according to claim 5, wherein the phase current acquisition unit is specifically configured to perform the following steps:
(1.1) measuring the electrical angle theta (k) of the motor and the phase current i of the remaining non-faulted phase of the motorabcuv(k) Setting the current sampling period as k time, phase current iabcuv(k) Component is Ia,Ib,Ic,Iu,IvWherein a, b, c, u and v respectively represent the windings of each non-fault phase;
(1.2) calculating the angular velocity ω from the electrical angle θ (k)e(k) Wherein e represents the rotation speed as an electrical angular speed;
(1.3) with the measured phase current iabcuv(k) Obtaining the current i under a decoupling coordinate system through park transformation and Clark transformationdqxy0(k) The decoupled current component is Id,Iq,Ix,Iy,I0In which Id,IqFor the direct and quadrature current components associated with the generation of electromagnetic torque, Ix,Iy,I0Is a current component that is not related to the generation of electromagnetic torque;
clark transformation matrix as TcAnd the park transformation matrix is TpWhere the index c represents a Clark transformation matrix, p represents a park transformation matrix, TcAnd TpRespectively, as follows:
wherein, I4An identity diagonal matrix representing 4x 4; the transformation from phase current to decoupled current is as follows:
[Id,Iq,Ix,Iy,I0]T＝Tp·Tc·[Ia,Ib,Ic,Iu,Iv]Twhere the superscript T represents the transpose of the matrix.
7. The motor fault-tolerant fault control apparatus of claim 5, wherein the voltage vector limiting unit is specifically configured to perform the following steps:
(3.1) by pairing voThe voltage of each phase (k +1) is separated under a decoupling coordinate system through park transformation and Clark coordinate transformation to obtain a component v participating in the generation of electromagnetic torqueodq(k+1)：
vodq(k+1)＝[vod(k+1),voq(k+1)]T；
(3.2) setting the limit value v of the output capacity of the frequency convertermaxTo v is to vodqThe amplitude of (k +1) is limited;
the output capacity of the frequency converter is expressed as vmaxV is this vmaxThe magnitude of the maximum voltage vector that can be output when the six-phase motor is normally operated is set,
wherein v isdcThe voltage of a direct current bus connected with the frequency converter is represented;
the voltage component amplitude limiting process associated with electromagnetic torque generation is as follows:
8. the fault-tolerant fault control device of claim 7, wherein the duty cycle modulation module is specifically configured to perform the following steps:
(4.1) converting the processed optimal voltage vector back to a phase coordinate system, determining a final optimal voltage vector, and determining the duty ratio D (k +1) of each phase bridge arm power device through the following formula;
And (4.2) obtaining a driving signal of the motor through modulation, inputting the driving signal into the six-phase frequency converter, and driving the motor to operate.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN201710820605.9A CN107623479B (en) | 2017-09-13 | 2017-09-13 | Fault-tolerant fault control method and device for motor |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN201710820605.9A CN107623479B (en) | 2017-09-13 | 2017-09-13 | Fault-tolerant fault control method and device for motor |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN107623479A CN107623479A (en) | 2018-01-23 |
| CN107623479B true CN107623479B (en) | 2020-01-21 |
Family
ID=61089536
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN201710820605.9A Expired - Fee Related CN107623479B (en) | 2017-09-13 | 2017-09-13 | Fault-tolerant fault control method and device for motor |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN107623479B (en) |
Families Citing this family (9)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP7053335B2 (en) * | 2018-03-28 | 2022-04-12 | 日立Astemo株式会社 | Motor control device, electric vehicle |
| CN110350833B (en) * | 2018-04-04 | 2021-05-25 | 中车株洲电力机车研究所有限公司 | Six-phase alternating current motor system and control method and control device thereof |
| CN110554313B (en) * | 2018-05-31 | 2021-10-19 | 蜂巢传动系统(江苏)有限公司保定研发分公司 | Current fault detection method and device |
| CN108988713B (en) * | 2018-09-14 | 2019-07-05 | 上海寰晟电力能源科技有限公司 | Double three-phase PMSG multiphase open-circuit fault error-tolerant operation control methods and system |
| CN109842343B (en) * | 2019-02-18 | 2020-09-25 | 清华大学 | Fault-tolerant operation control method and device for flywheel energy storage system based on twelve-phase motor |
| CN109981017B (en) * | 2019-04-30 | 2022-10-18 | 东南大学 | Fault-tolerant control system and method for two-unit same-phase permanent magnet synchronous motor under open-circuit fault |
| CN110690839B (en) * | 2019-09-29 | 2021-05-25 | 大连海事大学 | Topological control method for six-phase permanent magnet fault-tolerant rim propulsion motor |
| CN113311278A (en) * | 2021-07-01 | 2021-08-27 | 哈尔滨理工大学 | Six-phase motor driving system fault diagnosis method based on vector space decoupling |
| CN114604103B (en) * | 2022-03-17 | 2024-02-02 | 威睿电动汽车技术(宁波)有限公司 | Active heating method, device, equipment, storage medium and program product for motor |
Citations (3)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN105162380A (en) * | 2015-10-13 | 2015-12-16 | 哈尔滨工业大学 | Six-phase permanent-magnet synchronous motor model predictive control method |
| CN106505927A (en) * | 2016-12-26 | 2017-03-15 | 西南交通大学 | A kind of five-phase PMSM finite aggregate model prediction current control method |
| CN107069732A (en) * | 2017-04-18 | 2017-08-18 | 西南交通大学 | The active filter harmonic current compensation method predicted based on minimum current error model |
-
2017
- 2017-09-13 CN CN201710820605.9A patent/CN107623479B/en not_active Expired - Fee Related
Patent Citations (3)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN105162380A (en) * | 2015-10-13 | 2015-12-16 | 哈尔滨工业大学 | Six-phase permanent-magnet synchronous motor model predictive control method |
| CN106505927A (en) * | 2016-12-26 | 2017-03-15 | 西南交通大学 | A kind of five-phase PMSM finite aggregate model prediction current control method |
| CN107069732A (en) * | 2017-04-18 | 2017-08-18 | 西南交通大学 | The active filter harmonic current compensation method predicted based on minimum current error model |
Non-Patent Citations (1)
| Title |
|---|
| Reduction of Unbalanced Axial Magnetic Force in Post-fault Operation of a Novel Six-phase Double-stator Axial Flux PM Machine Using Model Predictive Control;Hanxiao Lu等;《IEEE》;20170216;第1-8页 * |
Also Published As
| Publication number | Publication date |
|---|---|
| CN107623479A (en) | 2018-01-23 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| CN107623479B (en) | Fault-tolerant fault control method and device for motor | |
| Tao et al. | Simplified fault-tolerant model predictive control for a five-phase permanent-magnet motor with reduced computation burden | |
| CN106936356B (en) | Vector screening and duty ratio combined motor model prediction control system and method | |
| Yim et al. | Modified current control schemes for high-performance permanent-magnet AC drives with low sampling to operating frequency ratio | |
| WO2022110277A1 (en) | Model predictive decomposition control method and apparatus for open-winding five-phase permanent magnet synchronous motor | |
| CN108631672B (en) | Permanent magnet synchronous motor prediction flux linkage control method considering optimal duty ratio modulation | |
| WO2011040159A1 (en) | Control device for an electric motor drive device | |
| CN109839830B (en) | Power level analog control method and device of three-phase alternating current motor | |
| CN106655936B (en) | It is a kind of to lack rare-earth permanent-magnet electric machine zero-sequence current inhibition control system and method | |
| CN107294447B (en) | Self-adaptive maximum torque current ratio control device of permanent magnet synchronous motor | |
| CN110086383B (en) | Model prediction control method and device for twelve-phase driving system | |
| Zhang et al. | Torque ripple RMS minimization in model predictive torque control of PMSM drives | |
| CN108448971B (en) | Control system of brushless doubly-fed generator and model prediction current control method | |
| CN105743330A (en) | Single support vector machine (SVM) simplified modulation algorithm of dual inverters | |
| Chi et al. | A ripple suppression of sensorless FOC of PMSM electrical drive system based on MRAS | |
| CN105871278A (en) | Direct feature control-based novel salient-pole permanent magnet synchronous motor control method | |
| Kakodia et al. | A comparative study of DFOC and IFOC for IM drive | |
| CN109600095B (en) | Phase failure fault-tolerant control system and method of permanent magnet synchronous motor based on four-bridge-arm inverter | |
| JP3939481B2 (en) | AC motor control device | |
| CN103532461B (en) | A kind of for steadily controlling the device that the little torque conditions of permagnetic synchronous motor low speed switches | |
| Chikh et al. | A novel fixed-switching-frequency DTC for PMSM drive with low torque and flux ripple based on Sinusoidal Pulse With Modulation and predictive controller | |
| CN115913038A (en) | Model prediction control method for double three-phase permanent magnet synchronous motor | |
| CN113258837B (en) | Robust model prediction current control method and device for permanent magnet synchronous motor | |
| Dang et al. | Model Predictive Direct Torque Control for PMSM Drives in M–T Frame | |
| CN113131829B (en) | Optimization control method and control system for harmonic loss suppression efficiency of five-phase induction motor |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| PB01 | Publication | ||
| PB01 | Publication | ||
| SE01 | Entry into force of request for substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| GR01 | Patent grant | ||
| GR01 | Patent grant | ||
| CF01 | Termination of patent right due to non-payment of annual fee | ||
| CF01 | Termination of patent right due to non-payment of annual fee |
Granted publication date: 20200121 Termination date: 20200913 |