CN117176258B

CN117176258B - Digital modulation method and device

Info

Publication number: CN117176258B
Application number: CN202311448545.4A
Authority: CN
Inventors: 樊玉成; 顾学恒; 胥国祥; 许人东
Original assignee: Jiangsu Hengtong Huahai Technology Co ltd; Suzhou University
Current assignee: Jiangsu Hengtong Huahai Technology Co ltd; Suzhou University
Priority date: 2023-11-02
Filing date: 2023-11-02
Publication date: 2024-01-26
Anticipated expiration: 2043-11-02
Also published as: CN117176258A

Abstract

The invention relates to the technical field of optical fiber communication, in particular to a digital modulation method and device. The digital modulation method comprises the following steps: before a high-frequency carrier signal is input to an IQ modulator at a transmitting end, a polarization control algorithm in a singlechip is utilized to coarsely adjust the bias voltage of the IQ modulator, the bias point is approximately locked near a linear point, then data is added to carry out fine adjustment, and the stability of the working state of the IQ modulator is ensured by detecting the direct current component and the alternating current component of optical power; and carrying out self-adaptive equalization processing by using a coherent demodulation algorithm at a receiving end, and carrying out frequency offset recovery and phase recovery on an output result of the coherent demodulation algorithm by using a carrier recovery algorithm. The polarization control algorithm ensures the polarization stability of the optical signal, so that the noise acceleration of the optical signal in transmission is alleviated, and the coherent demodulation algorithm and the carrier recovery algorithm realize the demodulation and correct data recovery of the signal, thereby effectively improving the transmission distance of the signal in light.

Description

Digital modulation method and device

Technical Field

The invention relates to the technical field of optical fiber communication, in particular to a digital modulation method and device.

Background

The optical communication technology is a communication technology for transmitting data and information through an optical fiber or free space by using light as a carrier for information transmission. The method has the advantages of high bandwidth, low loss, interference resistance and the like, and becomes one of the most important technologies in the modern communication field.

In the prior art, the optical communication uses a mature erbium-doped fiber amplifier and a wavelength division multiplexing technology, so that the problems of relay transmission and capacity expansion of the optical communication are simply and effectively solved. However, with the explosion of the mobile internet, the data traffic of the communication network increases rapidly, and the pressure faced by the backbone network increases rapidly. However, the potential of erbium-doped fiber amplifiers and wavelength division multiplexing technology is smaller and smaller, and the anti-interference performance of polarization modulation on optical signals is stronger, so that the distortion and the bit error rate of the signals can be effectively reduced.

The polarization state of an optical signal may change during transmission, for example, due to nonlinear effects of an optical fiber, chromatic dispersion, and other factors, resulting in phenomena such as polarization rotation and cross coupling. These effects can degrade the polarization retention of the optical signal, thereby limiting the transmission distance. Therefore, in order to ensure the polarization state of the optical signal, it is necessary to improve the accuracy of the bias control in the IQ modulator, thereby realizing a longer transmission distance.

Disclosure of Invention

Therefore, the technical problem to be solved by the invention is to overcome the problem that the transmission capability of optical communication needs to be improved in the prior art.

In order to solve the technical problems, the present invention provides a digital modulation method, including:

at a transmitting end, overlapping digital signals subjected to spread spectrum processing by different users, converting the overlapped digital signals into analog signals, performing radio frequency modulation, and modulating the analog signals into high-frequency carrier signals;

before the high-frequency carrier signal is input to the IQ modulator, the bias voltage of the IQ modulator is coarsely regulated by utilizing a polarization control algorithm in the singlechip, and the bias voltage is regulated within a preset bias range according to the minimum value and the maximum value of the optical power so as to lock the bias point near the linear point;

after the bias voltage is subjected to rough adjustment, a high-frequency carrier signal is input into an IQ modulator, the bias voltage is continuously tracked by utilizing a polarization control algorithm in a singlechip, the bias voltage which is increased and reduced firstly by one stepping value is used for obtaining direct current components of two different optical powers and alternating current components of two different optical powers, and the bias voltages corresponding to the smaller direct current components and the alternating current components are respectively taken as new bias voltages and are iterated continuously until the bias voltages are not changed;

The high-frequency carrier signal is converted into an optical signal through an IQ modulator, the power of the optical signal is amplified through a small signal-to-noise amplifier, and the optical signal enters an optical fiber for transmission after the completion;

at a receiving end, amplifying an optical signal transmitted by an optical fiber through a backward Raman amplifier, performing self-adaptive equalization processing by using a coherent demodulation algorithm, and performing frequency offset recovery and phase recovery on an output result of the coherent demodulation algorithm by using a carrier recovery algorithm to restore the output result to a high-frequency carrier signal;

the restored high-frequency carrier signals are demodulated into analog signals, then converted into digital signals, and the digital signals are despread, and mathematical signals of all users are separated from the mixed digital signals.

In one embodiment of the present invention, the method for coarsely adjusting the bias voltage of the IQ modulator by using a polarization control algorithm in a single chip microcomputer connected to the IQ modulator adjusts the bias voltage within a preset bias range according to the minimum value and the maximum value of the optical power so as to lock the bias point near the linear point, and specifically includes the steps of:

the single chip microcomputer respectively carries out quick scanning on bias voltages Vbias_XI and Vbias_XQ of an X polarized arm I path and an X polarized arm Q path, acquires bias voltages Vbias_XI and Vbias_XQ of the X polarized arm I path and the X polarized arm Q path which enable direct current components of signal light power received by the built-in diode to be maximum, and controls signals of the X polarized arm I path and the X polarized arm Q path to be positioned near a maximum power point;

After obtaining bias voltages vbias_xi and vbias_xq of an X-polarization arm I path and an X-polarization arm Q path which make the direct current component of the signal light power received by the built-in diode maximum, the singlechip rapidly scans the bias voltage vbias_xp of the X-polarization component to obtain bias voltage vbias_xp of the X-polarization component which makes the direct current component of the signal light power received by the built-in diode be the average value of the maximum value and the minimum value, and controls the X-polarization component signal to be located near a 90-degree phase point, wherein the bias voltage vbias_xp of the X-polarization component is the initial bias voltage of the X-polarization component;

after obtaining bias voltages vbias_xp of X polarization components enabling direct current components of signal light power received by the built-in diode to be maximum and minimum average values, the singlechip rapidly scans bias voltages vbias_xi and vbias_xq of an X polarization arm I path and an X polarization arm Q path to obtain bias voltages vbias_xi and vbias_xq of the X polarization arm I path and the X polarization arm Q path enabling the direct current components of the signal light power received by the built-in diode to be minimum, and controls signals of the X polarization arm I path and the X polarization arm Q path to be located near an optimal polarization point, wherein the bias voltages vbias_xi and vbias_xq of the X polarization arm I path and the X polarization arm Q path are initial bias voltages of the X polarization arm I path and initial bias voltages of the X polarization arm Q path;

The single chip microcomputer respectively carries out quick scanning on bias voltages Vbias_YI and Vbias_YQ of a Y polarization arm I path and a Y polarization arm Q path, acquires bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I path and the Y polarization arm Q path which enable direct current components of signal light power received by the built-in diode to be maximum, and controls signals of the Y polarization arm I path and the Y polarization arm Q path to be positioned near a maximum power point;

after obtaining bias voltages vbias_yi and vbias_yq of a Y polarization arm I path and a Y polarization arm Q path which make the direct current component of the signal light power received by the built-in diode maximum, the singlechip rapidly scans the bias voltage vbias_yp of the Y polarization component to obtain bias voltage vbias_yp of the Y polarization component which makes the direct current component of the signal light power received by the built-in diode be an average value of a maximum value and a minimum value, and controls the Y polarization component signal to be located near a 90-degree phase point, wherein the bias voltage vbias_yp of the Y polarization component is an initial bias voltage of the Y polarization component;

after obtaining the bias voltages vbias_yp of the Y polarization components, which make the direct current component of the signal light power received by the built-in diode be the average value of the maximum value and the minimum value, the singlechip rapidly scans the bias voltages vbias_yi and vbias_yq of the Y polarization arm I path and the Y polarization arm Q path to obtain the bias voltages vbias_yi and vbias_yq of the Y polarization arm I path and the Y polarization arm Q path, which make the direct current component of the signal light power received by the built-in diode be the minimum value, and controls the bias voltages vbias_yi and vbias_yq of the Y polarization arm I path and the Y polarization arm Q path to be near the optimal polarization point, and at this time, the bias voltages vbias_yi and vbias_yq of the Y polarization arm I path and the Y polarization arm Q path are the initial bias voltages of the Y polarization arm I path and the initial Y polarization arm Q path.

In one embodiment of the present invention, the high-frequency carrier signal is input to the IQ modulator, the polarization control algorithm in the single chip microcomputer continuously tracks the bias voltage, the bias voltage which is increased and decreased first by a step value obtains direct current components of two different optical powers and alternating current components of two different optical powers, and the bias voltages corresponding to the smaller direct current components and alternating current components are respectively taken as new bias voltages, and the iteration is continued until the bias voltages are not changed, and the specific steps include:

adding a high-frequency carrier signal for modulation to obtain optical signals output by an X polarization arm and a Y polarization arm;

the singlechip controls the initial bias voltage of the X polarization component to enable the stepping value of the initial bias voltage to be increased and then reduced, so as to obtain alternating current components of two different optical powers, and the bias voltage corresponding to the alternating current component with smaller power is taken as the bias voltage of the new X polarization component;

after the offset voltage of the new X polarization component is obtained, the singlechip controls the initial X polarization arm I offset voltage and the initial X polarization arm Q offset voltage to ensure that the stepping value of the initial X polarization arm I offset voltage is increased and then reduced to obtain direct current components of two optical powers with different front and back sizes, and the offset voltage corresponding to the smaller power direct current component is taken as the new X polarization arm I offset voltage and the new X polarization arm Q offset voltage;

The singlechip controls the initial bias voltage of the Y polarization component to enable the stepping value of the initial bias voltage to be increased and then reduced, so as to obtain alternating current components of two optical powers with different front and back sizes, and the bias voltage corresponding to the alternating current component with smaller power is taken as the bias voltage of the new Y polarization component;

after the offset voltage of the new Y polarization component is obtained, the singlechip controls the initial Y polarization arm I offset voltage and the initial Y polarization arm Q offset voltage to ensure that the stepping value of the initial Y polarization arm I offset voltage is increased and then reduced to obtain direct current components of two optical powers with different front and back sizes, and the offset voltage corresponding to the smaller power direct current component is taken as the new Y polarization arm I offset voltage and the new Y polarization arm Q offset voltage;

repeating the steps, and continuously iterating until the bias voltage of the X polarization arm I path, the bias voltage of the X polarization arm Q path, the bias voltage of the Y polarization arm I path and the bias voltage of the Y polarization arm Q path are not changed any more, determining the bias voltage as an optimal linear point, locking the bias voltage of the X polarization component of the bias voltage and the bias voltage of the Y polarization component to a 90-degree phase point, and keeping the quadrature phases of the I path and the Q path.

In one embodiment of the present invention, the adaptive equalization processing using a coherent demodulation algorithm includes: and eliminating the signal group delay by using an FIR filter, and then deciding and data decoding the signal with the eliminated group delay by using a decision device to output the original transmitting signal without carrier recovery.

In one embodiment of the present invention, the method uses an FIR filter to eliminate group delay, and the weighted square error cost function of the group delay model is:

，

wherein,for the passband of the FIR filter, < >>Is the stop band of the FIR filter, < >>Amplitude weights in the cost error function; />For the desired amplitude-frequency response of the channel, +.>For the desired group delay frequency response of the channel, +.>Is angular frequency;

as a channel frequency response function:

，

wherein,for group delay channel impulse response sequences: />，/>Is the channel filter length;

as a group delay frequency response function:

，

wherein,for the phase frequency characteristic function of the system->Representing a real part calculation;

discretizing the frequency, the weighted square error cost function becomes:

；

setting group delay channel impulse response sequenceThe weighted square error cost function is minimized, and the expression of the obtained group delay channel FIR filter model is as follows: />。

In one embodiment of the present invention, the performing frequency offset recovery and phase recovery on the output result of the coherent demodulation algorithm by adopting a carrier recovery algorithm includes: the frequency offset of the signal is compensated by using an M-th frequency offset estimation algorithm, and the phase of the signal is compensated by using a Viterbi-Viterbi phase estimation algorithm.

In one embodiment of the present invention, the M-th order frequency offset estimation algorithm includes a fourth order frequency offset estimation algorithm, and the specific steps include:

s1, removing phase damage introduced by laser linewidth:

setting the phase of the received signal to beWherein->Representing information phase +.>Representing the phase caused by the linewidth of the laser, +.>Representing noise phase;

for sampling value signalsConjugation and initial signal->Multiplying to obtain:

，

wherein the method comprises the steps ofIs a slowly varying signal, so that the front and back sample values are zero, and the laser linewidth can be eliminated after the step>Is a phase impairment caused by (a) a phase shift;

s2, removing residual modulation phases:

in the ideal case, the residual modulation phaseThe value of (2) is +.>Then->Is->Can be->Removing;

will beThe method is divided into a real part and an imaginary part for operation respectively, and the X polarization state is as follows:

，

the real part is:

，

the imaginary part is:

，

wherein,、/>initial signal of I, Q two signals respectively polarized by X,>、/>the sampling signals are I, Q signals with the X polarization state respectively;

dividing the X polarization state into two square operations, and performing fourth square operation to obtain the X polarization state as follows:

，

；

the calculation process of the Y polarization state is the same as that of the X polarization state;

s3, removing noise phases:

with noise phaseIn the case of- >Respectively calculating the arithmetic mean of the real part and the imaginary part of the X polarization state and the Y polarization state after the four times of calculation to obtain phase results which are all positioned at ideal phase valuesNearby, the obtained phase results are averaged again to obtain the ideal phase value +.>；

S4, calculating a carrier frequency offset estimation value:

the arithmetic mean of the real part and the imaginary part of the X polarization state and the Y polarization state obtained in S3 is、、/>、/>；

If the real part of the X polarization stateAnd imaginary part->Angle of irradiance；

If the real part of the X polarization stateThe argument is unchanged, ++>；

If the real part of the X polarization stateWhen the imaginary part is greater than 0, then the argument +>The method comprises the steps of carrying out a first treatment on the surface of the When the imaginary part is smaller than 0, then the argument +.>；

Calculating the same X polarization state by the amplitude angle of the Y polarization state;

wherein the method comprises the steps of，/>；

Obtaining the frequency offset value，/>；

Correcting the X polarization state and the Y polarization state by using the frequency deviation value respectively to obtain a real part and an imaginary part of the corrected X polarization state respectively as follows:

，

；

the real part and the imaginary part of the Y polarization state after correction are respectively:

，

；

and outputting the restored high-frequency carrier signal after the frequency offset correction.

In one embodiment of the invention, a CDMA spreading code is used to spread the original digital signal to generate a mask.

In one embodiment of the present invention, at the receiving end, each channel of the despread digital signal is quantized and sample mapped, and the digital signal is restored to the original binary data by the multilevel digital modulation decoding.

The invention also provides a digital modulation device, comprising:

a transmitting terminal, comprising:

the spread spectrum module is used for superposing digital signals subjected to spread spectrum processing of different users, converting the superposed digital signals into analog signals, carrying out radio frequency modulation, and modulating the analog signals into high-frequency carrier signals;

the input end of the IQ modulator is connected with the output end of the spread spectrum module and is used for converting the high-frequency carrier signal into an optical signal;

the single chip microcomputer is connected with the IQ modulator and is used for coarsely adjusting the bias voltage of the IQ modulator by utilizing a polarization control algorithm before the high-frequency carrier signal is input to the IQ modulator, and the bias voltage is adjusted within a preset bias range according to the minimum value and the maximum value of the optical power so as to lock the bias point near the linear point; after a high-frequency carrier signal is input into an IQ modulator, continuously tracking bias voltage by using a polarization control algorithm, obtaining direct current components of two different optical powers and alternating current components of two different optical powers by using bias voltage which is increased and reduced by a stepping value, respectively taking bias voltages corresponding to the smaller direct current components and the alternating current components as new bias voltages, and continuously iterating until the bias voltage is not changed;

The input end of the small signal-to-noise amplifier is connected with the IQ modulator, and the output end of the small signal-to-noise amplifier is connected with the optical fiber link and is used for amplifying the power of the optical signal;

a receiving terminal, comprising:

the input end of the backward Raman amplifier is connected with the optical fiber link and is used for amplifying the optical signal transmitted by the optical fiber;

the input end of the carrier recovery module is connected with the output end of the backward Raman amplifier, and is used for carrying out self-adaptive equalization processing by utilizing a coherent demodulation algorithm, carrying out frequency offset recovery and phase recovery on the output result of the coherent demodulation algorithm by adopting the carrier recovery algorithm, and restoring the output result into a high-frequency carrier signal;

and the despreading module is used for demodulating the restored high-frequency carrier signal into an analog signal, converting the analog signal into a digital signal and performing despreading.

Compared with the prior art, the technical scheme of the invention has the following advantages:

the invention relates to a digital modulation method, which is characterized in that a polarization control algorithm in a singlechip is utilized at a transmitting end to adjust bias voltage of an IQ modulator, the polarization control algorithm is divided into two stages of coarse adjustment and fine adjustment, signal data are not added in the coarse adjustment stage to quickly adjust, the bias point is approximately locked near a linear point, then data are added to finely adjust, and the stability of the working state of the IQ modulator is ensured by detecting direct current component and alternating current component of optical power, thereby ensuring the polarization stability of an optical signal, reducing the speed-up of noise of the optical signal in transmission, and realizing ultra-long distance transmission of the signal in an optical fiber.

The digital modulation method of the invention carries out self-adaptive equalization processing on the received signals by utilizing a coherent demodulation algorithm at a receiving end, effectively reduces the error rate in signal transmission, suppresses channel noise and interference by eliminating group delay distortion generated in the transmission process, can extract information in the original signals to the maximum extent, reduces errors and distortion introduced in the transmission process, and thus improves the reliability of data. The carrier recovery algorithm is adopted to carry out frequency offset recovery and phase recovery on the output result of the coherent demodulation algorithm, so that the carrier frequency and the phase of the transmitting end can be accurately recovered at the receiving end, the frequency and the phase between the receiving end and the transmitting end are synchronous, the demodulation and the correct data recovery of signals are realized, and the transmission distance of the signals in light is improved.

The digital modulation method of the invention can realize the decoding transmission capability aiming at special specific users by utilizing the orthogonality of multi-client communication of CDMA codes. The multi-frequency point signals are generated by using CDMA construction, and the spread spectrum mask in CDMA coding can ensure that a plurality of users can only extract and decode own transmission signals when simultaneously communicating and transmitting, thereby ensuring the safety and confidentiality of the whole transmission system for the simultaneous communication and transmission of a plurality of clients.

Drawings

In order that the invention may be more readily understood, a more particular description of the invention will be rendered by reference to specific embodiments thereof that are illustrated in the appended drawings, in which

FIG. 1 is a flow chart diagram of a digital modulation method of the present invention;

FIG. 2 is a schematic diagram of a polarization control algorithm in an embodiment of the present invention;

FIG. 3 is a block diagram of a coherent demodulation algorithm in an embodiment of the present invention;

FIG. 4 is a schematic block diagram of a fourth-order frequency offset estimation algorithm in an embodiment of the invention;

fig. 5 is a block diagram of a digital modulation apparatus according to the present invention.

Detailed Description

The present invention will be further described with reference to the accompanying drawings and specific examples, which are not intended to be limiting, so that those skilled in the art will better understand the invention and practice it.

Example 1

Referring to fig. 1, the present invention provides a digital modulation method, including:

at a transmitting end, binary data is input, multi-system digital modulation is carried out, the binary data is subjected to digital modulation coding, sampling mapping and quantization are carried out, and an initial digital signal is obtained; performing spread spectrum processing on the initial digital signal through a CDMA spread spectrum code to generate a mask; after the initial digital signal is transmitted through CPRI and processed through digital intermediate frequency, the digital signal is converted into an analog signal by using a digital-to-analog converter, radio frequency modulation is carried out, and the analog signal is modulated into a high-frequency carrier signal; the above process allows all users to transmit not only using wideband carrier signals of the same frequency, but the signals are also overlapping in time.

The embodiment of the invention can realize the decoding transmission capability aiming at special specific users by utilizing the multi-client communication orthogonality of the CDMA codes, generate multi-frequency point signals by utilizing the CDMA construction, and ensure that a plurality of users can only extract and decode own transmission signals when simultaneously transmitting the communication by utilizing the spread spectrum mask in the CDMA codes, thereby ensuring the safety and confidentiality of the whole transmission system for the simultaneous communication transmission of a plurality of clients.

Referring to fig. 2, before a high-frequency carrier signal is input to an IQ modulator, a polarization control algorithm in a single-chip microcomputer is utilized to coarsely adjust bias voltage of the IQ modulator, and bias voltage values are adjusted within a preset bias voltage range according to minimum and maximum values of optical power so as to lock bias points near linear points, and the specific steps include:

After the bias voltage is coarsely regulated, a high-frequency carrier signal is input into an IQ modulator, the bias voltage is continuously tracked by utilizing a polarization control algorithm in a singlechip, the bias voltage which is increased and reduced firstly by a stepping value is used for obtaining direct current components of two different optical powers and alternating current components of two different optical powers, the bias voltages corresponding to the smaller direct current components and the alternating current components are respectively taken as new bias voltages, and iteration is continuously carried out until the bias voltages are not changed, and the specific steps comprise:

The high-frequency carrier signal is converted into an optical signal through an IQ modulator, the power of the optical signal is amplified through a small signal-to-noise amplifier, and the optical signal enters a 500km optical fiber link after completion and is transmitted in an ultra-long distance.

When the polarization control algorithm is not added, the signal can cause error code surge because of signal noise increase after ultra-long distance transmission, but the signal noise increase speed of the signal is relaxed after the polarization control algorithm is added, so that the ultra-long distance transmission of the signal in the optical fiber can be realized.

At the receiving end, the optical signal transmitted by the optical fiber is amplified by the backward Raman amplifier, self-adaptive equalization processing is carried out by using a coherent demodulation algorithm, and frequency offset recovery and phase recovery are carried out on the output result of the coherent demodulation algorithm by using a carrier recovery algorithm, so that the high-frequency carrier signal is recovered. The following describes the signal processing procedure of the receiving end in detail.

Referring to FIG. 3, the original signal transmitted by the signal source isAfter 500km optical fiber link, the channel additive Gaussian white noise is combined>The equalizer receiving sequence of the receiving end is +.>. Equalizer is from receiving the sequence->Information is acquired in (1), and the adaptive adjustment length is +.>Equalizer weight coefficients +.>So that the output after equalizer convergence +.>For the original signal->Is a good estimate of the best estimate of (a).

The model construction process for eliminating the signal group delay by using the filter is as follows:

when a group signal passes through a communication system, the time delay that the communication system produces for the group signal wave group as a whole is called group time delay, wherein the group signal refers to a complex signal or wave group that has a frequency close to a plurality of frequency components and is composed in a certain way, namely. Group delay distortion describes the effect of the non-uniformity of the channel time over different frequency portions of a group signal.

The signal group delay concept and its characteristics are as follows:

let the frequency response characteristic function of the communication channel be:

，

wherein,and->The characteristic functions of group delay are respectively the channel amplitude frequency and phase frequencyThe definition is as follows:

，

the group delay frequency response characteristic function is equal to the system phase frequency characteristic functionDiagonal frequency->Is a derivative of the number of (a). Wherein a negative sign indicates that the system output signal is always lagging with respect to its input signal. For example- >Is a constant, i.e.)>And->The signal is linear, and the different frequency parts of the signal have the same group delay, so the signal cannot be distorted when passing through the channel. On the contrary, if->If not constant, the different frequency parts of the signal will generate group delay distortion through the channel.

Set group delay channel impulse response sequence asWherein->For the channel filter length, then the discrete form of the channel frequency response function is:

；

from the group delay characteristic functionThe definition of (1) yields the group delay frequency response function as:

，

wherein,for the phase frequency characteristic function of the system->Representing the real part arithmetic,/->Representing the imaginary part.

Desired amplitude-frequency response for a given channelAnd group delay frequency response->Establishing a weighted square error cost function of amplitude frequency and group delay frequency response, and participating in equalization calculation and coefficient update in an equalizer composite mode, and eliminating signal group delay by using an FIR filter, wherein the weighted square error cost function of a group delay model is as follows:

，

wherein, the embodiment of the invention uses an FIR filter as an equalizer,for the passband of the FIR filter, < >>Is the stop band of the FIR filter, < >>Amplitude weights in the cost error function; />For the desired amplitude-frequency response of the channel, +. >For the desired group delay frequency response of the channel, +.>Is the angular frequency.

Discretizing the frequency, the weighted square error cost function becomes:

。

setting group delay channel impulse response sequenceThe weighted square error cost function is minimized, and the expression of the obtained group delay channel FIR filter model is as follows:

。

from the group delay channel FIR filter model, a set of systems can be obtainedI.e. a digital group delay channel impulse response sequence, so that the weighted square error of the group delay model is minimum, and the group delay filter used in the embodiment of the invention is obtained.

Using FIR filters for sequencesAnd carrying out equalization processing for a plurality of times to eliminate the signal group delay, then using a decision device to make decision and data decoding on the signal after eliminating the group delay, and outputting the original transmitting signal without carrier recovery.

The method has the advantages that the self-adaptive equalization processing is carried out on the received signals by utilizing the coherent demodulation algorithm at the receiving end, the error rate in the signal transmission is effectively reduced, the channel noise and interference are restrained by eliminating the group delay distortion generated in the transmission process, the information in the original signals can be extracted to the maximum extent, the errors and distortion introduced in the transmission process are reduced, and therefore the reliability of the data is improved.

Because the optical signals are transmitted in the channel with too long a distance, phase offset occurs, and therefore, a carrier recovery algorithm needs to be introduced.

And carrying out frequency offset estimation and phase recovery on the signals subjected to decision and data decoding by carrier recovery to obtain a constellation diagram of a corresponding modulation format. The method corresponds to frequency offset recovery and phase recovery in a digital signal processing flow, wherein a frequency offset recovery algorithm corrects residual frequency errors by using an M-th-order frequency offset estimation algorithm, and a phase recovery algorithm corrects residual phase errors by using a Viterbi-Viterbi (V-V) phase estimation algorithm.

Referring to fig. 4, in an embodiment of the present invention, a fourth-order frequency offset estimation algorithm is used to correct the residual frequency error, and the specific steps include:

s1, removing phase damage introduced by laser linewidth:

setting the phase of the received signal to beWherein->Representing information phase +.>Representing the phase caused by the linewidth of the laser, +.>Represents noise phase +.>Serial number ± representing sample value>Representing a period; />

，

wherein the method comprises the steps ofIs a slowly varying signal, so that the front and back sample values are zero, and the laser linewidth can be eliminated after the step >Is a phase impairment caused by (a) a phase shift;

s2, removing residual modulation phases:

，

the real part is:

，

the imaginary part is:

，

；

the calculation process of the Y polarization state is the same as that of the X polarization state:

will beThe method is divided into a real part and an imaginary part for operation respectively, and the Y polarization state is as follows:

，

the real part is:

，

the imaginary part is:

，

wherein,、/>initial signal of I, Q two signals with Y polarization respectively,>、/>the sampling signals are I, Q signals with Y polarization states respectively;

dividing the Y polarization state into two square operations, and performing fourth square operation to obtain the Y polarization state as follows:

，/>

；

s3, removing noise phases:

in the ideal case, i.e. without noise phaseIn the case of taking->The signal samples are calculated by calculating the arithmetic mean of the real part and the imaginary part of the X polarization state and the Y polarization state after the four times operation, and the phase of each result is I.e. the value of each result is the same; but there is actually noise phase +>In the case of->The phase result obtained for each signal sample is at the ideal phase value +.>Nearby, so that averaging the phase results yields the ideal phase value；

S4, calculating a carrier frequency offset estimation value:

If the real part of the X polarization stateThe argument is unchanged, ++>；

The amplitude angle of the Y polarization state is calculated to be identical to the X polarization state:

if the real part of the Y polarization stateAnd imaginary part->Angle of irradiance；

If the real part of the Y polarization stateThe argument is unchanged, ++>；

If the real part of the Y polarization stateWhen the imaginary part is greater than 0, then the argument +>The method comprises the steps of carrying out a first treatment on the surface of the When the imaginary part is smaller than 0, then the argument +.>；

Wherein the method comprises the steps of，/>；

Obtaining the frequency offset value，/>；

，

；

，

；

The carrier recovery algorithm is adopted to carry out frequency offset recovery and phase recovery on the output result of the coherent demodulation algorithm, so that the carrier frequency and the phase of the transmitting end can be accurately recovered at the receiving end, the frequency and the phase between the receiving end and the transmitting end are synchronous, the demodulation and the correct data recovery of signals are realized, and the transmission distance of the signals in light is improved.

The restored high-frequency carrier signal is subjected to broadband filtering, the in-band signal amplitude is adjusted and limited, the in-band signal is demodulated into an analog signal, the analog signal is converted into a digital signal through a digital-to-analog converter, then the digital signal is despread, a mask is filtered out after the digital intermediate frequency processing and CPRI transmission are carried out, and the despread signal is subjected to baseband filtering to recover the digital baseband signal; and carrying out quantization processing and sampling mapping on each channel, and recovering the despread digital signal into initial binary data through multi-system digital modulation decoding.

After the digital modulation method is adopted, the optical signal can be transmitted for a longer distance on the original basis.

Example two

Referring to fig. 5, an embodiment of the present invention provides a digital modulation apparatus, including:

A transmitting terminal, comprising:

a receiving terminal, comprising:

It will be appreciated by those skilled in the art that embodiments of the present application may be provided as a method, system, or computer program product. Accordingly, the present application may take the form of an entirely hardware embodiment, an entirely software embodiment, or an embodiment combining software and hardware aspects. Furthermore, the present application may take the form of a computer program product embodied on one or more computer-usable storage media (including, but not limited to, disk storage, CD-ROM, optical storage, and the like) having computer-usable program code embodied therein.

The present application is described with reference to flowchart illustrations and/or block diagrams of methods, apparatus (systems) and computer program products according to embodiments of the application. It will be understood that each flow and/or block of the flowchart illustrations and/or block diagrams, and combinations of flows and/or blocks in the flowchart illustrations and/or block diagrams, can be implemented by computer program instructions. These computer program instructions may be provided to a processor of a general purpose computer, special purpose computer, embedded processor, or other programmable data processing apparatus to produce a machine, such that the instructions, which execute via the processor of the computer or other programmable data processing apparatus, create means for implementing the functions specified in the flowchart flow or flows and/or block diagram block or blocks.

These computer program instructions may also be stored in a computer-readable memory that can direct a computer or other programmable data processing apparatus to function in a particular manner, such that the instructions stored in the computer-readable memory produce an article of manufacture including instruction means which implement the function specified in the flowchart flow or flows and/or block diagram block or blocks.

These computer program instructions may also be loaded onto a computer or other programmable data processing apparatus to cause a series of operational steps to be performed on the computer or other programmable apparatus to produce a computer implemented process such that the instructions which execute on the computer or other programmable apparatus provide steps for implementing the functions specified in the flowchart flow or flows and/or block diagram block or blocks.

It is apparent that the above examples are given by way of illustration only and are not limiting of the embodiments. Other variations and modifications of the present invention will be apparent to those of ordinary skill in the art in light of the foregoing description. It is not necessary here nor is it exhaustive of all embodiments. And obvious variations or modifications thereof are contemplated as falling within the scope of the present invention.

Claims

1. A digital modulation method, comprising:

the high-frequency carrier signal is converted into an optical signal through an IQ modulator, the power of the optical signal is amplified through a small signal-to-noise amplifier, and the optical signal enters an optical fiber for transmission after the completion; at a receiving end, amplifying an optical signal transmitted by an optical fiber through a backward Raman amplifier, performing self-adaptive equalization processing by using a coherent demodulation algorithm, and performing frequency offset recovery and phase recovery on an output result of the coherent demodulation algorithm by using a carrier recovery algorithm; wherein the method comprises the steps of

And compensating the frequency offset of the signal by using an M-th frequency offset estimation algorithm, wherein the M-th frequency offset estimation algorithm comprises a fourth-order frequency offset estimation algorithm, and the specific steps comprise:

s1, removing phase damage introduced by laser linewidth:

setting the phase of the received signal to be theta _k ＝θ _s (k)+ΔωkT _i +θ _n +θ _ASE Wherein θ is _s (k) Representing the phase of the information, θ _n Represents the phase, θ, caused by the linewidth of the laser _ASE Representing noise phase, T representing period;

the conjugate of the sample value signal S (k-1) and the initial signal S (k) are multiplied to obtain:

wherein θ is _n Is a slowly varying signal, so that the front and back sample values are zero, and the linewidth theta of the laser can be eliminated after the step _n Is a phase impairment caused by (a) a phase shift;

s2, removing residual modulation phases:

in an ideal case, the residual modulation phase Δθ _s The value of (2) isThen [ S ] ^* (k-1)S(k)] ⁴ In (a) and (b)Can be delta theta _s Removing;

will S ^* (k-1) S (k) is divided into a real part and an imaginary part to be respectively operated, and the X polarization state is as follows:

S _k,x S ^* _k-1,x ＝(X _I,k +jX _Q,k )·(X _I,k-1 -jX _Q,k-1 )

the real part is:

S _{f_x_r} ＝X _I,k ·X _I,k-1 +X _Q,k ·X _Q,k-1

the imaginary part is:

S _{f_x_i} ＝X _Q,k ·X _I,k-1 -X _I,k ·X _Q,k-1

wherein X is _I,k 、X _Q,k Initial signals of I, Q two paths of signals respectively polarized by X _I,k-1 、X _Q,k-1 The sampling signals are I, Q signals with the X polarization state respectively;

S _{f_2_x} ＝S _{f_2_x_r} +jS _{f_2_x_i} ＝(S _{f_x_r} +jS _{f_x_i} ) ²

＝S ² _{f_x_r} -S ² _{f_x_i} +2jS _{f_x_r} S _{f_x_i}

S _{f_4_x} ＝S _{f_4_x_r} +jS _{f_4_x_i} ＝(S _{f_2_x_r} +jS _{f_2_x_i} ) ²

＝S ² _{f_2_x_r} -S ² _{f_2_x_i} +2jS _{f_2_x_r} S _{f_2_x_i}

S3, removing noise phases:

with noise phase theta _ASE In the case of (a), taking Nf signal samples, calculating the arithmetic mean of the real and imaginary parts of the X-polarization and Y-polarization after the above four-time operation, respectively, and obtaining a phase junctionThe result is located near the ideal phase value delta omega T, and the obtained phase result is averaged to obtain the ideal phase value delta omega T;

s4, calculating a carrier frequency offset estimation value:

the arithmetic mean of the real part and the imaginary part of the X polarization state and the Y polarization state obtained in S3 is sigma _Nf S _{f_4_x_r} 、∑ _Nf S _{f_4_x_i} 、∑ _Nf S _{f_4_y_r} 、∑ _Nf S _{f_4_y_i} ；

If the real part of the X polarization state is _Nf S _{f_4_x_r} <0 and imaginary part Σ _Nf S _{f_4_x_i} <0, then the argument ΔωT _x.4 ＝ΔωT _x,1,4 -π；

If the real part of the X polarization state is _Nf S _{f_4_x_r} <0 and imaginary part Σ _Nf S _{f_4_x_i} >0, then the argument ΔωT _x.4 ＝ΔωT _x,1,4 +π；

If the real part of the X polarization state is _Nf S _{f_4_x_r} >0, the argument is unchanged, ΔωT _x.4 ＝ΔωT _x,1,4 ；

If the real part of the X polarization state is _Nf S _{f_4_x_r} When the imaginary part is greater than 0, the amplitude angleWhen the imaginary part is smaller than 0, then the argument +.>

wherein the method comprises the steps ofObtaining the frequency offset value

Correcting the X polarization state and the Y polarization state respectively by using the frequency offset value to obtain the corrected X polarization state with the real part and the imaginary part respectively as follows:

S _{fout_k_x_r} ＝X _I,k ·cos(Δω _x kT)+X _Q,k ·sin(Δω _x kT)

S _{fout_k_x_i} ＝X _Q,k ·cos(Δω _x kT)-X _I,k ·sin(Δω _x kT)

S _{fout_k_y_r} ＝Y _I,k ·cos(Δω _y kT)+Y _Q,k ·sin(Δω _y kT)

S _{fout_k_y_i} ＝Y _Q,k ·cos(Δω _y kT)-Y _I,k ·sin(Δω _y kT)

outputting a restored high-frequency carrier signal after frequency offset correction;

2. The digital modulation method as claimed in claim 1, wherein the bias voltage of the IQ modulator is coarsely adjusted by using a polarization control algorithm in a single chip microcomputer connected to the IQ modulator, and the bias voltage is adjusted within a preset bias range according to a minimum value and a maximum value of the optical power so as to lock the bias point near the linear point, comprising the specific steps of:

3. The digital modulation method as set forth in claim 2, wherein the high frequency carrier signal is input to the IQ modulator, the polarization control algorithm in the single chip continuously tracks the bias voltage, the bias voltage increased and decreased by a step value to obtain dc components of two different optical powers, and ac components of two different optical powers, respectively taking the bias voltages corresponding to the smaller dc components and ac components as new bias voltages, and iterating until the bias voltages are no longer changed, the specific steps include:

4. The digital modulation method according to claim 1, wherein said performing adaptive equalization processing using a coherent demodulation algorithm comprises: and eliminating the signal group delay by using an FIR filter, and then deciding and data decoding the signal with the eliminated group delay by using a decision device to output the original transmitting signal without carrier recovery.

5. The digital modulation method of claim 4, wherein the group delay is eliminated by using an FIR filter, and the weighted square error cost function of the group delay model is:

e＝α∫ _ω∈P,S ||H(e ^jω )|-|H _d (e ^jω )|| ² dω+∫ _ω∈P |τ(ω)-τ _d (ω)| ² dω

wherein P is the passband of the FIR filter, S is the stopband of the FIR filter, and alpha is the amplitude weight in the cost error function; h _d (e ^jω ) For the desired amplitude-frequency response of the channel, τ _d (ω) is the expected group delay frequency response of the channel, ω is the angular frequency;

H(e ^jω ) As a channel frequency response function:

wherein h is a group delay channel impulse response sequence: h= [ h (0), h (1), …, h (N-1)] ^T N is the length of the channel filter;

τ (ω) is a group delay frequency response function:

wherein,re [. Cndot.]Representing a real part calculation;

discretizing the frequency, the weighted square error cost function becomes:

the group delay channel impulse response sequence h is set such that the weighted square error cost function is minimized,

the expression for obtaining the group delay channel FIR filter model is as follows:

6. the digital modulation method as claimed in claim 1, wherein the performing frequency offset recovery and phase recovery on the output result of the coherent demodulation algorithm by using a carrier recovery algorithm comprises:

the phase of the signal is compensated using a Viterbi-Viterbi phase estimation algorithm.

7. A digital modulation method according to claim 1 wherein the initial digital signal is spread using a CDMA spreading code to generate the mask.

8. A digital modulation method according to claim 1, wherein at the receiving end, each channel of the despread digital signal is quantized and sample mapped, and the digital signal is restored to the original binary data by the multilevel digital modulation decoding.

9. A digital modulation apparatus, comprising:

a transmitting terminal, comprising:

the single chip microcomputer is connected with the IQ modulator and is used for coarsely adjusting the bias voltage of the IQ modulator by utilizing a polarization control algorithm before the high-frequency carrier signal is input to the IQ modulator, and the bias voltage is adjusted within a preset bias range according to the minimum value and the maximum value of the optical power so as to lock the bias point near the linear point; after the high-frequency carrier signal is input into the IQ modulator, the bias voltage is continuously tracked by utilizing a polarization control algorithm, the bias voltage which is increased and then reduced by a stepping value is used for obtaining direct current components of two different optical powers,

and alternating current components of two different optical powers, namely respectively taking the smaller direct current component and the offset voltage corresponding to the alternating current component as new offset voltage, and continuously iterating until the offset voltage is not changed;

a receiving terminal, comprising:

the input end of the carrier recovery module is connected with the output end of the backward Raman amplifier, and is used for carrying out self-adaptive equalization processing by utilizing a coherent demodulation algorithm, and carrying out frequency offset recovery and phase recovery on the output result of the coherent demodulation algorithm by adopting the carrier recovery algorithm; the method comprises the following specific steps of:

s1, removing phase damage introduced by laser linewidth:

s2, removing residual modulation phases:

S _k,x S ^* _k-1,x ＝(X _I,k +jX _Q,k )·(X _I,k-1 -jX _Q,k-1 )

the real part is:

S _{f_x_r} ＝X _I,k ·X _I,k-1 +X _Q,k ·X _Q,k-1

the imaginary part is:

S _{f_x_i} ＝X _Q,k ·X _I,k-1 -X _I,k ·X _Q,k-1

S _{f_2_x} ＝S _{f_2_x_r} +jS _{f_2_x_i} ＝(S _{f_x_r} +jS _{f_x_i} ) ²

＝S ² _{f_x_r} -S ² _{f_x_i} +2jS _{f_x_r} S _{f_x_i}

S _{f_4_x} ＝S _{f_4_x_r} +jS _{f_4_x_i} ＝(S _{f_2_x_r} +jS _{f_2_x_i} ) ²

＝S ² _{f_2_x_r} -S ² _{f_2_x_i} +2jS _{f_2_x_r} S _{f_2_x_i}

s3, removing noise phases:

with noise phase theta _ASE Taking Nf signal samples, respectively calculating the arithmetic averages of the real part and the imaginary part of the X polarization state and the Y polarization state after the fourth-time operation, obtaining phase results which are all positioned near an ideal phase value delta omega T, and averaging the obtained phase results to obtain the ideal phase value delta omega T;

s4, calculating a carrier frequency offset estimation value:

wherein the method comprises the steps of

Obtaining the frequency offset value

S _{fout_k_x_r} ＝X _I,k ·cos(Δω _x kT)+X _Q,k ·sin(Δω _x kT)

S _{fout_k_x_i} ＝X _Q,k ·cos(Δω _x kT)-X _I,k ·sin(Δω _x kT)

S _{fout_k_y_r} ＝Y _I,k ·cos(Δω _y kT)+Y _Q,k ·sin(Δω _y kT)

S _{fout_k_y_i} ＝Y _Q,k ·cos(Δω _y kT)-Y _I,k ·sin(Δω _y kT)

and the input end of the despreading module is connected with the output end of the carrier recovery module and is used for demodulating the restored high-frequency carrier signal into an analog signal, converting the analog signal into a digital signal and performing despreading.