CN116885860A - Control method of underwater wireless charging system - Google Patents
Control method of underwater wireless charging system Download PDFInfo
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Classifications
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
- H02J50/10—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling
- H02J50/12—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling of the resonant type
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J7/00—Circuit arrangements for charging or depolarising batteries or for supplying loads from batteries
- H02J7/02—Circuit arrangements for charging or depolarising batteries or for supplying loads from batteries for charging batteries from ac mains by converters
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
- H02M3/33573—Full-bridge at primary side of an isolation transformer
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
- Y02T10/00—Road transport of goods or passengers
- Y02T10/60—Other road transportation technologies with climate change mitigation effect
- Y02T10/70—Energy storage systems for electromobility, e.g. batteries
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
- Y02T10/00—Road transport of goods or passengers
- Y02T10/60—Other road transportation technologies with climate change mitigation effect
- Y02T10/7072—Electromobility specific charging systems or methods for batteries, ultracapacitors, supercapacitors or double-layer capacitors
Abstract
The invention discloses a control method of an underwater wireless charging system, and relates to the field of underwater wireless charging. The invention solves the technical problem of improving the anti-offset characteristic and the power transmission capability of a wireless charging system on the premise of ensuring the power transmission power. The invention acquires a plurality of system resonance frequencies; selecting a coupling coefficient switching point, dividing a coupling coefficient range into a plurality of coefficient intervals by the coupling coefficient switching point, and respectively matching the system resonance frequency with each coefficient interval; the coupling coefficient is defined byiThe coefficient interval is switched to the firsti+1 or the firsti-1 coefficient interval, the firstiThe system resonant frequency is switched to the firsti+1 or the firsti-1 system resonant frequency, the switching time is defined bynThe time of the cycle is constituted by the time of the cycle,nthe cycle time varies in a linear increase. The invention is thatThe power, efficiency, anti-offset capability and power transmission capability of wireless power transmission are improved.
Description
Technical Field
The invention relates to the field of underwater wireless charging, in particular to a control method of an underwater wireless charging system.
Background
In recent years, wireless charging is increasingly widely used due to the advantages of safety, convenience and the like. The underwater wireless power transmission can be widely applied to electric energy supplementation of underwater unmanned aerial vehicles and robots, and can also provide reliable energy supply for ocean observation and monitoring systems. The magnetic coupling resonance type wireless charging mode utilizes the electromagnetic induction principle to transmit electric energy from the power grid end to the mobile end. Compared with the traditional contact type charging, the charging mode is safer and more reliable, has high flexibility and is more beneficial to maintenance. Because a certain gap exists between the two coils in the magnetic induction coupling type wireless charging, the main magnetic circuit has larger magnetic resistance, and the system needs extremely large reactive power, so that the capacity of equipment can be increased, the loss of the system is increased, and the efficiency of the system is reduced. To avoid this drawback, the primary and secondary ends of the magnetic coupling resonant wireless charging system are usually added with compensation networks, which improve the power factor and reduce the current carrying capacity of the switching device. The compensation network generally needs to be adopted on both the primary side and the secondary side, and has a low-order compensation and a high-order compensation. Because of the obvious short plates of the low-order compensation network, the research on the high-order compensation network is increasingly applied at present.
The bilateral LCC compensation network has a plurality of resonance frequencies, has the characteristics of a constant voltage source and a constant current source and has no-load protection characteristics compared with a low-order compensation network, and is increasingly widely applied due to the outstanding advantages of the bilateral LCC compensation network. In the prior art, the compensation network is improved and controlled, but when the underwater wireless charging system works, the coil can shift the relative position between the transmitting coil and the receiving coil based on factors such as water flow, so that the coupling coefficient is changed, and the electric energy transmission efficiency is affected.
Therefore, how to improve the anti-offset characteristic of the underwater wireless charging system under the premise of ensuring the transmission power is a problem to be solved.
Disclosure of Invention
In view of the above-mentioned drawbacks of the prior art, the present invention aims to improve the anti-offset performance and the power transmission capability of a wireless charging system on the premise of ensuring the power transmission.
The first aspect of the present invention provides a control method of an underwater wireless charging system, the underwater wireless charging system comprising a bilateral LCC compensation network, a primary side loop of the bilateral LCC compensation network being connected to a transmitting coil L P2 The secondary side loop is connected with the receiving coil L S2 The primary side loop comprises a primary side compensation capacitor C P1 、C P2 And primary side compensation inductance L P1 The secondary side comprises a secondary compensation capacitor C S1 、C S2 And secondary side compensation inductance L S1 ；
The control method comprises the following steps:
obtaining a plurality of system resonance frequencies { omega } 0 ，ω 1 ，ω 2 ，…，ω m }, wherein omega 0 Is the natural resonant frequency omega 1 ，ω 2 ，…，ω m Is the metastable resonant frequency;
selecting a coupling coefficient switching point { k c1 ，k c2 ，…，k cm Dividing the coupling coefficient range into a plurality of coefficient intervals by the coupling coefficient switching point, wherein the system resonance frequency is matched with each coefficient interval;
when the coupling coefficient is switched from the ith coefficient interval to the (i+1) th or the (i-1) th coefficient interval, the ith system resonant frequency is switched to the (i+1) th or the (i-1) th system resonant frequency, and the time length t required by the (i) th system resonant frequency to the (i+1) th or the (i-1) th system resonant frequency v The duration t v Consists of n cycle times which vary in a linear increase.
Further, the time period t required for the ith system resonant frequency to switch to the (i+1) th or (i-1) th system resonant frequency v The method comprises the following steps:
t v =T 1 +T 2 +…+T n ；
wherein T is n For the nth cycle time, it is: t (T) n =T 0 +n•ΔT。
Further, the method for obtaining the resonant frequency includes:
Calculating the input impedance of the bilateral LCC compensation network, and acquiring a resonance frequency point when the imaginary part of the input impedance is zero;
the resonance frequency point which is not affected by the load is taken as the natural resonance frequency, and the resonance frequency point which meets the condition that the imaginary part of the input impedance of the system is far smaller than the input impedance is taken as the metastable resonance frequency.
Further, when the bilateral LCC compensation network works in the constant voltage source mode, the primary side compensation capacitor C P1 、C P2 Primary side compensation inductance L P1 And a transmitting coil L P2 The constraint conditions are as follows:
L P2 =(1+λ p )L P1 ；
wherein lambda is p = C P1 / C P2 。
Further, the secondary compensation capacitor C S1 、C S2 Secondary side compensation inductance L S1 And a receiving coil L S2 The constraint conditions are as follows:
L S2 =(1+λ S )L S1 ；
wherein lambda is S = C S1 / C S2 。
Further, when the underwater wireless charging system is operated in the current source mode, lambda p And lambda (lambda) S The constraint relation between the two is as follows:
；
lambda when the underwater wireless charging system is operating in a voltage source mode p And lambda (lambda) S The constraint relation between the two is as follows:
。
further, the method for determining the coupling coefficient switching point comprises the following steps:
calculating inverter output power at natural resonant frequencyAnd inverter output power at m metastable resonant frequencies +.>；
Inverter output power at natural resonant frequencyInverter output power +_with the ith sub-stable resonant frequency >The coupling coefficient at the same time is the ith coupling coefficient switching point k ci 。
The second aspect of the present invention provides a control method of an underwater wireless charging system, the underwater wireless charging system including a bilateral LCC compensation network, a primary side of the bilateral LCC compensation network including a primary compensation capacitor C P1 、C P2 Primary side compensation inductance L P1 And a transmitting coil L P2 The secondary side comprises a secondary compensation capacitor C S1 、C S2 Secondary side compensation inductance L S1 And a receiving coil L S2 The control method is characterized by comprising the following steps:
obtaining a plurality of system resonance frequencies { omega } 0 ，ω 1 ，ω 2 ，…，ω m }；
Selecting a coupling coefficient switching point { k c1 ，k c2 ，…，k cm Dividing the coupling coefficient range into a plurality of coefficient intervals by the coupling coefficient switching point, wherein the system resonance frequency is matched with each coefficient interval;
when the coupling coefficient is switched from the ith coefficient interval to the (i+1) th or the (i-1) th coefficient interval, the time length required for switching the ith system resonance frequency to the (i+1) th or the (i-1) th system resonance frequency is approximately 0.
Compared with the prior art, the invention has the following technical effects:
1. according to the invention, a plurality of system resonance frequencies are found according to the actual condition of an actual wireless power transmission system, the power transmission capacity of the power transmission system is increased along with the increase of a coupling coefficient at a certain resonance frequency point, and the other resonance point is more suitable for power transmission at a low coupling coefficient, so that the power transmission system is matched with proper system resonance frequencies at different coupling coefficients, and proper resonance frequencies are selected according to the coupling coefficient change caused by the relative deflection of the coupling coils, thereby effectively improving the power and efficiency of the wireless power transmission of the system, realizing smaller current distortion rate and enabling the system to have better anti-deflection capacity and power transmission capacity.
2. The system resonant frequency switching process is realized by adopting periodic time linear smooth transformation, and the system can be kept to run stably in the resonant frequency switching process, and the current waveform output by the electric energy transmission system is a sine wave at the moment, so that the current amplitude change is stable.
3. The system resonance frequency switching process in the invention needs a length t v Is composed of n period times, time length t v Can be prolonged or shortened as required, when the time period t v When the time is prolonged, the switching process is more stable, and when the time period t is shortened v In this case, the system response speed can be increased. When t v When approaching to 0, namely a jump type switching mode, the switching time is shortest, and dynamic oscillation is easy to be caused under the condition, so that the system is unstable to operate.
The conception, specific structure, and technical effects of the present invention will be further described with reference to the accompanying drawings to fully understand the objects, features, and effects of the present invention.
Drawings
FIG. 1 is a schematic diagram of an underwater wireless charging system in accordance with an embodiment of the present invention;
FIG. 2 is a schematic circuit diagram of a dual-sided LCC compensation network in accordance with one embodiment of the invention;
FIG. 3 shows the different loads R according to an embodiment of the invention L A relationship between lower output current and operating frequency;
Fig. 4 shows waveforms of the output voltage and current of the inverter at different resonant frequencies when the coupling coefficient is 0.5, fig. 4a shows waveforms of the output voltage and current of the inverter at the natural resonant frequency (85K), fig. 4b shows waveforms of the output voltage and current of the inverter at the metastable resonant frequency (98.15K), and fig. 4c shows waveforms of the output voltage and current of the inverter at the metastable resonant frequency (69.4K), according to an embodiment of the present invention;
FIG. 5 is a graph showing the current harmonic analysis results at different resonant frequencies, wherein the coupling coefficient is 0.5, the graph in FIG. 5a is the current harmonic analysis results of the natural resonant frequency (85K) inverter, the graph in FIG. 5b is the current harmonic analysis results of the sub-stable resonant frequency (98.15K) inverter, and the graph in FIG. 5c is the current harmonic analysis results of the sub-stable resonant frequency (69.4K) inverter according to an embodiment of the present invention;
fig. 6 shows waveforms of the inverter output voltage and current at different resonant frequencies, fig. 6a shows waveforms of the inverter output voltage and current at a natural resonant frequency (85K), fig. 6b shows waveforms of the inverter output voltage and current at a metastable resonant frequency (98.15K), and fig. 6c shows waveforms of the inverter output voltage and current at a metastable resonant frequency (69.4K), according to an embodiment of the present invention;
FIG. 7 shows the current harmonic analysis results at different resonant frequencies, with the coupling coefficient being 0.3, FIG. 7a shows the current harmonic analysis results of the natural resonant frequency (85K) inverter, FIG. 7b shows the current harmonic analysis results of the metastable resonant frequency (98.15K) inverter, and FIG. 7c shows the current harmonic analysis results of the metastable resonant frequency (69.4K) inverter, according to an embodiment of the present invention;
FIG. 8 is a transmission characteristic of a constant-current type underwater wireless charging system according to an embodiment of the present invention, FIG. 8a is a transmission efficiency characteristic of the constant-current type underwater wireless charging system, and FIG. 8b is a transmission power characteristic of the constant-current type underwater wireless charging system;
FIG. 9 is a graph showing the relationship between the gain of the input/output voltage and the operating frequency of the system under different load conditions according to an embodiment of the present invention;
FIG. 10 shows waveforms of output voltage and current of the inverter at different resonant frequencies when the coupling coefficient is 0.2, FIG. 10a shows waveforms of output current and voltage of the inverter at a resonant frequency of 73.5KHz, and FIG. 10b shows waveforms of output current and voltage of the inverter at a resonant frequency of 90KHz according to an embodiment of the present invention;
FIG. 11 is a graph showing the output current harmonic of the inverter at a coupling coefficient of 0.2, FIG. 11a is a graph showing the output current harmonic of the inverter at a resonant frequency of 73.5KHz, and FIG. 11b is a graph showing the output current harmonic of the inverter at a resonant frequency of 90KHz, according to an embodiment of the present invention;
Fig. 12 is a waveform of output voltage and current of the inverter at different resonant frequencies when the coupling coefficient is 0.6, fig. 12a is a waveform of output current and voltage of the inverter at a resonant frequency of 73.5KHz, and fig. 12b is a waveform of output current and voltage of the inverter at a resonant frequency of 90KHz according to an embodiment of the present invention;
FIG. 13 is a graph showing the output current harmonic of the inverter at a coupling coefficient of 0.6, FIG. 13a is a graph showing the output current harmonic of the inverter at a resonant frequency of 73.5KHz, and FIG. 13b is a graph showing the output current harmonic of the inverter at a resonant frequency of 90KHz, according to an embodiment of the present invention;
fig. 14 is a diagram showing simulation results of transmission power and transmission efficiency according to an embodiment of the present invention and a comparative example, fig. 14a is a diagram showing transmission power simulation results, and fig. 14b is a diagram showing transmission efficiency simulation results;
FIG. 15 is a waveform of the output voltage of the inverter when the system resonant frequency is switched from 90KHz to 73.5KHz by a linear smooth switching method according to an embodiment of the present invention;
FIG. 16 is a diagram showing waveforms of a driving signal and an inverter input current when a system resonant frequency is switched from 90KHz to 73.5KHz by a linear smooth switching method according to an embodiment of the present invention, wherein FIG. 16a is a waveform of the driving signal, and FIG. 16b is a waveform of the inverter input current;
FIG. 17 is a voltage waveform input by a receiving side rectifier of the underwater wireless charging system when a linear smooth switching of the system resonant frequency occurs at 0.2 seconds in an embodiment of the present invention;
FIG. 18 is a graph showing the output voltage waveform and current waveform of the inverter when the system resonant frequency is switched from 90KHz to 73.5 KHz;
FIG. 19 is a graph showing the input voltage waveform of the rectifier when the system resonant frequency is switched from 90KHz to 73.5KHz in a step-change mode according to an embodiment of the present invention;
fig. 20 is a voltage waveform input by a receiving-end rectifier of the underwater wireless charging system when the system resonant frequency is switched in a linear transformation switching mode when the system resonant frequency is switched in a hopping frequency of 0.2 seconds in an embodiment of the invention.
Detailed Description
Other advantages and effects of the present invention will become apparent to those skilled in the art from the following disclosure, which describes the embodiments of the present invention with reference to specific examples. The invention may be practiced or carried out in other embodiments that depart from the specific details, and the details of the present description may be modified or varied from the spirit and scope of the present invention. It should be noted that the following embodiments and features in the embodiments may be combined with each other without conflict.
It should be noted that the illustrations provided in the following embodiments merely illustrate the basic concept of the present invention by way of illustration, and only the components related to the present invention are shown in the illustrations, not according to the number, shape and size of the components in actual implementation, and the form, number and proportion of each component in actual implementation may be arbitrarily changed, and the layout of the components may be more complex.
Some exemplary embodiments of the invention have been described for illustrative purposes, it being understood that the invention may be practiced otherwise than as specifically shown in the accompanying drawings.
The underwater wireless power transmission can be widely applied to electric energy supplementation of underwater unmanned aerial vehicles and robots, and can also provide reliable energy supply for ocean observation and monitoring systems. As shown in fig. 1, the magnetic coupling resonance type underwater wireless charging system adopted in the embodiment comprises a high-frequency inverter circuit, a coupling mechanism, a compensation network and a high-frequency rectifying circuit. The transmitting end is provided with direct current input by an underwater cable or a mother ship, and the input direct current is converted into high-frequency alternating current by a high-frequency inverter circuit to provide energy for the primary side of the coupling mechanism. Through electromagnetic induction's principle, the secondary side of coupling mechanism receives the energy that once side was launched, and the receiving end is through the high frequency rectifier circuit rectification of high frequency alternating current that receives into direct current, provides the energy for the battery of Autonomous Underwater Vehicle (AUV) after filtering.
Underwater wireless charging requires energy transmission in an underwater environment, the transmission of electromagnetic waves and energy loss are affected by the medium of water, the dielectric constant in water is also higher than that in air or vacuum, and more energy loss is caused. In addition, the transparency and the depth of water can also influence the effect of underwater wireless transmission, the transparency of water can influence the penetrating power of electromagnetic waves, the depth can increase the distance of signal transmission and reduce the signal intensity, the depth and the change of the transparency of water can finally react on the change of the coupling coefficient, thereby influencing the power transmission capacity of the system, and when the receiving coil and the transmitting coil deviate, the power transmission of the underwater wireless charging system can be greatly influenced.
In order to solve the above problems, a compensation network is added in the wireless charging system, the design structure of the low-order compensation network is relatively simple, the resonance current of the compensation network is smaller, but huge power is required to be provided during starting, the stability requirement on the resonance circuit is high, the sensitivity to coil position deviation is larger, and the charging efficiency is reduced when the position of the receiving and transmitting coil changes. The high-order resonance compensation technology can add additional inductance and capacitance elements into the resonance circuit, can optimize the working state of the resonance circuit, reduce power loss and energy leakage, improve the charging efficiency, and can also reduce the sensitivity of the resonance circuit to distance change and improve the stability of the charging efficiency. The high-order resonance compensation network can be flexibly adjusted and upgraded according to actual needs, and has better expandability and adaptability.
The compensation network in this embodiment adopts a bilateral LCC resonance compensation network, and the structure of the bilateral LCC resonance compensation network is shown in fig. 1. The LCC compensation network comprises a power supply U d Primary and secondary side loops, the coupling mechanism comprising a transmitting coil L P2 And a receiving coil L S2 The method comprises the steps of carrying out a first treatment on the surface of the The primary side comprises a primary compensation capacitor C P1 、C P2 And primary side compensation inductance L P1 Switch tube V 1 、V 2 、V 3 And V 4 Forms a full bridge circuit, and the primary side compensates the inductance L P1 Primary side compensation capacitor C P2 And a transmitting coil L P2 Respectively connected at the middle points of the two bridge arms after being connected in series, and the primary side compensation capacitor C P1 Is connected in parallel with the primary compensation capacitor C P2 And a transmitting coil L P2 Is provided; the secondary side comprises a secondary compensation capacitor C S1 、C S2 And secondary side compensation inductance L S1 Receiving coil L S2 Secondary side compensation capacitor C S2 And secondary side compensation inductance L S1 And equivalent resistance R L In series in turn, compensating capacitor C S1 Is connected in parallel with the two ends of the secondary compensation capacitor C S2 And a receiving coil L S2 Is provided with a transmitting coil L P2 And a receiving coil L S2 And forming a mutual inductance model. Based on the bilateral LCC compensation network, two control methods of the wireless charging control system are provided through the following embodiments.
Example 1:
a control method of an underwater wireless charging system comprises the following steps:
S1, acquiring a plurality of system resonance frequencies { omega } 0 ，ω 1 ，ω 2 ，…，ω m }，ω 0 Is the natural resonant frequency omega 1 ，ω 2 ，…，ω m To sub-stabilize the resonant frequency.
As shown in fig. 1, for the primary side loop of the dual-side LCC compensation network of the present embodiment, due to the primary dc power internal resistance r and the primary compensation inductance L P1 And a transmitting coil L P2 Is small and is omitted in this embodiment. Definition lambda P To represent the primary compensation capacitance C of the system P1 And C P2 The ratio of the values of (2) is shown as the formula (1):
compensating the total input impedance Z of the primary side loop of the network in The method comprises the following steps:
wherein, the liquid crystal display device comprises a liquid crystal display device, 。
when the system is in a resonance state, the input impedance is expressed as pure resistance, namely the input impedance Z in The imaginary part is zero, so that a plurality of resonance frequency points of the double LCC resonance compensation network can be obtained. The resonance frequency point generally comprises a stable natural resonance frequency omega P0 And a plurality of sub-stable resonant frequencies { omega ] P1 ，ω P2 ，…，ω Pm }. The natural resonant frequency is not affected by the load, the output end of the natural resonant frequency presents the output characteristic of a current source, the metastable resonant frequency can change along with the change of the load, but the influence of the change of the load on the metastable resonant frequency is negligible only when the imaginary part of the input impedance of the system is far smaller than the real part of the input impedance. A frequency point satisfying the condition that the imaginary part of the input impedance of the system is far smaller than the real part of the input impedance is taken as a metastable resonance frequency.
In the present embodiment, the input impedance Z in The imaginary expression of (2) is shown in the formula (3):
the primary side compensation capacitor C P1 、C P2 Primary side compensation inductance L P1 Transmitting coil L P2 The constraint conditions are as follows:
when equation (4) is satisfied, equation (4) is taken into equation (3), and when the input impedance Zin is setWhen the imaginary part is 0, a natural resonant frequency omega which does not change with load can be obtained P0 And a resonance frequency point omega P1 、ω P2 Specifically, the method is shown as a formula (5):
at the resonant frequency omega P1 ，ω P2 If equation (6) is satisfied, the imaginary part of the system is considered to be much smaller than the real part, i.e., ω P1 ，ω P2 The two resonance frequencies are not changed with the change of the load, and are two metastable resonance frequency points.
R p The input impedance expression at the natural resonant frequency and the secondary resonant frequency can be obtained by substituting the expression (4) into the input impedance expression (2) for the primary impedance as shown in the expression (7). The output voltage and the output current of the double LCC resonance compensation system can be further calculated, and the system can be seen to be at the resonance frequency omega P0 Having parallel resonance characteristics, i.e. the output exhibits the output characteristics of a current source, as shown in formula (8), the resonance frequency being ω P1 ，ω P2 The output has series resonance characteristics, namely, the output characteristics of the output presentation voltage source are shown in formulas (9) and (10).
In the above-mentioned method, the step of,I p for the primary side current, the current is,Z ref is equivalent to impedance, U p Is the primary voltage. For the secondary side loop, similar to the primary side loop, the secondary side power transfer circuit does not consider the parasitic internal resistance of the element, defining lambda S Secondary compensation capacitor C for representing system S1 And C S2 The ratio of the values of (2) is represented by the formula (11):
input impedance Z of secondary LCC network of underwater wireless charging system S The method comprises the following steps:
wherein, the liquid crystal display device comprises a liquid crystal display device,ω=2pi f is the angular frequency of the input voltage.
Input impedance Z S The expression of the real and imaginary parts of (2) is:
the secondary compensation capacitor C S1 、C S2 Secondary side compensation inductance L S1 Receiving coil L S2 The constraint conditions are as follows:
the reactive power flowing through the inverter bridge causes additional loss, so that the LCC compensation network is made to be pure resistive, i.e. the imaginary part is made to be zero, and when the constraint condition shown in the formula (14) is satisfied, a natural resonant frequency omega is calculated S0 And two resonant frequencies omega S1 、ω S2 Resonant frequency omega S1 、ω S2 And the load R L In relation, when the equation (15) is satisfied, the imaginary part of the input impedance of the system is far smaller than the real part of the input impedance, and the equation is considered to accurately describe the resonant frequency thereof, and meets the requirement of the metastable resonant frequency.
And then the natural resonant frequency omega in the secondary side loop is obtained S0 And two metastable resonant frequencies omega S1 、ω S2 The method comprises the following steps:
as can be seen from equation (16), when the system is operating at ω S0 When the system is operated at omega, the input impedance is related to the inductance-capacitance value and the load impedance value of LCC compensation S1 And omega S2 The input impedance is related to the load and the compensation capacitance, and is independent of other parameters. Respectively omega S0 、ω S1 And omega S2 Substitution of input impedance Z S In the above, the input impedance of LCCs of the system operating at the three resonance points is calculated as:
when the resonant frequency is omega S0 When the output current is shown as a formula (18), the magnitude of the output current is irrelevant to a load, the LCC network shows constant current output characteristics, and the load current is as follows:
when the resonant frequency is omega S1 And omega S2 At this time, the LCC compensation network may exhibit a constant voltage output characteristic, the output voltage is independent of the load, and the load voltage is:
the LCC compensation network has a plurality of resonance frequencies irrelevant to load impedance in the primary side and secondary side loops, and lambda can be seen P ，λ S The method has influence on the resonance frequency, transmission power and coupling coefficient of the bilateral LCC resonance compensation network.
In the prior art, the charging mode of the storage battery is divided into constant current charging and constant voltage charging. In the initial stage of general charging, because the electromotive force of the storage battery is low, a constant-current charging mode is often adopted to avoid excessive charging current; and (3) in the later stage of charging, changing the electromotive force of the storage battery to a constant voltage charging mode to finish the residual charging as the electromotive force of the storage battery rises to a preset voltage value.
In this embodiment, in the constant current charging mode, the fixed resonant frequency ω of the primary side is obtained based on the method of the above embodiment Ip0 The metastable resonant frequency is omega Ip1 And omega Ip2 The fixed resonance frequency of the secondary side is omega IS0 The metastable resonant frequency is omega IS1 And omega IS2 In the constant voltage charging mode, the fixed resonance frequency ω of the primary side is obtained based on the method of the above embodiment Up0 And a metastable resonant frequency of omega Up1 The fixed resonance frequency of the secondary side is omega US0 And a metastable resonant frequency of omega US1 。
S2, selecting a coupling coefficient switching point { k } c1 ，k c2 ，…，k c(m-1) Dividing the coupling coefficient range into a plurality of coefficient intervals by the coupling coefficient switching point, wherein the system resonance frequency is matched with each coefficient interval;
when the bilateral LCC resonance compensation network in the underwater wireless charging system works under the characteristic of a constant current source, the current does not change along with the change of the load, and at the moment, omega is led to be changed Ip0 =ω IS0 =ω I0 To fix the resonant frequency omega Ip1 =ω IS1 ，ω Ip2 =ω IS2 For metastable resonant frequency, lambda P =λ S =λ, at natural resonant frequencyAnd obtaining the underwater wireless charging system with constant current source characteristics by the rate point and the two secondary stable resonance frequency points.
As shown in fig. 2, at ω Ip0 =ω IS0 =ω I0 When the secondary side impedance acts on the equivalent impedance Z of the primary side ref As shown in the formula (20), the load R flows without considering the loss L Is (1) the current of the (a)As shown in equation (21), the power output by the inverter is shown in equation (22).
At omega Ip1 =ω IS1 =ω I1 ，ω Ip2 =ω IS2 =ω I2 When ω is replaced by ω I1 、ω I2 Equivalent impedance Z of secondary side impedance acting on primary side ref As shown in the formula (23), the load R flows without considering the loss L Is (1) the current of the (a)As shown in equation (24), the power output by the inverter is shown in equation (25).
In order to enable the underwater wireless charging system to have a better connection process during frequency switching, the frequency switching is performed on the premise of ensuring stable transmission power, and the wireless charging system is used by a strategy of mutually switching multiple resonance frequency points to have better offset resistance. When the power at the natural resonant frequency is made equal to the power at the metastable resonant frequency point, even omega I0 Where the transmission power is equal to ω, respectively I1 ，ω I2 Transmission power at, i.e. commandThe coupling coefficient k at the resonant frequency switching position can be obtained Ic1 ，k Ic2 As shown in formulas (26) and (27).
In this embodiment, when the bilateral LCC resonance compensation network in the underwater wireless charging system works with the constant voltage source output characteristic, in order to make the system output at constant voltage, ω is made to be up1 =ω us0 =ω u1 ，ω pu0 =ω su1 =ω u2 The frequency relationship corresponds to the following:
lambda can be obtained according to formula (28) p And lambda is s The constraint relation of (2) is as follows:
Two resonant frequencies omega u1 And omega u2 The reflection impedance of the secondary side acting on the primary side is respectively as shown in the specification30 (31):
where k is a real-time coupling coefficient, where the coil structure, the magnetic core structure, and the magnetic medium are determined, the coupling coefficient k is only related to the spatial position, and when the relative position between the transmitting coil and the receiving coil is shifted, the coupling coefficient k is also changed correspondingly, and if the shift amount is larger, the coupling coefficient k is lower, and conversely, if the shift amount is smaller, the coupling coefficient and k are higher, and the change in the coupling coefficient causes a change in the power transmission characteristic.
Two resonant frequencies omega u1 And omega u2 The following output voltages are shown in the formulas (32) and (33), respectively:
two resonant frequencies omega u1 And omega u2 The following output powers are shown in the formulas (34) and (35), respectively:
as can be seen from equation (33), in the present embodiment, the resonance frequency ω is used u2 When power transmission is carried out, the coupling coefficient k gradually decreases along with the increase of the offset, and the output voltage of the system decreases along with the increase of the offset, so that the power is generatedThe transmission quality is degraded. If after the offset occurs, switch to ω u1 And (3) carrying out electric energy transmission, wherein the coupling coefficient is reduced as shown in a formula (32), the output voltage of the system is increased instead, and the electric energy transmission capability of the system can be ensured.
When the underwater wireless charging system charges the storage battery, excessive power fluctuation is avoided, particularly when the resonance frequency point is switched, the transmission power is ensured not to be changed, and a proper multi-resonance point switching strategy is needed to be adopted for the purpose. From equations (34) and (35), the output power is related not only to the resonant frequency, but also to the coupling coefficient, at resonant frequency ω u1 The lower output power decreases with decreasing coupling coefficient at the resonant frequency omega u2 The lower output power increases with decreasing coupling coefficient. Therefore, on the basis of the plurality of resonance points, the power transmission device is capable of matching the plurality of resonance points with the power output characteristics, so that the transmission power and the transmission efficiency of the wireless charging system are effectively improved, and compared with the mode of single resonance point adopted in the prior art, the power transmission device is capable of transmitting power in a manner of greatly limiting the freedom of power transmission.
In one embodiment, the coupling coefficient k corresponding to the power equal is found Uc At k Uc The frequency switching is performed, so that the transmission power can be kept unchanged. In the case of constraint (29), the coupling coefficient at the time of the available frequency switching is calculated as shown in equation (36):
will couple coefficient k Uc And as a switching point, the matched resonant frequency is selected according to the coefficient interval where the actual coupling coefficient is located, so that the stability and the high efficiency of electric energy transmission are improved.
S3, when the coupling coefficient is switched from the ith coefficient section to the (i+1) th or (i-1) th coefficient section, the (i)The time length required for switching the ith system resonant frequency to the ith+1 or the ith-1 system resonant frequency is t v The duration t v From n cycle times { T 1 ，T 2 ，…, T n The composition, i.e., satisfies the expression (37):
t v =T 1 +T 2 +…+T n （37）
wherein T is n Is the nth cycle time.
To connect the resonance frequency difference at both sides of the switching time, a time length t is formed v Is a linear increase in the time of the n cycles, i.e.:
T n =T 0 +n•ΔT（38）
wherein DeltaT is the increment of each period relative to the time of the previous period, and the duration T is v The cycle time of (a) gradually increases linearly. Each cycle time is linearly and slowly changed, so that the output waveform is always kept in a stable state, the waveform distortion of the system to a greater extent is prevented, and the influence on the power transmission efficiency is reduced.
Example 2:
the embodiment provides a control method of an underwater wireless charging system, which is different from the embodiment in that in the step S3, when the coupling coefficient is switched from the ith coefficient interval to the (i+1) th or (i-1) th coefficient interval, the ith system resonant frequency is switched to the (i+1) th or (i-1) th system resonant frequency, and the time period t required for the ith system resonant frequency to be switched to the (i+1) th or (i-1) th system resonant frequency v Approaching to 0, namely adopting a jump type switching mode, the jump type switching mode has high response speed, and realizes switching of proper resonant frequency according to the actual coupling coefficient. In comparison with example 1, dynamic oscillation is easily caused, and the system operation stability is affected.
The above embodiments are further described below in connection with specific simulation experiments.
In the constant current source wireless charging mode, parameters of the bilateral LCC resonance compensation system are selected as shown in table 1, and simulation analysis is carried out on multiple resonance points of the LCC resonance compensation system.
TABLE 1 simulation parameters for underwater Wireless charging System
And performing simulation analysis on multiple resonance points of the bilateral LCC resonance compensation system in a Matlab environment. FIG. 3 shows different loads R L A relationship between the lower output current and the operating frequency. The system has a plurality of constant current resonance points, wherein the output currents are 24.9A, 7.3A and 17.1A respectively at 69.4KHz, 85KHz and 98.15KHz, and the system has constant current output characteristics irrelevant to loads. The three resonance frequency points are all in resonance state and show pure resistance, wherein the inherent resonance frequency omega I0 At 85KHz, the metastable resonant frequency omega I1 And omega I2 98.15KHz and 64.9KHz, respectively.
Fig. 4a shows waveforms of the inverter output voltage and current at the natural resonant frequency when the coupling coefficient is equal to 0.5, and fig. 4b and fig. 4c show waveforms of the inverter output voltage and current at the metastable resonant frequency when the coupling coefficient is equal to 0.5, respectively. The AUV has good transmission capability at the natural resonant frequency in the state of alignment of the receiving coil and the transmitting coil in the charging dock. At the metastable resonant frequency, the transmission capability is obviously reduced, the phase difference of the voltage and current waveforms of the output of the inverter is larger, and the system generates excessive reactive power. The result of fourier analysis of the current waveform in fig. 4 is shown in fig. 5, which shows that the current distortion rate is small at the natural resonant frequency, the system loss is small, and the current distortion rate is large at the metastable resonant frequency, so that more loss is generated. Fig. 4 and 5 show that the natural resonant frequency is more favorable for energy transmission in a state of higher coupling coefficient, and the transmission power and transmission efficiency of the system are higher.
Fig. 6a shows waveforms of output voltage and current of the inverter at the natural resonant frequency of 85KHz when the coupling coefficient is equal to 0.3, and fig. 6b and fig. 6c show waveforms of output voltage and current of the inverter at the metastable resonant frequency when the coupling coefficient is equal to 0.3, respectively. The AUV may shift in the charging dock, resulting in a decrease in coupling coefficient, in this embodiment, the coupling coefficient is 0.3, as shown in fig. 6a, the output current of the inverter is distorted, and a larger peak current is generated during switching of the device, resulting in very large switching loss, and lower power and transmission efficiency of the system. When the resonant frequency is switched to 98.15KHz, the output waveform of the inverter is obviously improved and is more similar to a sine wave, the transmission power and the transmission efficiency are obviously improved, but a larger phase difference exists between the voltage and the current output by the inverter, so that more reactive power and loss are generated in the system. When the resonance frequency is switched to 64.9KHz, the waveform of the output current of the inverter is similar to a sine wave compared with the output current of the inverter at the natural resonance frequency, the phase difference between the output voltage and the current of the inverter is almost zero, and compared with the resonance frequency of 98.15KHz, the output power and the output efficiency of the underwater wireless charging system are further improved.
Fourier analysis was performed on the inverted currents in fig. 6a, 6b and 6c, and the results are shown in fig. 7. Fig. 7a shows harmonic analysis of the inverter current at the natural resonant frequency of the system, and it can be found that the total current distortion is 26.15% when the coupling coefficient is 0.3, the harmonic current is large in duty ratio, and the duty ratio is the third harmonic except the fundamental current. Fig. 7b and 7c are current harmonic analyses at the metastable resonant frequency, and it can be found that the system works at the metastable resonant frequency point, the three times, the five times of harmonic duty ratio are greatly inhibited, and the total distortion rate of the current is obviously reduced. According to the analysis, when the double LCC resonance compensation system operates at the natural resonance frequency, dislocation is generated, so that after the coupling coefficient is reduced, the third harmonic content is high, the current waveform is severely distorted, and the system quality factor is poor. In contrast, under the same parameter condition, the third harmonic content is obviously reduced when the system works at the subresonance frequency, the waveform of the current approximates to a sine wave, the system has a higher quality factor, the function of well inhibiting the third harmonic is achieved, the problem of current waveform distortion caused by dislocation is effectively avoided, and the anti-offset capability of the system is improved.
As shown in fig. 8, the power of the underwater wireless charging increases with an increase in the coupling coefficient at the natural resonant frequency, and the charging power is not in direct proportion to the coupling coefficient at the metastable resonant frequency. When the coupling coefficient is reduced from 0.4 to 0.3 only with the natural resonant frequency, the power is reduced by 0.7KW, the efficiency is reduced from 85.9% to 78.1%, and when the coupling coefficient is reduced from 0.4 to 0.3 with the metastable resonant frequency only, the power is increased by 1.55KW, and the efficiency is increased from 75.3% to 92.4%. It is understood that the higher the coupling coefficient is, the better the power transmission effect is at the natural resonant frequency, and the lower the coupling coefficient is, the better the transmission effect is at the metastable resonant frequency. If the charging strategy with the double resonant frequencies being switched is adopted, the fluctuation range of power can be greatly reduced, the efficiency of system power transmission can be improved, and the anti-offset capability of the underwater wireless charging system is improved to a certain extent.
In the underwater wireless charging system, when the constant voltage source wireless charging condition is verified, parameters of the selected bilateral LCC resonance compensation system are shown in table 2, and at the moment, the output voltage is not changed along with the change of the load. Lambda in Table 2 P 、λ S Is carried into a formula (36) to obtain a coupling coefficient k during frequency switching Uc 0.4. When the real-time coupling coefficient is greater than 0.4, ω is used U2 Carrying out underwater wireless charging system transmission; when the real-time coupling coefficient is less than 0.4, ω is used U1 Carrying out underwater wireless charging system transmission; when the real-time coupling coefficient is equal to 0.4, the frequency switching can effectively ensure the stable transmission power of the system.
Table 2 simulation parameters of underwater Wireless charging System
In the case of different loads R L In the case of scanning the system frequency at 30KHz-110KHz, the system has multiple constant voltage resonance points. FIG. 9 is a graph showing the relationship between the gain of the input and output voltage and the operating frequency of the system under different load conditions, the gain of the voltage is independent of the load at 68KHz, 73.5KHz, 89.7KHz and 95KHz, and the experience is thatThe system shows pure resistance when the resonance frequency is 73.5KHz and 89.7KHz, which is consistent with the calculation result, so that the constant voltage charge of the system adopts 73.5KHz and 90KHz as the resonance frequency of the system.
Since the AUV may shift in position in the charging dock, the coupling coefficient may decrease, and a case where the coupling coefficient is less than 0.4 may occur. Fig. 10 is a waveform of voltage and current output by the inverter at the resonance frequency of 73.5kHz and 90kHz when the real-time coupling coefficient is 0.2, and fig. 10a is a waveform of voltage and current output by the inverter at the resonance frequency of 73.5kHz when the coupling coefficient is 0.2, and it can be seen that the current waveform is a sine wave, the current amplitude is obviously increased, and the phase difference between the output voltage and the current of the inverter is almost zero, which indicates that the output power and the output efficiency of the underwater wireless charging system are obviously improved when the resonance frequency is switched to 73.5 kHz. Fig. 10b shows waveforms of voltage and current output by the inverter when the coupling coefficient is 0.2 and the resonant frequency is 90kHz, and the waveform of the current is non-sinusoidal, which indicates that harmonic loss is easy to generate due to larger harmonic content, particularly, larger peak current is easy to generate in the switching process of the device, so that the switching loss is very large, the phase difference between the output voltage and the current of the inverter is relatively large, and the transmission power and the transmission efficiency of the system are reduced.
Fourier analysis was performed on the inverted currents in fig. 10a and 10b, and the results are shown in fig. 11. FIG. 11a shows the result of harmonic analysis of current at a resonant frequency of 73.5kHz, with the ratio of third and fifth harmonics significantly reduced, and the total current distortion rate of 1.12%. From the above analysis, it can be seen that if the system is at the resonance frequency ω after the coil is shifted to cause the coupling coefficient to decrease U2 Continuing to run, the current waveform is seriously distorted, so that the quality factor of the system is deteriorated; if the resonant frequency is switched to omega U1 When working downwards, the harmonic content of the system is obviously reduced, and the quality factor of the system can be improved. Therefore, the resonance frequency point switching mode is adopted, the problem of current waveform distortion caused by offset can be effectively avoided, and the transmission efficiency and the offset resistance of the system can be improved. FIG. 11b shows the result of harmonic analysis of current at a resonance frequency of 90kHz, showing that the total distortion of the current is 10.16% and the third harmonic is about when the coupling coefficient is 0.2The fundamental content is 10%, and the fifth harmonic is about 5% of the fundamental content.
If the AUV has a small offset between the receiving coil and the transmitting coil in the charging dock, a coupling coefficient greater than 0.4 may occur. Fig. 12 is a simulation waveform of voltage and current output by the inverter at 73.5kHz and 90kHz resonant frequency when the real-time coupling coefficient is equal to 0.6. As can be seen from the figure, the system is at resonant frequency ω U2 With good transmission capacity at the resonance frequency omega U1 The lower transmission capability is obviously reduced, and the phase difference of the voltage and current waveforms of the output voltage and current of the inverter is larger, so that the system generates excessive reactive power. Fourier analysis of the current waveforms in fig. 12a and 12b, the results being shown in fig. 13, shows that at the resonance frequency ω U2 The harmonic content of the lower current is small and at the resonant frequency omega U1 The lower current has larger harmonic content and can generate more harmonic loss.
From the simulation results of fig. 12 and 13, it is shown that the coupling coefficient is larger at the resonance frequency ω when the AUV is less positionally offset in the charging dock U2 The lower work is more beneficial to energy transmission, the current waveform distortion rate is small, and the power and the efficiency of the system transmission are higher. When the resonant frequency is matched with the coupling coefficient, the amplitude of the output current of the inverter is large, the waveform distortion is small, the waveform sine degree is good, the phase difference between the output voltage and the current is small, the generated reactive power is small, and the system has higher transmission power and transmission efficiency. Because the wireless charging system adopts the constant voltage source output system to carry out electric energy transmission, when the system has an open circuit fault, the transmission power becomes zero, and the system can be automatically protected.
In order to further illustrate that the multi-resonant frequency transmission adopted by the invention has the same coupling structure, the conventional double-side LCC resonance compensation system is taken as a comparison example, and simulation comparison analysis is carried out on the double-side LCC resonance compensation system switched by the multi-resonant points of the embodiment. In the comparative example, the conventional bilateral LCC resonance compensation system works at a resonance frequency of 85kHz, and the bilateral LCC resonance compensation systems with multi-resonance-point switching are respectively arranged at omega U1 And omega U2 Operating at resonance frequency, the simulation result of the transmission power and transmission efficiency is shown in FIG. 14As shown. As can be seen from the figure, the transmission characteristics of the conventional bilateral LCC resonance compensation system and the resonance frequency ω thereof U2 The transmission characteristics are similar, the coupling coefficient is larger than 0.4, the transmission capability is good, the transmission capability is poor when the coupling coefficient is smaller than 0.4, but the system is at the resonance frequency omega under the low coupling coefficient U1 The transmission capacity is obviously enhanced, so that the transmission power and the transmission efficiency of the system can be improved by adopting a multi-resonance point switching mode.
As shown by the simulation results of the comparative example and the embodiment, in the process that the coupling coefficient is changed from 0.2 to 0.6, the transmission power of the traditional bilateral LCC compensation system is reduced by about 1.9kW, and the transmission efficiency is reduced by about 8%; the transmission power of the system adopting multi-resonance point switching is reduced by about 1.2kW, and the transmission efficiency is reduced by about 5.5%. Therefore, the bilateral LCC resonance compensation system adopting multi-resonance point switching is described to be capable of remarkably improving transmission power and transmission efficiency, so that the system has higher stability and anti-offset capability.
In one embodiment, the effect of a linearly varying resonant frequency switching pattern, which refers to a pattern that allows a period of time to vary linearly during frequency switching, is verified. As shown in fig. 15, t 1 Time=0.2s is the time point when the frequency starts to switch, t 2 The moment is the time point of completion of the frequency switch, t v The duration required for frequency switching. Taking the example of the dual resonance point switching constant voltage output system of the above embodiment as well, i.e. at t 1 Before the moment, the working frequency of the system is 90KHz, at t 2 After the moment, the working frequency of the system is 73.5KHz, and the time period t v The cycle time is linearly increased.
FIG. 16 is a waveform of the driving signal and the inverter input current during the linear variable frequency switching, and the simulation setting time period t v The switching of the resonant frequency from 90kHz to 73.5kHz is completed by 1000 times of transformation within 0.1s, fig. 16 only shows the waveform change condition of five periods after the switching is started, the system works near the resonant point, the obtained current waveform is a sine wave, and the output waveform is very stable due to the linear slow change of each period, and the current amplitude change is stable.
As shown in fig. 17, the voltage waveform input by the rectifier at the receiving end of the underwater wireless charging system is shown, after the frequency switching occurs in the frequency switching mode of linear change, the input voltage of the rectifier generally shows a linear reduction trend, and finally, the voltage tends to be stable, and the fluctuation of the voltage and the frequency is smaller.
In a specific embodiment, the effect of the direct jump resonant frequency switching mode is verified, the direct jump resonant frequency switching mode is changed immediately at the instant of frequency switching, taking the underwater wireless charging system with dual resonance point constant voltage output of the above embodiment as an example, the switching from 90KHz to 73.5KHz is completed, and the output voltage of the inverter is shown in fig. 18. the time t is the frequency switching point, the working frequency is 90KHz before the period of the time t, and the frequency is switched to 73.5KHz immediately after the period is finished, namely after the time t.
Fig. 19 shows waveforms of driving signals and input voltage of the inverter during the jump frequency switching, that is, the resonant frequency is 90kHz before 0.2s, and the jump is to 73.5kHz instantly after 0.2s, as can be seen by comparing fig. 19 with fig. 16, when the jump frequency switching, the output current of the inverter generates a greater degree of distortion, and in several periods after the switching time of 0.2s, the system works in a non-resonant state, the current waveform is non-sinusoidal, the inverter circuit works unstably, and a greater electromagnetic interference is possibly generated, so that the risk of direct connection occurs between two power tubes of one bridge arm.
As shown in fig. 20, the voltage waveform input by the rectifier at the receiving end of the underwater wireless charging system is switched in frequency at 0.2 s. After the frequency switching occurs, the input voltage of the rectifier generally presents a situation that the input voltage is increased and then reduced and then tends to be stable, and larger frequency and voltage fluctuation is generated in the switching period, so that electromagnetic interference is increased and abnormal system operation is possibly caused.
According to the underwater wireless charging system taking the bilateral LCC as the compensation network, based on a mutual inductance coupling model, a plurality of resonance points and electric energy transmission characteristics are matched through a frequency relation, proper resonance frequencies are selected under different coupling coefficients, and comparison analysis of output power, transmission efficiency and current harmonic waves at the two resonance points of the system proves that the system has stronger power output capacity at a secondary resonance point after the coupling coefficient is reduced to a certain degree due to offset. By comparing the difference between the jump-type frequency switching and the linear change-type frequency switching and combining the simulation analysis of the underwater wireless charging system compensated by the bilateral LCC, the method proves that the linear change-type frequency switching mode has a larger advantage compared with the jump-type frequency switching in terms of the output of the inverter at the transmitting end of the system and the input of the rectifier at the receiving end of the system, the distortion of the voltage and current waveforms is obviously reduced, the switching process is safer and more reliable, and the feasibility is verified. The resonant frequency adopts a linear smooth switching mode and a jump type switching mode, so that the linear smooth switching mode can be adopted for places with high system stability requirements, and the jump type switching mode can be completely adopted for places with low precision and stability requirements.
The above embodiments are merely illustrative of the principles of the present invention and its effectiveness, and are not intended to limit the invention. Modifications and variations may be made to the above-described embodiments by those skilled in the art without departing from the spirit and scope of the invention. Accordingly, it is intended that all equivalent modifications and variations of the invention be covered by the claims, which are within the ordinary skill of the art, be within the spirit and scope of the present disclosure.
Claims (8)
1. Control method of underwater wireless charging system, wherein the underwater wireless charging system comprises a bilateral LCC compensation network, and a primary side loop of the bilateral LCC compensation network is connected with a transmitting coil L P2 The secondary side loop is connected with the receiving coil L S2 The primary side loop comprises a primary side compensation capacitor C P1 、C P2 And primary side compensation inductance L P1 The secondary side comprises a secondary compensation capacitor C S1 、C S2 And secondary side compensation inductance L S1 The control method is characterized by comprising the following steps:
obtaining a plurality of system resonance frequencies { omega } 0 ，ω 1 ，ω 2 ，…，ω m }, wherein omega 0 Is the natural resonant frequency omega 1 ，ω 2 ，…，ω m Is the metastable resonant frequency;
selecting a coupling coefficient switching point { k c1 ，k c2 ，…，k cm Dividing the coupling coefficient range into a plurality of coefficient intervals by the coupling coefficient switching point, wherein the system resonance frequency is matched with each coefficient interval;
When the coupling coefficient is switched from the ith coefficient interval to the (i+1) th or the (i-1) th coefficient interval, the ith system resonant frequency is switched to the (i+1) th or the (i-1) th system resonant frequency, and the time length t required by the (i) th system resonant frequency to the (i+1) th or the (i-1) th system resonant frequency v The duration t v Consists of n cycle times which vary in a linear increase.
2. A control method of an underwater wireless charging system according to claim 1, wherein a time period t required for the i-th system resonance frequency to be switched to the i+1-th or i-1-th system resonance frequency v The method comprises the following steps:
t v =T 1 +T 2 +…+T n ；
wherein T is n For the nth cycle time, it is: t (T) n =T 0 +n•ΔT，T 0 For the period time before the switching time, Δt is the increment of each period time relative to the previous period time.
3. The control method of an underwater wireless charging system according to claim 1, wherein the method of obtaining the resonance frequency comprises:
calculating the input impedance of the bilateral LCC compensation network, and acquiring a resonance frequency point when the imaginary part of the input impedance is zero;
the resonance frequency point which is not affected by the load is taken as the natural resonance frequency, and the resonance frequency point which meets the condition that the imaginary part of the input impedance of the system is far smaller than the input impedance is taken as the metastable resonance frequency.
4. The method of claim 1, wherein when the bilateral LCC compensation network is operated in a constant voltage source mode, the primary compensation capacitor C P1 、C P2 Primary side compensation inductance L P1 And a transmitting coil L P2 The constraint conditions are as follows:
L P2 =(1+λ p )L P1 ；
wherein lambda is p = C P1 / C P2 。
5. The method of claim 4, wherein the secondary compensation capacitor C S1 、C S2 Secondary side compensation inductance L S1 And a receiving coil L S2 The constraint conditions are as follows:
L S2 =(1+λ S )L S1 ；
wherein lambda is S = C S1 / C S2 。
6. The method of claim 5, wherein λ is determined when the underwater wireless charging system is operating in a current source mode p And lambda (lambda) S The constraint relation between the two is as follows:
；
lambda when the underwater wireless charging system is operating in a voltage source mode p And lambda (lambda) S The constraint relation between the two is as follows:
。
7. the control method of an underwater wireless charging system according to claim 1, wherein the coupling coefficient switching point determining method is as follows:
calculating inverter output power at natural resonant frequencyAnd inverter output power at m metastable resonant frequencies +.>；
Inverter output power at natural resonant frequency Inverter output power at ith metastable resonant frequencyThe coupling coefficient at the same time is the ith coupling coefficient switching point k ci 。
8. A control method of an underwater wireless charging system is characterized in that the underwater wireless charging system comprises a bilateral LCC compensation network, and a primary side of the bilateral LCC compensation network comprises a primary side compensation capacitor C P1 、C P2 Primary side compensation inductance L P1 And a transmitting coil L P2 The secondary side comprises a secondary compensation capacitor C S1 、C S2 Secondary side compensation inductance L S1 And a receiving coil L S2 The control method is characterized by comprising the following steps:
obtaining a plurality of system resonance frequencies { omega } 0 ，ω 1 ，ω 2 ，…，ω m }；
Selecting a coupling coefficient switching point { k c1 ，k c2 ，…，k cm Dividing the coupling coefficient range into a plurality of coefficient intervals by the coupling coefficient switching point, wherein the system resonance frequency is matched with each coefficient interval;
when the coupling coefficient is switched from the ith coefficient interval to the (i+1) th or the (i-1) th coefficient interval, the time length required for switching the ith system resonance frequency to the (i+1) th or the (i-1) th system resonance frequency is approximately 0.
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