CN116582006A - Coordination control method for three-phase-single-phase multi-level converter - Google Patents
Coordination control method for three-phase-single-phase multi-level converter Download PDFInfo
- Publication number
- CN116582006A CN116582006A CN202310714906.9A CN202310714906A CN116582006A CN 116582006 A CN116582006 A CN 116582006A CN 202310714906 A CN202310714906 A CN 202310714906A CN 116582006 A CN116582006 A CN 116582006A
- Authority
- CN
- China
- Prior art keywords
- phase
- voltage
- current
- axis
- inverter
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
- 238000000034 method Methods 0.000 title claims abstract description 35
- 230000001629 suppression Effects 0.000 claims abstract description 58
- 239000003990 capacitor Substances 0.000 claims description 57
- 230000014509 gene expression Effects 0.000 claims description 37
- 230000009466 transformation Effects 0.000 claims description 25
- 238000006243 chemical reaction Methods 0.000 claims description 19
- 238000011217 control strategy Methods 0.000 claims description 19
- 230000003068 static effect Effects 0.000 claims description 16
- 239000011159 matrix material Substances 0.000 claims description 12
- 230000008859 change Effects 0.000 claims description 5
- 230000010363 phase shift Effects 0.000 claims description 5
- 239000013598 vector Substances 0.000 description 17
- 238000005070 sampling Methods 0.000 description 11
- 238000010586 diagram Methods 0.000 description 10
- 230000009286 beneficial effect Effects 0.000 description 8
- 238000004422 calculation algorithm Methods 0.000 description 7
- 239000004973 liquid crystal related substance Substances 0.000 description 6
- 238000004458 analytical method Methods 0.000 description 4
- 230000002829 reductive effect Effects 0.000 description 4
- 230000007704 transition Effects 0.000 description 4
- 230000000295 complement effect Effects 0.000 description 3
- 238000007599 discharging Methods 0.000 description 3
- 238000001914 filtration Methods 0.000 description 3
- 230000001939 inductive effect Effects 0.000 description 3
- 230000007935 neutral effect Effects 0.000 description 3
- 230000008569 process Effects 0.000 description 3
- 230000004044 response Effects 0.000 description 3
- 230000001360 synchronised effect Effects 0.000 description 3
- 238000004364 calculation method Methods 0.000 description 2
- 230000000694 effects Effects 0.000 description 2
- 238000005516 engineering process Methods 0.000 description 2
- 230000000670 limiting effect Effects 0.000 description 2
- 230000002441 reversible effect Effects 0.000 description 2
- 230000009471 action Effects 0.000 description 1
- 238000000098 azimuthal photoelectron diffraction Methods 0.000 description 1
- 230000005540 biological transmission Effects 0.000 description 1
- 230000015572 biosynthetic process Effects 0.000 description 1
- 238000000354 decomposition reaction Methods 0.000 description 1
- 230000007547 defect Effects 0.000 description 1
- 238000009795 derivation Methods 0.000 description 1
- 230000004069 differentiation Effects 0.000 description 1
- 230000009977 dual effect Effects 0.000 description 1
- 238000000605 extraction Methods 0.000 description 1
- 230000006698 induction Effects 0.000 description 1
- 230000002401 inhibitory effect Effects 0.000 description 1
- 238000002347 injection Methods 0.000 description 1
- 239000007924 injection Substances 0.000 description 1
- 238000011068 loading method Methods 0.000 description 1
- 238000003786 synthesis reaction Methods 0.000 description 1
Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M5/00—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases
- H02M5/40—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc
- H02M5/42—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters
- H02M5/44—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac
- H02M5/453—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal
- H02M5/458—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/12—Arrangements for reducing harmonics from ac input or output
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/14—Arrangements for reducing ripples from dc input or output
Landscapes
- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Rectifiers (AREA)
Abstract
The invention discloses a coordination control method of a three-phase-single-phase multi-level converter, which belongs to the technical field of electric energy quality, and comprises the steps of connecting a three-phase rectifier with a single-phase inverter through an extensible symmetrical half-bridge decoupling circuit to obtain the three-phase-single-phase converter; performing secondary ripple suppression and characteristic sub-current harmonic suppression of the three-phase rectifier according to the three-phase-single-phase converter, and adding the secondary ripple suppression signal and the characteristic sub-current harmonic suppression signal which are respectively obtained to obtain a third bridge arm modulation wave; controlling a third bridge arm of the three-phase rectifier by using the third bridge arm modulation wave; and according to the three-phase-single-phase converter, characteristic subvoltage harmonic suppression of the single-phase inverter is carried out, an inverter modulation wave is obtained, and coordination control of the three-phase-single-phase multi-level converter is completed. The invention solves the problems that the three-phase-single-phase multi-level converter has unbalance of direct-current side voltage and secondary ripple, and the three-phase input current and single-phase inversion output voltage have harmonic waves.
Description
Technical Field
The invention belongs to the technical field of electric energy quality, and particularly relates to a coordination control method of a three-phase-single-phase multi-level converter.
Background
The output side of the three-phase-single-phase converter is a single-phase inverter, and the current of the single-phase converter at the alternating-current side can be controlled to be sinusoidal current with the same frequency as the voltage no matter in a rectifying or inverting state, so that the conversion system has a very high power factor. However, sinusoidal fluctuating currents and voltages generate pulsating power that is doubled to the grid voltage frequency, which in turn generates secondary voltage ripple on the dc side. The secondary ripple wave of the direct current side voltage injects three positive sequence harmonic waves and single negative sequence harmonic waves into the three-phase input current through the voltage control loop, and injects three order harmonic waves and single harmonic waves into the single-phase output voltage. The existing method for inhibiting the secondary voltage fluctuation at the direct current side mainly comprises two types, namely, a control algorithm is improved to prevent secondary ripple waves from entering a control loop and inhibit the generation of low-order harmonic waves at the network side; the first type is to eliminate secondary ripple wave at DC side and inhibit generation of low harmonic wave at net side by adding hardware filter circuit.
However, the control algorithm is rarely adopted in the current engineering application, because the control algorithm not only has the problem of compensation precision, but also can narrow the bandwidth of the voltage loop to different degrees, and the response speed of the control loop is reduced. Meanwhile, the control algorithm can only prevent secondary ripple waves generated on the direct current side from entering the control loop, introduces other subharmonics to the alternating current input side of the front end, and cannot directly inhibit secondary fluctuation voltage on the direct current side, and the secondary fluctuation voltage on the direct current side still exists. The traditional hardware filtering adopts a large capacitor or an LC resonant circuit, has large volume and short service life, influences the reliability of the system and reduces the power density of the system. The active power decoupling APD (active power decoupling) can reduce inductance, capacitance and system volume while filtering out secondary ripple, reduce cost and improve power density.
The existing APD circuit is applied to a two-level circuit, however, in a multi-level converter circuit, the problem of unbalance of upper and lower voltages exists, a voltage balancing circuit is needed to be added, a software voltage balancing mode and a hardware voltage balancing mode exist, the software voltage balancing control method is used for counteracting or eliminating the offset of midpoint potential by adjusting the acting time of a redundancy vector, the balance of midpoint potential is realized, the software voltage balancing control method can reduce the size, the weight and the cost of an inverter and improve the power density, but the control algorithm becomes complex and difficult to realize, and the response speed of a control loop is reduced. The hardware voltage equalizing mainly comprises inductance voltage equalizing and capacitance voltage equalizing, and because the current of the capacitance element can be suddenly changed, a resistance element is needed to be added for current limiting in actual use, and the efficiency of the converter is reduced after the current limiting resistor is added, so that an inductance voltage equalizing circuit is mostly used in actual use.
The three-phase-single-phase three-level converter not only needs an inductance voltage equalizing circuit and a secondary ripple suppression circuit, but also increases the number, the volume, the weight and the cost of devices of the converter.
Disclosure of Invention
Aiming at the defects in the prior art, the coordination control method of the three-phase-single-phase multi-level converter solves the problems that the three-phase-single-phase multi-level converter has unbalance of direct-current side voltage and secondary ripple and has harmonic waves of three-phase input current and single-phase inversion output voltage.
In order to achieve the aim of the invention, the invention adopts the following technical scheme: a coordination control method of a three-phase-single-phase multi-level converter comprises the following steps:
s1, connecting a three-phase rectifier and a single-phase inverter through an extensible symmetrical half-bridge decoupling circuit to obtain a three-phase-single-phase converter;
s2, performing secondary ripple suppression of the three-phase rectifier according to the three-phase-single-phase converter to obtain a secondary ripple suppression signal;
s3, carrying out characteristic sub-current harmonic suppression of the three-phase rectifier according to the three-phase-single-phase converter to obtain a characteristic sub-current harmonic suppression signal;
s4, adding the secondary ripple suppression signal and the characteristic secondary current harmonic suppression signal to obtain a third bridge arm modulation wave;
s5, controlling a third bridge arm of the three-phase rectifier by utilizing the third bridge arm modulation wave;
s6, according to the three-phase-single-phase converter, characteristic subvoltage harmonic suppression of the single-phase inverter is carried out, inverter modulation waves are obtained, the single-phase inverter is controlled based on the inverter modulation waves, and coordination control of the three-phase-single-phase multi-level converter is completed.
The beneficial effects of the invention are as follows: the invention can effectively inhibit secondary ripple of the DC bus voltage of the three-phase-single-phase converter and realize capacitor voltage balance, and simultaneously, the three-phase input current and the single-phase output voltage have good sine; on the basis of the direct-current side supporting capacitor of the three-phase-single-phase converter, the filter inductor is increased, and compared with the traditional active filter and voltage equalizing circuit, the filter inductor has the advantages of simple structure, fewer devices, small volume and easiness in engineering application.
Further, the scalable symmetrical half-bridge decoupling circuit in the step S1 specifically includes: for an n-level three-phase rectifier, the scalable symmetrical half-bridge decoupling circuit includes (n-1)/2 symmetrical half-bridge decoupling circuits; the symmetrical half-bridge decoupling circuits comprise two direct-current side supporting capacitors and a second direct-current side supporting capacitor, wherein the two direct-current side supporting capacitors and the second direct-current side supporting capacitor are arranged between the three-phase rectifier and the single-phase inverter; a filter inductor is arranged between the two direct-current side supporting capacitors; the filter inductor is connected with the three-phase rectifier.
The beneficial effects of the above-mentioned further scheme are: the switching tube of the third bridge arm of the rectifier is used for replacing the switching tube of the symmetrical half-bridge decoupling circuit, the filter inductance is increased only on the basis of the direct-current side supporting capacitance of the three-phase-single-phase converter, and compared with the traditional active filter and voltage equalizing circuit, the filter inductance is simple in structure, few in devices, small in size and easy to engineer and apply.
Further, the step S2 specifically includes:
s201, obtaining a direct-current side secondary ripple voltage according to a three-phase-single-phase converter;
s202, obtaining a basic component in a static coordinate system according to the secondary ripple voltage of the direct current side;
s203, obtaining a secondary ripple suppression signal according to the basic components in the static coordinate system.
The beneficial effects of the above-mentioned further scheme are: and the switching-off of a switching tube of a third bridge arm of the three-phase rectifier is controlled by collecting the secondary ripple voltage and the inductive current of the direct current side, so that the alternating current components of the supporting capacitor voltage of the direct current side are complemented, and the direct current side has no voltage fluctuation. And outputting PR controller in the secondary ripple suppression loop as a secondary ripple suppression signal, adding the secondary ripple suppression signal with a characteristic secondary current harmonic suppression signal of a third bridge arm output by a three-phase rectifier control strategy, finally generating a third bridge arm modulation wave to the third bridge arm of the three-phase rectifier, controlling the AC component of the DC side supporting capacitor voltage to complement, compensating the secondary power, suppressing the secondary ripple of the DC side, and finally enabling the electric energy quality of the three-phase network side current and the output voltage to be good.
Further, the step S201 specifically includes:
s2011, obtaining direct-current voltage and a given direct-current voltage value according to a three-phase-single-phase converter;
s2012, obtaining a DC voltage deviation value according to the DC voltage and a given DC voltage value;
s2013, obtaining the secondary ripple voltage of the direct current side by utilizing a double-frequency band-pass filter according to the direct current voltage deviation value.
The beneficial effects of the above-mentioned further scheme are: and acquiring the secondary ripple voltage of the direct current side, and preparing for acquiring the basic components in the static coordinate system.
Further, the step S202 specifically includes:
s2021, obtaining output voltage according to the three-phase-single-phase converter;
s2022, obtaining an output side phase by using a single-phase-locked loop according to the output voltage;
s2023, shifting the phase of the secondary ripple voltage of the direct current side by 1/4 period, setting the delay time to be 2.5ms, and obtaining a virtual control variable;
s2024, obtaining a basic component in a static coordinate system by utilizing a transformation matrix according to the virtual control variable, the direct-current side secondary ripple voltage and the output side phase; the expression of the transformation matrix is as follows:
wherein T is trans Is a transformation matrix; ωt is the output side phase; θ is the phase angle of the filtered inductor current.
The beneficial effects of the above-mentioned further scheme are: and acquiring a basic component in the static coordinate system, and preparing for acquiring the secondary ripple suppression signal.
Further, the step S203 specifically includes:
s2031, obtaining inductance reference current according to basic components in a static coordinate system;
s2032, obtaining inductance current according to the three-phase-single-phase converter;
s2033, obtaining inductance current deviation according to the inductance current and the inductance reference current;
and S2034, obtaining a secondary ripple suppression signal by using the PR controller according to the inductance current deviation.
The beneficial effects of the above-mentioned further scheme are: and obtaining an inductance reference current according to the basic components in the static coordinate system, and obtaining a secondary ripple suppression signal through the PR controller.
Further, the step S3 specifically includes:
s301, according to the DC voltage deviation value, a band-stop filter and a PI controller are utilized to inhibit voltage ripple, and a d-axis current reference value is obtained:
wherein i is sd * Is a d-axis current reference value;the d-axis reference current direct current is used as the reference current direct current;
s302, performing a three-phase rectifier DQ decoupling control strategy according to the three-phase-single-phase converter and the d-axis current reference value to obtain a three-phase rectifier q-axis voltage modulation signal and d-axis voltage direct current under a three-phase rectifier DQ coordinate system;
s303, acquiring the secondary fluctuation amplitude value of the d-axis modulation signal of the three-phase rectifier and the secondary fluctuation phase of the d-axis modulation signal of the three-phase rectifier:
wherein u is d2 The amplitude of the secondary wave quantity of the d-axis modulation signal of the three-phase rectifier is obtained; beta is the phase of the secondary fluctuation of the d-axis modulation signal of the three-phase rectifier; m is m s The modulation degree of the rectifier;a direct current amount which is a direct current side voltage; c is the equivalent capacitance value of the direct current side; phi (phi) o The phase angle of the output current and the voltage is inverted; alpha is the phase angle of the inverted output voltage; u (U) o Is the output voltage amplitude; i o Is the output current amplitude; omega s The voltage frequency is three-phase network side voltage frequency;
s304, obtaining a three-phase power frequency negative sequence modulation signal and a three-phase positive sequence modulation signal according to the secondary wave amplitude value of the d-axis modulation signal of the three-phase rectifier and the secondary wave phase of the d-axis modulation signal of the three-phase rectifier:
wherein u is a- * (t) is the single negative sequence component of the a-phase voltage modulated signal; u (u) b- * (t) is the single negative sequence component of the b-phase voltage modulated signal; u (u) c- * (t) is the single negative sequence component of the c-phase voltage modulated signal; u (u) a3 * (t) is the positive three-order component of the a-phase voltage modulation signal; u (u) b3 * (t) is the b-phase voltage modulation signal cubic positive sequence component; u (u) c3 * (t) is the c-phase voltage modulation signal cubic positive sequence component; omega is the frequency; t is time; ωt is the output side phase;
s305, performing DQ conversion according to the three-phase power frequency negative sequence modulation signal and the three-phase positive sequence modulation signal to obtain a secondary component of d-axis current of the three-phase rectifier:
wherein i is d2 (t) is the secondary component of the d-axis current of the three-phase rectifier; l (L) s The filter inductance is input;
s306, carrying out phase shift and amplitude change on a secondary component of d-axis current of the three-phase rectifier to obtain secondary wave quantity;
s307, obtaining d-axis voltage modulation signals of the three-phase rectifier according to the secondary fluctuation and d-axis voltage direct current under the DQ coordinate system of the three-phase rectifier:
u d (t)＝u d0 +u d2 cos(2ωt+β)
Wherein u is d (t) is a d-axis voltage modulation signal of a three-phase rectifier; u (u) d0 D-axis voltage direct current under a DQ coordinate system of the three-phase rectifier; u (u) d2 cos (2ωt+β) is the amount of the secondary wave;
and S308, obtaining a characteristic sub-current harmonic suppression signal according to the q-axis voltage modulation signal of the three-phase rectifier and the d-axis voltage modulation signal of the three-phase rectifier.
The beneficial effects of the above-mentioned further scheme are: the three-phase current harmonic component is extracted, and the rectifier is controlled to generate opposite current harmonic, so that a characteristic sub-current harmonic suppression signal is obtained, and harmonic components in the three-phase current are suppressed.
Further, the step S6 specifically includes:
s601, acquiring single-phase inversion output voltage, amplitude of second harmonic injected by d-axis of the single-phase inverter and phase of the second harmonic injected by d-axis of the single-phase inverter:
wherein u is md2 Amplitude of second harmonic injected for d-axis of single-phase inverter; gamma is the phase of the second harmonic injected by the d axis of the single-phase inverter; omega is the frequency; m is the modulation degree of the inverter;
s602, performing PARK coordinate transformation on third harmonic waves of single-phase inversion output voltage to obtain d-axis voltage secondary components of the single-phase inverter;
s603, obtaining a second harmonic modulation signal of the single-phase inverter according to the d-axis voltage secondary component of the single-phase inverter, the amplitude of the second harmonic injected by the d-axis of the single-phase inverter and the phase of the second harmonic injected by the d-axis of the single-phase inverter;
S604, performing voltage-current double-loop control by using a low-pass filter according to the single-phase inversion output voltage to obtain a q-axis voltage modulation signal and a first d-axis voltage modulation signal of the single-phase inverter;
s605, obtaining a d-axis voltage modulation signal of the single-phase inverter according to the first d-axis voltage modulation signal and a second harmonic modulation signal of the single-phase inverter;
s606, obtaining an inverter modulation wave according to a d-axis voltage modulation signal of the single-phase inverter and a q-axis voltage modulation signal of the single-phase inverter, controlling the single-phase inverter based on the inverter modulation wave, and completing coordination control of the three-phase-single-phase multi-level converter, wherein the expression of the inverter modulation wave is as follows:
u mo (t)＝u md cosωt+u mq sinωt+u mc
u mc (t)＝u md2 cos(3ωt+γ)/2+u md2 cos(ωt+γ)/2
wherein u is mo (t) is an inverter modulation wave; u (u) md Modulating a signal for a first d-axis voltage; u (u) mq A q-axis voltage modulation signal of the single-phase inverter; u (u) mc Modulating signals for the second harmonic of single-phase invertersThe method comprises the steps of carrying out a first treatment on the surface of the t is time; ωt is the output side phase.
The beneficial effects of the above-mentioned further scheme are: by injecting corresponding second harmonic modulation signals into d-axis voltage modulation signals of the single-phase inverter, third harmonic and single harmonic modulation waves with the same amplitude can be obtained, inverter modulation waves are obtained, and single-phase inversion output voltage harmonics are restrained.
Drawings
FIG. 1 is a flow chart of the method of the present invention.
Fig. 2 is a typical three-phase-single-phase three-level PWM converter circuit.
Fig. 3 is a block diagram of a three-phase rectifier d-q decoupling control.
Fig. 4 is a d-axis voltage-current dual closed-loop control system structure of a three-phase rectifier.
Fig. 5 is a block diagram of a single-phase inverter d-q decoupling control.
Fig. 6 is a schematic diagram of a three-phase-single-phase three-level converter structure of a conventional active power decoupling circuit and an inductance voltage equalizing circuit.
Fig. 7 is a schematic diagram of a three-phase-single-phase three-level converter based on a symmetrical half-bridge decoupling circuit.
Fig. 8 is a schematic diagram of an improved three-phase-single-phase three-level converter based on an extensible symmetrical half-bridge decoupling circuit according to the present invention.
Fig. 9 shows a comprehensive coordination control strategy of a three-phase-single-phase converter according to the present invention.
Fig. 10 shows a coordinated control strategy of characteristic subharmonic suppression and secondary ripple suppression of a three-phase rectifier according to the present invention.
Fig. 11 shows a characteristic subharmonic suppression strategy for a single-phase inverter according to the present invention.
Fig. 12 is a schematic diagram of a typical three-phase-single-phase five-level converter configuration.
Fig. 13 is a schematic diagram of a three-phase-single-phase five-level converter based on an extensible symmetrical half-bridge decoupling circuit according to the present invention.
Detailed Description
The following description of the embodiments of the present invention is provided to facilitate understanding of the present invention by those skilled in the art, but it should be understood that the present invention is not limited to the scope of the embodiments, and all the inventions which make use of the inventive concept are protected by the spirit and scope of the present invention as defined and defined in the appended claims to those skilled in the art.
Example 1
As shown in fig. 1, in one embodiment of the present invention, a coordinated control method of a three-phase-single-phase multi-level converter includes the steps of:
s1, connecting a three-phase rectifier and a single-phase inverter through an extensible symmetrical half-bridge decoupling circuit to obtain a three-phase-single-phase converter;
s2, performing secondary ripple suppression of the three-phase rectifier according to the three-phase-single-phase converter to obtain a secondary ripple suppression signal;
s3, carrying out characteristic sub-current harmonic suppression of the three-phase rectifier according to the three-phase-single-phase converter to obtain a characteristic sub-current harmonic suppression signal;
s4, adding the secondary ripple suppression signal and the characteristic secondary current harmonic suppression signal to obtain a third bridge arm modulation wave;
S5, controlling a third bridge arm of the three-phase rectifier by utilizing the third bridge arm modulation wave;
s6, according to the three-phase-single-phase converter, characteristic subvoltage harmonic suppression of the single-phase inverter is carried out, inverter modulation waves are obtained, the single-phase inverter is controlled based on the inverter modulation waves, and coordination control of the three-phase-single-phase multi-level converter is completed.
In this embodiment, aiming at the problems that the three-phase-single-phase multilevel converter has unbalance of direct-current side voltage and secondary ripple and harmonic exists between three-phase input current and output voltage of the single-phase inverter, the invention provides a coordination control method of the three-phase-single-phase multilevel converter, which can realize voltage balance and secondary ripple suppression on the three-phase-single-phase converter and ensure that the three-phase input current and the single-phase output voltage have good sine. The expandable symmetrical half-bridge APD circuit has expansibility and is suitable for three, five, seven and other multi-level structures, and the three-phase-single-phase converter consists of a three-phase rectifier of a front stage, a direct-current side supporting capacitor and a filter inductor in the middle and a single-phase inverter of a rear stage.
The technical scheme for solving the technical problems is as follows: the method comprises the steps that a characteristic secondary current harmonic suppression and direct current side secondary ripple suppression coordination control strategy is provided in a three-phase rectifier of a three-phase-single-phase converter, a symmetrical half-bridge type power decoupling circuit is formed by a third bridge arm of the three-phase rectifier, a direct current side supporting capacitor and a filter inductor, the secondary ripple suppression control strategy is provided based on the circuit, a secondary component of direct current voltage of a control loop is further suppressed by a trap, a mathematical relation between a network side current harmonic component of the three-phase rectifier and a d-axis modulation voltage expression is established, and the influence of the secondary component of direct current voltage on the current of the three-phase network side through port voltage is suppressed;
The method comprises the steps that a characteristic subvoltage harmonic suppression strategy is provided for a single-phase inverter of a three-phase-single-phase converter, a mathematical relation between an inverter output voltage harmonic component and a d-axis modulation voltage expression is established, the influence of a direct-current voltage secondary component on the single-phase inversion output voltage through a port voltage is suppressed, the single-phase inverter adopts an SVPWM modulation strategy, and upper and lower capacitor voltages are balanced by using a redundancy vector; the two controls together form a comprehensive coordination control strategy of the three-phase-single-phase three-level converter, so that secondary ripple waves on the direct current side can be eliminated, upper and lower capacitor voltage balance is ensured, and meanwhile, good sine property of three-phase input current and single-phase output voltage is ensured.
The scalable symmetrical half-bridge decoupling circuit in step S1 specifically comprises: for an n-level three-phase rectifier, the scalable symmetrical half-bridge decoupling circuit includes (n-1)/2 symmetrical half-bridge decoupling circuits; the symmetrical half-bridge decoupling circuits comprise two direct-current side supporting capacitors and a second direct-current side supporting capacitor, wherein the two direct-current side supporting capacitors and the second direct-current side supporting capacitor are arranged between the three-phase rectifier and the single-phase inverter; a filter inductor is arranged between the two direct-current side supporting capacitors; the filter inductor is connected with the three-phase rectifier.
The step S2 specifically comprises the following steps:
s201, obtaining a direct-current side secondary ripple voltage according to a three-phase-single-phase converter;
s202, obtaining a basic component in a static coordinate system according to the secondary ripple voltage of the direct current side;
s203, obtaining a secondary ripple suppression signal according to the basic components in the static coordinate system.
The step S201 specifically includes:
s2011, obtaining direct-current voltage and a given direct-current voltage value according to a three-phase-single-phase converter;
s2012, obtaining a DC voltage deviation value according to the DC voltage and a given DC voltage value;
s2013, obtaining the secondary ripple voltage of the direct current side by utilizing a double-frequency band-pass filter according to the direct current voltage deviation value.
The step S202 specifically includes:
s2021, obtaining output voltage according to the three-phase-single-phase converter;
s2022, obtaining an output side phase by using a single-phase-locked loop according to the output voltage;
s2023, shifting the phase of the secondary ripple voltage of the direct current side by 1/4 period, setting the delay time to be 2.5ms, and obtaining a virtual control variable;
s2024, obtaining a basic component in a static coordinate system by utilizing a transformation matrix according to the virtual control variable, the direct-current side secondary ripple voltage and the output side phase; the expression of the transformation matrix is as follows:
Wherein T is trans Is a transformation matrix; ωt is the output side phase; θ is the phase angle of the filtered inductor current.
The step S203 specifically includes:
s2031, obtaining inductance reference current according to basic components in a static coordinate system;
s2032, obtaining inductance current according to the three-phase-single-phase converter;
s2033, obtaining inductance current deviation according to the inductance current and the inductance reference current;
and S2034, obtaining a secondary ripple suppression signal by using the PR controller according to the inductance current deviation.
The step S3 specifically comprises the following steps:
s301, according to the DC voltage deviation value, a band-stop filter and a PI controller are utilized to inhibit voltage ripple, and a d-axis current reference value is obtained:
wherein i is sd * Is a d-axis current reference value;the d-axis reference current direct current is used as the reference current direct current;
s302, performing a three-phase rectifier DQ decoupling control strategy according to the three-phase-single-phase converter and the d-axis current reference value to obtain a three-phase rectifier q-axis voltage modulation signal and d-axis voltage direct current under a three-phase rectifier DQ coordinate system;
s303, acquiring the secondary fluctuation amplitude value of the d-axis modulation signal of the three-phase rectifier and the secondary fluctuation phase of the d-axis modulation signal of the three-phase rectifier:
wherein u is d2 The amplitude of the secondary wave quantity of the d-axis modulation signal of the three-phase rectifier is obtained; beta is the phase of the secondary fluctuation of the d-axis modulation signal of the three-phase rectifier; m is m s The modulation degree of the rectifier;a direct current amount which is a direct current side voltage; c is the equivalent capacitance value of the direct current side; phi (phi) o The phase angle of the output current and the voltage is inverted; alpha is the phase angle of the inverted output voltage; u (U) o Is the output voltage amplitude; i o Is the output current amplitude; omega s The voltage frequency is three-phase network side voltage frequency;
s304, obtaining a three-phase power frequency negative sequence modulation signal and a three-phase positive sequence modulation signal according to the secondary wave amplitude value of the d-axis modulation signal of the three-phase rectifier and the secondary wave phase of the d-axis modulation signal of the three-phase rectifier:
wherein u is a- * (t) is the single negative sequence component of the a-phase voltage modulated signal; u (u) b- * (t) is the single negative sequence component of the b-phase voltage modulated signal; u (u) c- * (t) is the single negative sequence component of the c-phase voltage modulated signal; u (u) a3 * (t) is the positive three-order component of the a-phase voltage modulation signal; u (u) b3 * (t) is the b-phase voltage modulation signal cubic positive sequence component; u (u) c3 * (t) is the c-phase voltage modulation signal cubic positive sequence component; omega is the frequency; t is time; ωt is the output side phase;
s305, performing DQ conversion according to the three-phase power frequency negative sequence modulation signal and the three-phase positive sequence modulation signal to obtain a secondary component of d-axis current of the three-phase rectifier:
wherein i is d2 (t) is the secondary component of the d-axis current of the three-phase rectifier; l (L) s The filter inductance is input;
s306, carrying out phase shift and amplitude change on a secondary component of d-axis current of the three-phase rectifier to obtain secondary wave quantity;
s307, obtaining d-axis voltage modulation signals of the three-phase rectifier according to the secondary fluctuation and d-axis voltage direct current under the DQ coordinate system of the three-phase rectifier:
u d (t)＝u d0 +u d2 cos(2ωt+β)
wherein u is d (t) is a d-axis voltage modulation signal of a three-phase rectifier; u (u) d0 D-axis voltage direct current under a DQ coordinate system of the three-phase rectifier; u (u) d2 cos (2ωt+β) is the amount of the secondary wave;
and S308, obtaining a characteristic sub-current harmonic suppression signal according to the q-axis voltage modulation signal of the three-phase rectifier and the d-axis voltage modulation signal of the three-phase rectifier.
The step S6 specifically includes:
s601, acquiring single-phase inversion output voltage, amplitude of second harmonic injected by d-axis of the single-phase inverter and phase of the second harmonic injected by d-axis of the single-phase inverter:
wherein u is md2 Amplitude of second harmonic injected for d-axis of single-phase inverter; gamma is the phase of the second harmonic injected by the d axis of the single-phase inverter; omega is the frequency; m is the modulation degree of the inverter;
s602, performing PARK coordinate transformation on third harmonic waves of single-phase inversion output voltage to obtain d-axis voltage secondary components of the single-phase inverter;
s603, obtaining a second harmonic modulation signal of the single-phase inverter according to the d-axis voltage secondary component of the single-phase inverter, the amplitude of the second harmonic injected by the d-axis of the single-phase inverter and the phase of the second harmonic injected by the d-axis of the single-phase inverter;
S604, performing voltage-current double-loop control by using a low-pass filter according to the single-phase inversion output voltage to obtain a q-axis voltage modulation signal and a first d-axis voltage modulation signal of the single-phase inverter;
s605, obtaining a d-axis voltage modulation signal of the single-phase inverter according to the first d-axis voltage modulation signal and a second harmonic modulation signal of the single-phase inverter;
s606, obtaining an inverter modulation wave according to a d-axis voltage modulation signal of the single-phase inverter and a q-axis voltage modulation signal of the single-phase inverter, controlling the single-phase inverter based on the inverter modulation wave, and completing coordination control of the three-phase-single-phase multi-level converter, wherein the expression of the inverter modulation wave is as follows:
u mo (t)＝u md cosωt+u mq sinωt+u mc
u mc (t)＝u md2 cos(3ωt+γ)/2+u md2 cos(ωt+γ)/2
wherein u is mo (t) is an inverter modulation wave; u (u) md Modulating a signal for a first d-axis voltage; u (u) mq A q-axis voltage modulation signal of the single-phase inverter; u (u) mc The second harmonic modulation signal is a single-phase inverter; t is time; ωt is the output side phase.
Example 2
Fig. 2 shows a three-phase-single-phase three-level PWM converter circuit, in which the output side of the three-phase-single-phase converter is a single-phase inverter, and the current of the single-phase converter on the ac side can be controlled to be sinusoidal current with the same frequency as the voltage no matter in the rectifying or inverting state, so that the conversion system has a very high power factor.
However, sinusoidal fluctuating currents and voltages generate pulsating power that is doubled to the grid voltage frequency, which in turn generates secondary voltage ripple on the dc side.
Neglecting harmonic distortion influence of input voltage and current of the three-phase network side, and assuming that the expression of the input voltage of the three-phase network side is as follows:
wherein u is a (t) is an a-phase input voltage; u (u) b (t) is a b-phase input voltage; u (u) c (t) is the c-phase input voltage; u (U) s Is the input voltage amplitude; t is time; omega s Is the three-phase network side voltage frequency.
The expression of the three-phase input current is:
i a (t)＝I s cos(ω s t+φ s )
wherein i is a (t) input current for phase a; i.e b (t) is a b-phase input current; i.e c (t) is the c-phase input current; i s Is input voltage and current; phi (phi) s Is the phase angle of the net side current to the voltage.
Similarly, the expression of the single-phase inversion output voltage and current is assumed to be:
wherein u is o (t) is the output voltage; i.e o (t) is the output current; u (U) o Is the output voltage amplitude; i o Is the output current amplitude; omega o Is the output voltage frequency; alpha is the phase angle of the inverted output voltage; phi (phi) o To invert the phase angle of the output current to the voltage.
The three-phase network side adopts unit power control, the power factor is 1, no reactive power exists, and the three-phase network side input power expression can be expressed as:
Wherein p is in (t) three-phase network side input power;
instantaneous power p on input inductance Ls The expression (t) is:
wherein p is Ls (t) is the instantaneous power on the input inductor; l (L) s The filter inductance is input; d is a differential sign;
three-phase of the preceding stage of a three-phase-to-single-phase converterRectifier input power p s The expression (t) is:
wherein p is s (t) is the three-phase rectifier input power of the three-phase-to-single-phase converter front stage;
in a three-phase single-phase converter based on a flexible traction power supply system, the frequency of a three-phase rectification input voltage is equal to that of a single-phase inversion output voltage, and the single-phase inversion output instantaneous output power expression p is given o (t) is:
wherein p is o (t) is a single-phase inversion output instantaneous output power expression;
the difference value between the three-phase network side input power and the single-phase inversion output power is the power on the middle direct current side supporting capacitor:
wherein p is dc (t) is the power on the dc side supporting capacitor;
since the two direct current side supporting capacitors of the middle direct current side do not consume active power, the active power between the input and the output is equal, namely:
the power expression on the direct current side supporting capacitor can be simplified as:
the power on the dc side supporting capacitor can also be expressed as:
substituting this formula into the above formula yields:
Wherein u is dc (t) is a dc side voltage; c is the equivalent capacitance value of the direct current side;
the two-sided integral can be obtained:
where λ is a constant representing the dc component of the dc voltage. Can be solved as follows:
wherein, the liquid crystal display device comprises a liquid crystal display device,a direct current amount which is a direct current side voltage;
it can be seen from the above that the input power and the output power of the three-phase-single-phase converter based on the flexible traction power supply system are unbalanced, so that the intermediate direct-current voltage consists of a constant direct-current voltage and a secondary ripple wave twice the frequency of the network side voltage, and the magnitude of the secondary ripple wave is directly proportional to the transmission power of the converter and inversely proportional to the magnitude of the direct-current voltage and the magnitude of the direct-current capacitor.
The dq decoupling control of the three-phase rectifier of the front stage is shown in fig. 3, the d-axis control active power component, the q-axis control reactive power component, the q-axis current component is controlled to be 0, the q-axis component is controlled to be independent of the voltage of the direct current side and is not affected by secondary ripple, the q-axis component is controlled to enable the power factor of the three-phase network side to be 1, the d-axis component is controlled to enable the output voltage of the direct current side to be stable, the d-axis voltage and current double closed loop control system structure of the three-phase rectifier is shown in fig. 4, and the d-axis voltage and current double closed loop control system structure of the three-phase rectifier participates in the synthesis and decomposition of the three-phase current.
The purpose of the voltage outer ring in the double closed loop control strategy is to control the stable value of the output direct current side voltage, ensure no error when the direct current side output voltage is stable, and because the PI controller can control the direct current, the voltage outer ring adopts the PI controller, and the expression is as follows:
Wherein G is vd (s) is the rectifier voltage sampling delay; s is the variable sign of Law transformation; k (K) pl And K is equal to il The P and I coefficients of the PI controller of the rectifier voltage loop respectively, the calculation delay and the sampling delay exist in the digital circuit of the actual converter, and G vd (s) is the rectifier voltage sampling delay, which can be equivalently a computation delay comprising 1 sampling period, expressed as:
wherein G is vc (s) is a computation delay comprising 1 sampling period; t (T) s Is a sampling time;
G PWM (s) is a rectifier PWM modulation function, equivalent to a delay of PWM loading segment of 0.5 sample delay.
Wherein G is PWM (s) is a three-phase rectifier PWM modulation function; k (K) pwm The conversion ratio is PWM;
the power factor of the three-phase rectifier of the current inner loop action control front stage in the double closed loop control is 1, and DQ conversion is adoptedThe three-phase current in the natural coordinate system is converted into d-axis and q-axis current components in the synchronous rotation coordinate system, and the current obtained through DQ conversion is DC, so that a PI controller can be adopted by the current inner loop. G ic (s) is the rectifier current sampling delay, G id (s) is an expression of the PI controller of the rectifier, both expressions being in the form of the same sampling delay and PI controller of the voltage control loop. The calculation formula of the d-axis current reference value of the rectifier active current under the DQ coordinate system is as follows:
Wherein i is d * D-axis current reference value of rectifier active current under DQ coordinate system; u (U) dc * Is a direct current side voltage reference value;is the fluctuation component of the DC side voltage;
because the current inner loop adopts the PI controller to increase the system bandwidth and the system response speed, the steady-state precision is not strictly required, and therefore, the current inner loop can be simplified into a proportional controller for simplifying analysis. The PI controller can realize the no-difference tracking of the direct current, but has weaker control effect on the alternating current, so the PI controller can be simplified into a proportional controller when controlling the alternating current. The d-axis current reference value can be rewritten as:
wherein, the liquid crystal display device comprises a liquid crystal display device,is d-axis reference current direct current; omega is the frequency;
in the control of the rectifier, the phase-locked loop technology is adopted to realize the unit power factor operation, namely the network side current and the network side voltage are in phase, so that the reference value of the network side current under a natural coordinate system can be calculated as follows:
wherein i is a * A phase current reference value; i.e b * A b-phase current reference value; i.e c * A c-phase current reference value;harmonic component of a phase current reference value; />Harmonic component of b-phase current reference value; />Harmonic component of c-phase current reference value;
the three-phase current under the natural coordinate system can be expressed as two terms, the first term is a power frequency positive sequence three-phase current component and is an ideal value of a three-phase current reference value, the second term is a harmonic component of the three-phase current reference value, and expressions of the second term are respectively:
From the above analysis, assuming that the three-phase network side voltage and current are ideally free of harmonics, the intermediate dc side voltage of the converter fluctuates secondarily due to the unbalance of the three-phase network side and the inverter output side power as the inverter output is a single-phase system. The pulsating component enters the control system of the rectifier through the direct-current voltage control loop, not only the 3-order positive sequence harmonic component is introduced into the instruction current of the rectifier, so that the actual network side current also contains a large amount of third harmonic, but also the fundamental wave negative sequence current with the same amplitude as the third harmonic is injected. Similarly, the 3 rd harmonic component of the grid-side current will in turn introduce a frequency-doubled voltage ripple into the dc-side voltage along with the fundamental component of the grid-side voltage and into the control loop such that the reference current contains the 5 th positive sequence harmonic component and the 3 rd negative sequence harmonic component. Therefore, the network side current contains 3, 5, 7 positive sequence harmonic components and 1, 3, 5 negative sequence harmonic with the same amplitude, and the harmonic content is reduced along with the increase of the harmonic frequency. In addition, because the intermediate direct-current voltage has larger secondary ripple waves, harmonic waves can be generated on the port voltage of the converter, and the port voltage of the three-phase rectifier has the following expression:
Wherein u is an For inputting a phase a port voltage; u (u) bn For inputting b-phase port voltage; u (u) cn For inputting b-phase port voltage; u (u) dc For inputting a c-phase port voltage; s is S a The input a bridge arm switching function; s is S b The switching function of the bridge arm b is input; s is S c The switching function of the bridge arm is input c;
the three-phase rectifier circuit adopts SPWM modulation, and when the rectifier modulation degree m s When the impulse equivalent principle is less than or equal to 1, the switching function can be equivalent to sine wave according to the impulse equivalent principle and the SPWM symmetrical rule sampling method, and three switching functions of the three-phase rectifier can be obtained through a state space average method:
wherein m is s The modulation degree of the rectifier;
the three-phase rectifier port voltage can be obtained by a state space average method:
wherein, the liquid crystal display device comprises a liquid crystal display device,harmonic components of the input a-phase port voltage; />Harmonic components of the input b-phase port voltage; />A harmonic component that is the input c-phase port voltage;
the first term is the voltage component of the port of the power frequency positive sequence three-phase rectifier, which is the ideal value of the three-phase voltage reference value, the second term is the harmonic component of the three-phase voltage reference value, and the expressions of the second term are respectively:
after SPWM modulation, the secondary ripple wave introduces a third harmonic component and a single negative sequence harmonic component into the voltage of the rectifier port, and the larger the secondary ripple wave is, the larger the third harmonic voltage is. After the rectifier filters the inductance, a large amount of third harmonic and single negative sequence harmonic exist in the network side current.
As shown in fig. 5, since the control loop of the single-phase inverter at the rear stage does not include the dc side bus voltage, the secondary ripple voltage at the dc side does not generate harmonic wave to the single-phase inverter output voltage through the control loop, and only the harmonic wave is injected into the single-phase inverter output voltage through the hardware port, and the expression of the single-phase inverter output voltage is as follows:
wherein S is ao A bridge arm switch function is output; s is S bo The switching function of the bridge arm b is output;
the single-phase inverter also adopts a carrier laminated SPWM (sinusoidal pulse width modulation) strategy, when the modulation degree m of the inverter is less than or equal to 1, the switching function can be equivalent to a sine wave according to the impulse equivalent principle and the SPWM symmetrical rule sampling method, and the switching function of the single-phase inverter can be obtained through a state space average method:
wherein m is the modulation degree of the inverter;
the output port voltage of the single-phase inverter can be obtained by a state space average method:
after SPWM modulation, the secondary ripple wave introduces a third harmonic component into the port voltage of the single-phase inverter, the larger the secondary ripple wave is, the larger the third harmonic voltage is, and sine voltage with the same amplitude as the third harmonic is introduced, so that the amplitude and the phase of the output voltage are changed.
The three-level converter SPWM modulation algorithm can divide the voltage vector into 5 major classes, namely zero vector, positive small vector, negative small vector, medium vector and large vector. The 5 kinds of vectors have different effects on the midpoint potential, wherein when the vectors are zero, the three phases of the load are short-circuited, so that the midpoint voltage is not affected; when the vector is a large vector, the corresponding output end is not connected with a neutral point, but is connected with the positive bus and the negative bus respectively, so that the midpoint voltage is not influenced; when the small vector or the medium vector works, at least one output end is connected with the zero bus, and the zero bus and the positive electrode or the negative electrode of the direct current bus form a loop, so that the phenomenon of charging and discharging the direct current side supporting capacitor occurs, and the neutral point potential balance is influenced.
Fig. 6 shows a three-phase-single-phase three-level converter structure of an active power decoupling circuit and an inductance voltage equalizing circuit, which are symmetrical half-bridge circuits, wherein four full-control switching tubes, two capacitors and two inductors are added on the basis of two direct-current side supporting capacitors. The number of components used increases the volume and cost of the system.
The single-phase inverter adopts SVPWM modulation technology, realizes counteracting midpoint current by selecting reasonable redundant vectors, controls midpoint potential, replaces an inductance auxiliary voltage-sharing circuit by a software voltage-sharing method, adopts a main circuit after software voltage sharing as shown in figure 7, replaces the switching tube of a symmetrical half-bridge decoupling circuit by the switching tube of a third bridge arm of a rectifier, and adopts a main circuit as shown in figure 8. The control strategy consists of a coordination control strategy of characteristic sub-current harmonic suppression and secondary ripple suppression of the three-phase rectifier and a characteristic sub-voltage harmonic suppression strategy of the single-phase inverter.
In this embodiment, taking a three-level three-phase-single-phase converter based on an expandable symmetrical half-bridge decoupling circuit as an example, the direct current side realizes the active power decoupling function of the secondary power through the expandable symmetrical half-bridge decoupling circuit, and the half-bridge branches are modulated to make the direct current components of the two direct current side supporting capacitor voltages equal to half of the direct current link voltage, namely U dc And/2, but at the same time, a basic alternating current component is superimposed in each direct current side supporting capacitor voltage, and the phase shift is 180 degrees, so that the two alternating current component components are complementary.
Two DC side supporting capacitor voltages u c1 (t) and u c2 (t) is:
wherein u is c1 (t) is the upper capacitor voltage in the dc side supporting capacitor; u (u) c2 (t) is the lower capacitor voltage in the dc side supporting capacitor;
by differentiation, the upper and lower capacitance currents can be expressed as:
wherein i is c1 (t) is the upper capacitance current in the DC side supporting capacitance; i.e c2 (t) is the lower capacitance current in the dc side supporting capacitance;
under the condition of balancing the voltages of the upper capacitor and the lower capacitor, as the switching frequency of the three-phase rectification and the single-phase inversion is 1kHz, the average value of the midpoint current is zero in a short time, and under the condition that the midpoint input current and the single-phase inversion midpoint output current of the three-phase rectifier are not considered, the upper current of the filter inductor is as follows:
wherein i is f (t) is the filter inductor current; c (C) d Each capacitance value is the direct current side; u (U) c Supporting effective values of alternating current components of capacitor voltage for two direct current sides; θ is the phase angle of the alternating component of the two direct side support capacitor voltages;
the power of the two capacitors and the filter inductor on the symmetrical half-bridge decoupling circuit is as follows:
wherein p is f (t) is the power of two capacitors and a filter inductor on the symmetrical half-bridge decoupling circuit; l (L) f The filter inductor is a direct current side filter inductor; i.e f The inductor current is filtered;
by controlling the output side secondary power and the power on the APD capacitor inductance to cancel each other, the effective value and the phase angle of the alternating current component of the voltage of the two direct current side supporting capacitors can be deduced:
wherein L is g Filtering inductance for output; u (U) R To output a load voltage; r is an output load;
u calculated by the method c And theta theory calculated value is used for calculating induction current reference value i f * The control block diagram is shown in fig. 10, and the specific steps are as follows:
step one: for DC voltage u dc Sampling with a given DC voltage value u dc * Comparing to obtain output DC voltage deviation delta udc . The DC voltage deviation value is subjected to a frequency doubling band-pass filter to obtain a secondary ripple voltage u at the DC side dc2 。
Step two: collecting output voltage uo Obtaining the phase omega of the output side by a single-phase-locked loop o t. However, the extracted ripple voltage is second order, and in order to adjust such components, the following transformation matrix T is used trans The control variable may be made the primary component in the stationary coordinate system.
Transformation matrix T trans Is a virtual control variable obtained by virtually orthorhombic ripple voltage To get->Will be udc2 Phase shifted by a quarter cycle and set the delay time to 2.5ms.
Step three: multiplying the component under the static coordinate system obtained in S2 by a coefficient to obtain an inductance reference current i f * 。
And comparing the acquired inductive current with the obtained reference current to obtain the deviation amount of the inductive current, and controlling the deviation amount by adopting a PR controller.
And the PR controller output in the secondary ripple suppression loop is used as a modulation wave, the modulation wave is added with the modulation wave of the third bridge arm output by the three-phase rectifier control strategy, the PWM wave is finally generated to the third bridge arm of the three-phase rectifier, the two capacitance voltage alternating current components are controlled and controlled to be complementary, the secondary power is compensated, the secondary ripple on the direct current side is suppressed, and finally the three-phase network side current and the output voltage have good electric energy quality.
In order to further restrain the influence of the secondary ripple on the three-phase current, a frequency doubling trap can be added in the outer voltage loop, and the secondary ripple is filtered out in the control loop. However, the influence of secondary ripple and midpoint voltage fluctuation on the three-phase power grid through the port voltage cannot be solved. If the three-phase current harmonic components can be extracted, and the rectifier is controlled to generate opposite current harmonics, harmonic components in the three-phase current can be suppressed.
Because direct current side secondary ripple passes through port voltage and not only injects three positive sequence harmonic components into three-phase network side rectifier, still contains single negative sequence harmonic components moreover, only can't draw it out through the filter, can obtain after passing through DQ conversion with the third harmonic:
wherein i is 3d (t)，i 3q (t) are the current components of the third harmonic of the three-phase input current on the d-axis and q-axis, respectively, of the dq synchronous rotation coordinate system; from the above, it can be seen that the third harmonic is subjected to DQ conversion and then secondary ripple is injected into the d-axis and q-axis current components.
The DQ conversion of the single negative sequence harmonic component of the three-phase network side can be obtained:
wherein i is d- (t)，i q- And (t) are current components of the single negative sequence current of the three-phase input current on the d axis and the q axis on the DQ synchronous rotation coordinate system respectively, and it can be seen from the above that the single negative sequence harmonic wave of the three-phase input current is injected into the d axis and the q axis current components after DQ conversion.
The two formulas can be added to obtain:
it can be seen that after the third positive sequence harmonic and the single negative sequence harmonic with the same amplitude are added, the second wave momentum related to the d-axis current component is obtained after DQ decoupling transformation, and similarly, the third positive sequence harmonic and the single negative sequence harmonic with the same amplitude can be obtained after the d-axis second current component is subjected to inverse transformation. In the decoupling transformation of the three-phase rectifier DQ, d-axis secondary ripple can be extracted through specific secondary ripple extraction, and then certain harmonic waves are injected into modulation waves controlled by the three-phase rectifier according to the relation between secondary ripple current and three-phase network side current as well as DC voltage, so that the influence of DC side voltage ripple on three-phase input current is further suppressed.
According to analysis, the d-axis and q-axis modulation signals in DQ decoupling control need to contain a certain secondary component to inhibit the influence of the second harmonic of the DC side voltage on the three-phase network side current, and the d-axis and q-axis modulation signals are assumed to be:
wherein u is d (t) is a d-axis voltage modulation signal of a three-phase rectifier; u (u) q (t) is a three-phase rectifier q-axis voltage modulated signal; u (u) d0 D-axis voltage direct current under a DQ coordinate system of the three-phase rectifier; u (u) d2 cos (2ωt+β) is the amount of the secondary wave; u (u) d2 The amplitude of the secondary fluctuation of the d-axis voltage modulation signal is obtained; beta is the phase of the secondary fluctuation of the d-axis modulation signal; u (u) q0 Modulating the direct current of the signal for the q axis of the three-phase rectifier;
first item u d0 Under unit power factor control, q-axis current component u d0 =0, second term u d2 cos (2ωt+β) is a secondary fluctuation amount, and the modulation signal under the DQ axis is transformed to a coordinate system of a, b, and c, and the expression is:
wherein u is a * (t) is an a-phase voltage modulated signal; u (u) b * (t) is a b-phase voltage modulated signal; u (u) c * (t) is a c-phase voltage modulated signal; u (u) a1 * (t) is the positive sequence component of the power frequency of the a-phase voltage modulated signal; u (u) b1 * (t) is the positive sequence component of the power frequency of the b-phase voltage modulated signal; u (u) c1 * (t) is the power frequency positive sequence component of the c-phase voltage modulation signal; u (u) a- * (t) is the single negative sequence component of the a-phase voltage modulated signal; u (u) b- * (t) is the single negative sequence component of the b-phase voltage modulated signal; u (u) c- * (t) is the single negative sequence component of the c-phase voltage modulated signal; u (u) a3 * (t) is the positive three-order component of the a-phase voltage modulation signal; u (u) b3 * (t) is the b-phase voltage modulation signal cubic positive sequence component; u (u) c3 * (t) is the c-phase voltage modulation signal cubic positive sequence component;
a. the first term of the modulation signal under the b and c coordinate systems is converted into a three-phase power frequency positive sequence modulation signal, and the expression is as follows:
and (3) obtaining a three-phase power frequency negative sequence modulation signal and a three-phase positive sequence modulation signal by inverse transformation of the second secondary component, wherein the expression of the three-phase power frequency negative sequence modulation signal is as follows:
the expression of the three-phase three-time positive sequence modulation signal is as follows:
according to impulse equivalence principle, the expression of the three-phase network side port voltage can be obtained:
wherein u is an For inputting a phase a port voltage; u (u) bn For inputting b-phase port voltage; u (u) cn For inputting a c-phase port voltage;is the fluctuation component of the DC side voltage;
from the above, it can be seen that the port voltage component generated by using the dc component of the dc bus voltage and the single negative sequence harmonic wave and the third positive sequence harmonic wave counteracts the corresponding harmonic component, and because the ac component of the dc bus voltage and the d-axis injected modulated wave are small signals, the product is negligible, and therefore, the expression of the d-axis second harmonic amplitude and the phase can be obtained:
The expression of the three-phase current at the net side is:
wherein i is a Inputting current for phase a; i.e b Inputting current for phase b; i.e c Inputting current for phase c; u (u) a Input voltage for a phase; u (u) b Input voltage for phase b; u (u) c Input voltage for c phase; j is an imaginary symbol; l (L) s The filter inductance is input;
the three-phase network side current is subjected to coordinate transformation to obtain an expression under a DQ coordinate system, and the single negative sequence harmonic and the three positive sequence harmonic are subjected to DQ transformation to obtain a secondary component of the d-axis current of the three-phase rectifier according to the derivation of the previous voltage coordinate transformation, wherein the expression is as follows:
wherein i is d2 (t) is the secondary component of the d-axis current of the three-phase rectifier;
the secondary component of the d-axis current in the formula is subjected to phase shift and amplitude change, so that a d-axis voltage modulation signal to be injected can be obtained.
In this embodiment, the voltage harmonic is converted into the current harmonic after passing through the filter inductor, so that the characteristic sub-current harmonic suppression signal can be obtained according to the q-axis voltage modulation signal of the three-phase rectifier and the d-axis voltage modulation signal of the three-phase rectifier.
In summary, a DQ decoupling control strategy block diagram with specific sub-current harmonic suppression can be obtained by combining the pre-stage three-phase rectifier base DQ decoupling control strategy as shown in fig. 10.
Since the control loop of the single-phase inverter does not contain the DC bus voltage, the DC side secondary voltage ripple does not inject harmonic waves into the inversion output voltage from the control loop, and the DC side secondary voltage ripple and the midpoint voltage fluctuation inject three-time voltage ripple and single-time voltage harmonic waves into the single-phase output voltage through the port voltage.
Similar to the control strategy for suppressing harmonic wave influence of a three-phase rectifier, since the direct-current side secondary voltage ripple not only injects a third harmonic component into the single-phase inversion output voltage through the port voltage, but also contains a single harmonic component, the third harmonic component cannot be extracted completely through a filter, and the second harmonic modulation signal is transformed into a natural coordinate system by injecting a corresponding second harmonic into a d-axis modulation signal, the same amplitude third harmonic and single harmonic modulation wave can be obtained, and further, the single-phase inversion output voltage harmonic is suppressed, and the second harmonic expression of d-axis injection is assumed to be:
u md2 (t)＝u md2 cos(2ωt+γ)
wherein u is md2 (t) is the secondary of the single-phase inverter d-axis injectionHarmonics; u (u) md2 Amplitude of second harmonic injected for d-axis of single-phase inverter; gamma is the phase of the second harmonic injected by the d axis of the single-phase inverter;
The second harmonic of the d axis is transformed into a natural coordinate system, and the following can be obtained:
u mc (t)＝u md2 cos(3ωt+γ)/2+u md2 cos(ωt+γ)/2
wherein u is mc (t) is an expression of the second harmonic injected on the d axis in a natural coordinate system;
from the above, it can be seen that the second ripple of the d-axis is transformed into the natural coordinate system, and the third harmonic and the single harmonic are obtained, so that the modulation wave of the single-phase inverter can be obtained as follows:
u mo (t)＝u md cosωt+u mq sinωt+u mc
wherein u is mo (t) is an inverter modulation wave; u (u) md Modulating a signal for a first d-axis voltage; u (u) mq A q-axis voltage modulation signal of the single-phase inverter; u (u) mc The second harmonic modulation signal is a single-phase inverter;
according to impulse equivalence principle, the expression of the output port voltage of the single-phase inverter can be obtained:
wherein u is on (t) is the single phase inverter output port voltage;
from the above, it can be seen that, by using the direct current component of the direct current bus voltage and the port voltage component generated by the third harmonic and the single harmonic modulation wave to cancel the corresponding harmonic component, since the alternating current component of the direct current bus voltage and the d-axis injected modulation wave are small signals, the product is negligible, and therefore, by calculating the expression of the amplitude and the phase of the second harmonic to be introduced by the d-axis modulation wave:
the voltage modulation signal needing to be injected into the d axis is obtained through the third harmonic of the single-phase inversion output voltage, and the third harmonic is obtained after PARK coordinate transformation:
Wherein C is αβ/dq A transformation matrix from an alpha beta coordinate system to a dq coordinate system; u (u) q2 The amplitude of the q-axis second harmonic component; u (u) o3 The third harmonic amplitude of the output voltage; u (u) d2 ' is the amplitude of the secondary fluctuation of the d-axis voltage modulation signal of the inversion output control link;
and extracting the secondary component in the d-axis voltage after the transformation, and carrying out corresponding amplitude change on the secondary component to obtain the d-axis voltage modulation signal to be injected.
The problem of unbalanced midpoint potential is solved by adopting a switching state conversion method in the single-phase inverter. The core idea is as follows: the power flow direction of the single-phase inverter is judged first, and then the charge and discharge conditions of the upper capacitor and the lower capacitor at the direct current side are changed by switching state transition when appropriate.
From the specific switch state analysis, (S) a ,S b ) When (1, 0), (0, 1), the capacitance C 1 Charged by a forward current or discharged by a reverse current. (S) a ,S b ) In the case of (-1, 0), (0, -1), the capacitance C 2 Charged by a forward current or discharged by a reverse current. Therefore, these 4 switching patterns are the main cause of the shift in the midpoint potential.
The principle of software equalizing is shown in fig. 11. Wherein u is c1 、u c2 The voltages of the upper capacitor and the lower capacitor of the direct current side are respectively, u o Output instantaneous voltage for single-phase inverter, i o The instantaneous current is output for the single-phase inverter. Substituting it into the conversion formula to obtainThe switch state transition signal T. The switching state transition is completed through the transition process shown in fig. 11:
T＝(u c1 -u c2 )i o u o
and then the switching state conversion is completed through the conversion process:
based on a space vector pulse width modulation voltage equalizing algorithm, the switching state is generated after the space vector pulse width modulation is performed. And judging the current power flow direction, and converting the existing switch state to weaken the offset of the neutral point.
For example, when u c1 >u c2 ，u s >0，i s >At 0, T>0, the upper capacitor C is required for balancing the midpoint potential 1 Discharging or aligning the lower capacitor C 2 And (5) discharging. Therefore T>At 0, if it occurs (S a ,S b ) In the case of (1, 0), it should be converted to (0, -1), i.e., the upper capacitor charge is adjusted to the operating state for charging the lower capacitor. Thereby bringing the midpoint potential toward equilibrium. The conversion process of the other three cases can be analyzed in a similar way. After the switching state conversion, the balance of the midpoint potential can be realized, the output voltage at the direct current side still can be stabilized near the given value, the phase of the current at the net side is the same as that of the voltage, and the voltage at the net side U ab And still be a sine wave equivalent to a five level signal. Waveforms other than the capacitor voltage are balanced are not affected. To prevent the switching frequency from being too high or even unstable caused by too frequent switching of the switching state, u can be set c1 -u c2 The conversion is performed only when the value of (c) is greater than a positive constant.
In summary, by combining the control strategy of the single-phase inverter of the subsequent stage, a DQ decoupling control strategy block diagram with specific subvoltage harmonic suppression can be obtained as shown in fig. 11:
the comprehensive coordination control strategy of the three-phase-single-phase converter is also applicable to five-level and seven-level multi-level circuits, fig. 12 is a typical three-phase-single-phase five-level PWM converter circuit, the symmetrical half-bridge APD is applied to the five-level circuit as shown in fig. 13, the working principle of two symmetrical half-bridge APDs is the same as that of a three-level converter, and the same applies to the working principle and control of more level converters.
Claims (8)
1. The coordination control method of the three-phase-single-phase multi-level converter is characterized by comprising the following steps of:
s1, connecting a three-phase rectifier and a single-phase inverter through an extensible symmetrical half-bridge decoupling circuit to obtain a three-phase-single-phase converter;
s2, performing secondary ripple suppression of the three-phase rectifier according to the three-phase-single-phase converter to obtain a secondary ripple suppression signal;
s3, carrying out characteristic sub-current harmonic suppression of the three-phase rectifier according to the three-phase-single-phase converter to obtain a characteristic sub-current harmonic suppression signal;
S4, adding the secondary ripple suppression signal and the characteristic secondary current harmonic suppression signal to obtain a third bridge arm modulation wave;
s5, controlling a third bridge arm of the three-phase rectifier by utilizing the third bridge arm modulation wave;
s6, according to the three-phase-single-phase converter, characteristic subvoltage harmonic suppression of the single-phase inverter is carried out, inverter modulation waves are obtained, the single-phase inverter is controlled based on the inverter modulation waves, and coordination control of the three-phase-single-phase multi-level converter is completed.
2. The method for coordinated control of a three-phase-single-phase multilevel converter according to claim 1, wherein the scalable symmetrical half-bridge decoupling circuit in step S1 is specifically: for an n-level three-phase rectifier, the scalable symmetrical half-bridge decoupling circuit includes (n-1)/2 symmetrical half-bridge decoupling circuits; the symmetrical half-bridge decoupling circuits comprise two direct-current side supporting capacitors and a second direct-current side supporting capacitor, wherein the two direct-current side supporting capacitors and the second direct-current side supporting capacitor are arranged between the three-phase rectifier and the single-phase inverter; a filter inductor is arranged between the two direct-current side supporting capacitors; the filter inductor is connected with the three-phase rectifier.
3. The method for coordinated control of a three-phase-single-phase multilevel converter according to claim 1, wherein the step S2 is specifically:
S201, obtaining a direct-current side secondary ripple voltage according to a three-phase-single-phase converter;
s202, obtaining a basic component in a static coordinate system according to the secondary ripple voltage of the direct current side;
s203, obtaining a secondary ripple suppression signal according to the basic components in the static coordinate system.
4. The method for coordinated control of a three-phase-single-phase multilevel converter according to claim 3, wherein the step S201 specifically comprises:
s2011, obtaining direct-current voltage and a given direct-current voltage value according to a three-phase-single-phase converter;
s2012, obtaining a DC voltage deviation value according to the DC voltage and a given DC voltage value;
s2013, obtaining the secondary ripple voltage of the direct current side by utilizing a double-frequency band-pass filter according to the direct current voltage deviation value.
5. The method for coordinated control of a three-phase-single-phase multilevel converter according to claim 3, wherein the step S202 is specifically:
s2021, obtaining output voltage according to the three-phase-single-phase converter;
s2022, obtaining an output side phase by using a single-phase-locked loop according to the output voltage;
s2023, shifting the phase of the secondary ripple voltage of the direct current side by 1/4 period, setting the delay time to be 2.5ms, and obtaining a virtual control variable;
S2024, obtaining a basic component in a static coordinate system by utilizing a transformation matrix according to the virtual control variable, the direct-current side secondary ripple voltage and the output side phase; the expression of the transformation matrix is as follows:
wherein T is trans Is a transformation matrix; ωt is the output side phase; θ is the phase angle of the filter inductance current; omega is the frequency; t is time.
6. The method for coordinated control of a three-phase-single-phase multilevel converter according to claim 3, wherein the step S203 is specifically:
s2031, obtaining inductance reference current according to basic components in a static coordinate system;
s2032, obtaining inductance current according to the three-phase-single-phase converter;
s2033, obtaining inductance current deviation according to the inductance current and the inductance reference current;
and S2034, obtaining a secondary ripple suppression signal by using the PR controller according to the inductance current deviation.
7. The method for coordinated control of a three-phase-single-phase multilevel converter according to claim 4, wherein step S3 is specifically:
s301, according to the DC voltage deviation value, a band-stop filter and a PI controller are utilized to inhibit voltage ripple, and a d-axis current reference value is obtained:
wherein i is sd * Is a d-axis current reference value; The d-axis reference current direct current is used as the reference current direct current;
s302, performing a three-phase rectifier DQ decoupling control strategy according to the three-phase-single-phase converter and the d-axis current reference value to obtain a three-phase rectifier q-axis voltage modulation signal and d-axis voltage direct current under a three-phase rectifier DQ coordinate system;
s303, acquiring the secondary fluctuation amplitude value of the d-axis modulation signal of the three-phase rectifier and the secondary fluctuation phase of the d-axis modulation signal of the three-phase rectifier:
wherein u is d2 The amplitude of the secondary wave quantity of the d-axis modulation signal of the three-phase rectifier is obtained; beta is the phase of the secondary fluctuation of the d-axis modulation signal of the three-phase rectifier; m is m s The modulation degree of the rectifier;a direct current amount which is a direct current side voltage; c is the equivalent capacitance value of the direct current side; phi (phi) o The phase angle of the output current and the voltage is inverted; alpha is the phase angle of the inverted output voltage; u (U) o Is the output voltage amplitude; i o Is the output current amplitude; omega s The voltage frequency is three-phase network side voltage frequency;
s304, obtaining a three-phase power frequency negative sequence modulation signal and a three-phase positive sequence modulation signal according to the secondary wave amplitude value of the d-axis modulation signal of the three-phase rectifier and the secondary wave phase of the d-axis modulation signal of the three-phase rectifier:
wherein u is a- * (t) is the single negative sequence component of the a-phase voltage modulated signal; u (u) b- * (t) is the single negative sequence component of the b-phase voltage modulated signal; u (u) c- * (t) is the single negative sequence component of the c-phase voltage modulated signal; u (u) a3 * (t) is the positive three-order component of the a-phase voltage modulation signal; u (u) b3 * (t) is the b-phase voltage modulation signal cubic positive sequence component; u (u) c3 * (t) modulation of c-phase voltageA signal three-time positive sequence component; omega is the frequency; t is time; ωt is the output side phase;
s305, performing DQ conversion according to the three-phase power frequency negative sequence modulation signal and the three-phase positive sequence modulation signal to obtain a secondary component of d-axis current of the three-phase rectifier:
wherein i is d2 (t) is the secondary component of the d-axis current of the three-phase rectifier; l (L) s The filter inductance is input;
s306, carrying out phase shift and amplitude change on a secondary component of d-axis current of the three-phase rectifier to obtain secondary wave quantity;
s307, obtaining d-axis voltage modulation signals of the three-phase rectifier according to the secondary fluctuation and d-axis voltage direct current under the DQ coordinate system of the three-phase rectifier:
u d (t)＝u d0 +u d2 cos(2ωt+β)
wherein u is d (t) is a d-axis voltage modulation signal of a three-phase rectifier; u (u) d0 D-axis voltage direct current under a DQ coordinate system of the three-phase rectifier; u (u) d2 cos (2ωt+β) is the amount of the secondary wave;
and S308, obtaining a characteristic sub-current harmonic suppression signal according to the q-axis voltage modulation signal of the three-phase rectifier and the d-axis voltage modulation signal of the three-phase rectifier.
8. The method for coordinated control of a three-phase-single-phase multilevel converter according to claim 1, wherein the step S6 is specifically:
s601, acquiring single-phase inversion output voltage, amplitude of second harmonic injected by d-axis of the single-phase inverter and phase of the second harmonic injected by d-axis of the single-phase inverter:
wherein u is md2 Amplitude of second harmonic injected for d-axis of single-phase inverter; gamma is the phase of the second harmonic injected by the d axis of the single-phase inverter; omega is the frequency; m is the modulation degree of the inverter;
s602, performing PARK coordinate transformation on third harmonic waves of single-phase inversion output voltage to obtain d-axis voltage secondary components of the single-phase inverter;
s603, obtaining a second harmonic modulation signal of the single-phase inverter according to the d-axis voltage secondary component of the single-phase inverter, the amplitude of the second harmonic injected by the d-axis of the single-phase inverter and the phase of the second harmonic injected by the d-axis of the single-phase inverter;
s604, performing voltage-current double-loop control by using a low-pass filter according to the single-phase inversion output voltage to obtain a q-axis voltage modulation signal and a first d-axis voltage modulation signal of the single-phase inverter;
s605, obtaining a d-axis voltage modulation signal of the single-phase inverter according to the first d-axis voltage modulation signal and a second harmonic modulation signal of the single-phase inverter;
S606, obtaining an inverter modulation wave according to a d-axis voltage modulation signal of the single-phase inverter and a q-axis voltage modulation signal of the single-phase inverter, controlling the single-phase inverter based on the inverter modulation wave, and completing coordination control of the three-phase-single-phase multi-level converter, wherein the expression of the inverter modulation wave is as follows:
u mo (t)＝u md cosωt+u mq sinωt+u mc
u mc (t)＝u md2 cos(3ωt+γ)/2+u md2 cos(ωt+γ)/2
wherein u is mo (t) is an inverter modulation wave; u (u) md Modulating a signal for a first d-axis voltage; u (u) mq A q-axis voltage modulation signal of the single-phase inverter; u (u) mc The second harmonic modulation signal is a single-phase inverter; t is time; ωt is the output side phase.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202310714906.9A CN116582006B (en) | 2023-06-15 | 2023-06-15 | Coordination control method for three-phase-single-phase multi-level converter |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202310714906.9A CN116582006B (en) | 2023-06-15 | 2023-06-15 | Coordination control method for three-phase-single-phase multi-level converter |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN116582006A true CN116582006A (en) | 2023-08-11 |
| CN116582006B CN116582006B (en) | 2024-03-12 |
Family
ID=87534189
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN202310714906.9A Active CN116582006B (en) | 2023-06-15 | 2023-06-15 | Coordination control method for three-phase-single-phase multi-level converter |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN116582006B (en) |
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN117394708A (en) * | 2023-12-13 | 2024-01-12 | 四川大学 | Current-mode PWM rectifier control system and method suitable for input unbalance |
Citations (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN102916572A (en) * | 2012-06-12 | 2013-02-06 | 华中科技大学 | Control method and system for inhibiting secondary ripple current and improving dynamic characteristic |
| CN106787671A (en) * | 2016-11-22 | 2017-05-31 | 张欣 | Suppress the circuit of power factor correction of the no electrolytic capacitor of function and fast dynamic response speed with secondary ripple wave |
| US20190190276A1 (en) * | 2017-12-15 | 2019-06-20 | Delta Electronics (Shanghai) Co., Ltd | Method and device for controlling distribution of unbalanced and harmonic power among parallel inverters |
| CN111490685A (en) * | 2020-04-14 | 2020-08-04 | 湖南工业大学 | Decoupling vector modulation method for three-phase high-frequency chain matrix converter |
| CN112290567A (en) * | 2020-12-23 | 2021-01-29 | 西南交通大学 | Three-phase power quality compensation device and method based on half-bridge converter |
| CN115313346A (en) * | 2022-08-19 | 2022-11-08 | 福州大学 | Low-frequency harmonic suppression method for load input current of single-phase voltage type inverter rectifier |
-
2023
- 2023-06-15 CN CN202310714906.9A patent/CN116582006B/en active Active
Patent Citations (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN102916572A (en) * | 2012-06-12 | 2013-02-06 | 华中科技大学 | Control method and system for inhibiting secondary ripple current and improving dynamic characteristic |
| CN106787671A (en) * | 2016-11-22 | 2017-05-31 | 张欣 | Suppress the circuit of power factor correction of the no electrolytic capacitor of function and fast dynamic response speed with secondary ripple wave |
| US20190190276A1 (en) * | 2017-12-15 | 2019-06-20 | Delta Electronics (Shanghai) Co., Ltd | Method and device for controlling distribution of unbalanced and harmonic power among parallel inverters |
| CN111490685A (en) * | 2020-04-14 | 2020-08-04 | 湖南工业大学 | Decoupling vector modulation method for three-phase high-frequency chain matrix converter |
| CN112290567A (en) * | 2020-12-23 | 2021-01-29 | 西南交通大学 | Three-phase power quality compensation device and method based on half-bridge converter |
| CN115313346A (en) * | 2022-08-19 | 2022-11-08 | 福州大学 | Low-frequency harmonic suppression method for load input current of single-phase voltage type inverter rectifier |
Non-Patent Citations (1)
| Title |
|---|
| 何晓琼等: "基于级联－并联变换器的贯通式牵引变电所系统研究", 铁道学报, vol. 39, no. 7, pages 52 - 61 * |
Cited By (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN117394708A (en) * | 2023-12-13 | 2024-01-12 | 四川大学 | Current-mode PWM rectifier control system and method suitable for input unbalance |
| CN117394708B (en) * | 2023-12-13 | 2024-02-20 | 四川大学 | Current-mode PWM rectifier control system and method suitable for input unbalance |
Also Published As
| Publication number | Publication date |
|---|---|
| CN116582006B (en) | 2024-03-12 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| CN106100402B (en) | A kind of T-type three-level inverter and its neutral balance control method | |
| CN110707958B (en) | Modulation wave interval division-based midpoint voltage control method | |
| CN104578884B (en) | A kind of low-voltage micro-capacitance sensor multi-inverter parallel Voltage unbalance control method | |
| CN110739678B (en) | Control method for series virtual impedance of grid-connected converter | |
| CN109713925B (en) | DC side voltage-sharing control method of three-phase four-wire three-level inverter | |
| CN109347354B (en) | Midpoint voltage ripple suppression device and method based on third harmonic injection | |
| CN116582006B (en) | Coordination control method for three-phase-single-phase multi-level converter | |
| CN111478565B (en) | Design method of higher harmonic suppression controller of VIENNA rectifier | |
| CN114583702A (en) | Parallel current control system and control method of LCL type active power filter | |
| CN101304172B (en) | Method for real time pre-compensating harmonic wave field dead region | |
| CN107046288B (en) | Structure of hybrid harmonic suppressor and control method thereof | |
| He et al. | Advanced traction power supply system based on modular multilevel converters | |
| CN112737383A (en) | Three-level NPC inverter carrier frequency ordered change inverted phase laminated pulse width modulation method | |
| CN116722750A (en) | Single-phase-three-phase converter and comprehensive coordination control method thereof | |
| CN204290329U (en) | A kind of SVG DC bus capacitor neutral-point potential balance device | |
| Huang et al. | Neutral-point potential self-balancing analysis and improved topology of three-level converter with APOD modulation | |
| CN115912489A (en) | LMS-SOGI three-phase-locked loop design method and system suitable for non-ideal power grid | |
| CN112332426B (en) | Unified power quality regulator system based on MMC technology and control method | |
| CN111786390B (en) | Vector repetition control method and system based on harmonic sequence extraction | |
| Zhiwu et al. | Research on STATCOM simulation of H bridge cascade multilevel inverter based on CPS-SPWM control | |
| Gao et al. | A DC-link Balancing Control Strategy for NPC Three-level Shunt Active Power Filter | |
| Patel et al. | A new fundamental active current extraction scheme for multifunctional control of V2G interface | |
| CN114285304B (en) | Neutral point potential balance control method for NPC three-level DVR system | |
| CN116707313A (en) | Comprehensive coordination control method for single-phase-DC-DC converter | |
| Li | Model predictive control based D-STATCOM using three-level simplified neutral point clamped inverter |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| PB01 | Publication | ||
| PB01 | Publication | ||
| SE01 | Entry into force of request for substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| GR01 | Patent grant | ||
| GR01 | Patent grant |