CN116569292A - Improved converter performance - Google Patents

Abstract

An electromagnetic device (1) comprising a plurality of inductors (10, 11) is provided. Each inductor (10, 11) has a winding arranged near or on a single core (12), wherein the device (1) is configured such that the plurality of inductors (10, 11) are substantially independent of each other or magnetically isolated from each other.

Description

Improved converter performance

Technical Field

The field of the invention relates to power converters, and more particularly to power converters comprising an electromagnetic device having a plurality of inductors and methods of operating or controlling the power converters.

A portion of the disclosure of this patent document contains material which is subject to copyright protection. The copyright owner has no objection to the facsimile reproduction by anyone of the patent document or the patent disclosure, as it appears in the patent and trademark office patent file or records, but otherwise reserves all copyright rights whatsoever.

Background

Magnetic elements are fundamental components in many circuits, such as switching circuits. They are used, for example, in many converter circuit topologies such as buck, boost, buck-boost, flyback, forward, LLC, LCC, or class E and class F.

Depending on the size of the inductor and the on-time of the switch, different modes of operation may be used: continuous current mode, critical/boundary current mode, or discontinuous current mode.

Critical current mode and discontinuous current mode are typically used to achieve very good efficiency due to the reduction of switching losses. However, these modes also increase ripple in the inductor at the expense of losses in the inductor due to hysteresis and increased losses in eddy currents, skin and proximity effects in the inductor lines, radiation and conduction emissions, and thus generally require the use of larger cores and additional filters (both at the input and output). Thus, these topologies are typically larger compared to continuous current mode topologies.

Generally, for each mode of operation, the inductor is typically very bulky.

A common technique for reducing the size of the inductor includes the use of a multiphase system in which the converter circuit includes several parallel branches.

Multiphase converters are also known as interleaving systems, in which a plurality of parallel branches act out of phase with each other in order to reduce the current in each branch and to reduce the current ripple. The interleaving system also reduces dissipated power and component size due to the reduction in losses. Interleaving systems are typically used for high power applications, such as above 1kW applications.

However, there is still a need to provide a solution that will provide an inductor that achieves high efficiency while being smaller in physical size, even in low power applications.

The wireless power alliance (WPC) defines a switching frequency between 110kHz and 205kHz in its Qi wireless inductance standard, which can deliver up to 30W and is widely adopted by most phone manufacturers. The coupling factor k is in the range of 0.75 to 0.5 (charging distance of about 2mm to 7 mm).

When the distance increases beyond 10mm, the coupling factor k between the transmitter coil and the receiver coil becomes low and the Qi system is generally unable to transfer energy.

WPC recommends tuning the natural frequency of Qi wireless systems to about 100kHz using LC resonance tanks, where the inductance L is given by the transmit coil. This allows the wireless charger to function close to maximum power transfer while remaining safely away from resonance and thus simplifying feedback loop control.

Inductive wireless power transfer is a well-known technology that is capable of recharging and supplying electronic devices from a few watts up to several kilowatts.

Conventional wireless power transfer systems for low power applications are typically composed of several power conversion stages, including:

● A transmitter comprising an AC/DC isolation adapter, an optional DC/DC power converter (typically a buck converter, a boost converter or a buck-boost converter), and a DC/AC converter for coil driving.

● An isolation barrier between a transmit coil located in the transmitter and a receive coil located in the receiver.

● A receiver comprising an AC/DC rectifier circuit and a DC/DC power converter (linear converter or switching converter) capable of functioning as a battery charger, voltage regulator or current regulator.

Because the multiple stages in the wireless power transfer system each add some inefficiency, the overall system can typically achieve only up to 60% of the efficiency measured from AC input to DC output.

The wireless power transmitter that excites the transmit coil may be configured to use a variety of topologies including a class D topology, a class E topology, or a push-pull class E amplifier.

However, standard topologies still suffer from high radiation emissions or use a large number of components, very low efficiency or efficiency is strongly dependent on load conditions. There is a need for a simplified solution that will meet the radiation emission regulations and provide high efficiency over a wide range of load variations, while being able to provide high power.

The present invention solves the above mentioned vulnerabilities and also solves other problems not described above.

Disclosure of Invention

The present invention relates to an electromagnetic device comprising a plurality of inductors as described in the claims. The apparatus is configured such that the plurality of inductors are substantially independent or magnetically isolated from each other.

A comprehensive list of key features is at the appendix.

Drawings

Aspects of the invention will now be described, by way of example, with reference to the following drawings, each of which shows a feature of the invention:

fig. 1 shows a top view of a single inductor used in a converter system and two inductors in parallel used in an interleaving system.

Fig. 2 shows a cross-sectional view of a single inductor used in the converter system and two inductors implemented in the interleaving system.

Fig. 3 shows an electromagnetic device having multiple independent inductors wound on a single core of magnetic material.

Fig. 4 shows a cross-sectional view of an assembled electromagnetic device comprising a plastic part or coil former.

Fig. 5 shows different views of two inductors implemented in parallel in an interleaved converter system compared to an assembled electromagnetic device having two independent inductors.

Fig. 6 shows a further example of a winding configuration including a calibration gap.

Fig. 7 shows an electromagnetic device configuration with three multiple inductors.

Fig. 8 shows a weakly coupled transformer.

Fig. 9 shows a graph of drain voltage of an interleaved boost converter implementing a weakly coupled transformer (where k=0).

Fig. 10 shows a graph of inductor current for an interleaved boost converter implementing a weakly coupled transformer (where k=0).

Fig. 11 shows a graph of drain voltage of an interleaved boost converter implementing a weakly coupled transformer (where k is about 0.35).

Fig. 12 shows a graph of inductor current for an interleaved boost converter implementing a weakly coupled transformer (where k is about 0.35).

Fig. 13 shows a circuit diagram of a Power Factor Conversion (PFC) circuit including a weakly coupled transformer.

Fig. 14 shows a graph of waveforms associated with branch 131 of the boost converter.

Fig. 15 shows a perspective view of a 3D CAD model of a plastic part (or former) for winding wires of a weakly coupled transformer.

Fig. 16 shows a top view of a 3D CAD model of a plastic part for winding wires of a weakly coupled transformer. When not specified, dimensions are expressed in mm.

Fig. 17 shows a 3D CAD model cut-away view of a plastic part for winding a wire of a weakly coupled transformer.

Fig. 18 shows an assembled weakly coupled transformer.

Fig. 19 shows an LLC converter comprising a half-bridge implementation of a switching network and a full-wave rectifier.

Fig. 20 shows a graph of output voltage in dB, illustrating a change in resonant frequency associated with an increase in load.

Fig. 21 shows an example of an LLC resonant converter.

Fig. 22 shows an LLC resonant converter.

Fig. 23 shows an example of a winding configuration of an LLC resonant converter.

Fig. 24 shows a further example of a winding configuration of an LLC resonant converter.

Fig. 25 shows a further example of a winding configuration of an LLC resonant converter.

Fig. 26 shows a further example of a winding configuration comprising virtual wire windings.

Fig. 27 shows an example of a winding configuration comprising planar windings printed on a PCB.

Fig. 28 shows another example of a winding configuration including planar windings printed on a PCB.

Fig. 29 shows a graph of control signals for different winding configurations.

Fig. 30 shows a graph of control signals for different winding configurations.

Fig. 31 shows a graph of the voltage vout of fig. 22 using different winding configurations.

Fig. 32 shows a graph of the voltage vout of fig. 22 using different winding configurations.

Fig. 33 shows an example of a circuit capable of generating an active low enable signal for an ideal diode controller.

Fig. 34 shows a graph of the voltage of the different signals of fig. 33.

Fig. 35 shows a conventional wireless charging circuit.

Fig. 36 shows a conventional wireless power transmitter implemented using a class D topology.

Fig. 37 illustrates a conventional wireless power transmitter implemented using a class E topology.

Fig. 38 shows a conventional wireless power transmitter implemented using an E-analogized topology.

Fig. 39 shows a graph of the signal of the circuit of fig. 38.

Fig. 40 shows a single isolation device integrating an AC/DC converter and a wireless charger.

Fig. 41 shows an integrated AC/DC converter and wireless power transmitter arranged as a single stage.

Fig. 42 shows a proposed short-range topology of an integrated AC/DC converter.

Fig. 43 shows a graph of the current measured at the inductor L1.

Fig. 44 shows line trends for push-pull, switching nodes, coil voltage, and coil current.

Fig. 45 shows a proposed length Cheng Tapu of an integrated AC/DC converter.

Fig. 46 shows the line trends for push-pull, switching nodes, coil voltage and coil current.

Fig. 47 shows a circuit diagram of a single-stage bridgeless and capacitor-less wireless architecture.

Fig. 48 shows a circuit including a resonant class D inverter with the addition of a sensing network.

Fig. 49 shows a graph of the voltage at the node between L1 and C1.

Fig. 50 shows a graph of the voltage at the node between R1 and R2.

Fig. 51 shows a graph of the voltage at the node between C2 and R3.

Fig. 52 shows a diagram of a distance calibration setting.

Fig. 53 shows the working principle of the calibration arrangement.

Fig. 54 shows a diagram of an insulated converter with a secondary side circuit configured as a voltage multiplier.

Fig. 55 shows a diagram of an insulated converter with a secondary side circuit configured as a full bridge circuit.

Fig. 56 shows a diagram illustrating different phases of an insulated converter.

Fig. 57 shows a diagram illustrating different phases of an insulated converter.

Fig. 58 shows a diagram illustrating different phases of an insulated converter.

Fig. 59 shows a diagram illustrating different phases of an insulated converter.

Fig. 60 shows a diagram illustrating different phases of an insulated converter.

Fig. 61 shows a diagram illustrating different phases of an insulated converter.

Fig. 62 shows a diagram illustrating different phases of an insulated converter.

Fig. 63 shows a diagram illustrating different phases of an insulated converter.

Fig. 64 shows a diagram of an isolated converter used as an isolated PFC.

Index

Electromagnetic device 1 with multiple inductors

First inductor 10 wound on a single magnetic core

Second inductor 11 wound around a single core

Magnetic core 12

Gap 13 of first lateral leg of magnetic core

Gap 14 of second lateral leg of magnetic core

The magnetic flux path 15 of the first inductor

Magnetic flux path 16 of the second inductor

Center leg 17 of the core

Base part 41 of plastic part or coil former

Top part 42 of plastic part or coil former

First inductor 70 of an electromagnetic device comprising three inductors

Second inductor 71 of electromagnetic device comprising three inductors

Third inductor 72 of electromagnetic device comprising three inductors

Center leg 74 of electromagnetic device including three inductors

Visible gap 75 on one of the lateral legs

Visible gap 76 on the other lateral leg

Weakly coupled transformer 80

Primary winding 81 of a weakly coupled transformer

Secondary winding 82 of a weakly coupled transformer

Magnetic core 83 of weakly coupled transformer

Center air gap 84 of weakly coupled transformer

First lateral air gap 85 of weakly coupled transformer

Second lateral air gap 86 of weakly coupled transformer

First branch 131 of the boost converter

Second branch 132 of boost converter

Switch or MOSFET 133 of branch 131

Switch or MOSFET 134 of branch 132

First switch 220 of LLC converter

DC voltage input 221 of LLC converter

Half bridge or switching node 222

Second switch 223 of LLC converter

Ground input 224 of LLC converter

Center leg 230 of single core of weakly coupled transformer

Lateral leg 231 of single core of weakly coupled transformer

Lateral leg 232 of single core of weakly coupled transformer

End member 233 of a single core of a weakly coupled transformer

End member 234 of the single core of a weakly coupled transformer

Air gap 235 of center leg

Primary winding 236 of a transformer

Secondary winding 237 of transformer

Center leg 240 of a single core of a weakly coupled transformer

Lateral leg 241 of the single core of a weakly coupled transformer

Single core lateral leg 242 of a weakly coupled transformer

End member 243 of a single core of a weakly coupled transformer

End member 244 of a single core of a weakly coupled transformer

Air gap 245 of center leg

Primary winding 246 of transformer

Secondary winding 247 of transformer

Primary winding 250

First secondary winding 251

Second secondary winding 252

Center leg 253 of magnetic core

Air gap 254 of center leg

Primary winding 260

Coil former 261

Opening 262 for insertion of magnetic core

Virtual winding 263

Secondary winding 264

Separation layer 266 between primary and secondary windings

Calibration coil former 267 for secondary winding

Substrate or PCB 280 for printed planar inductor

Transmitter coil 520 of wireless charger

Wireless repeater 521

First inductor 523 of a wireless repeater

Second inductor 524 of the wireless repeater

Series resonant capacitor 525 for wireless repeater

Detailed Description

The specification is organized around the following categories or core technologies:

section I improved converter performance

Chapter II improved LLC converter

Chapter III Wireless charger

Section IV. insulation converter

Section I improved converter performance

1.1 multiple independent inductors on a single core

Multiphase converters, also known as interleaved converters, generally achieve the following objectives:

● Reducing the dissipated power (in the case of a two-branch system, a 2-fold reduction in current results in a 4-fold reduction in the dissipated power in each branch by R x 2 and thus halving the overall dissipated power).

● The size of the component is reduced due to the reduction of losses. In general, the multiplication of the number of components far exceeds the compensation for the high power (e.g. hundreds or thousands of watts) that is generated, in particular, by the size reduction.

● Output ripple is reduced.

● Radiation and conduction emissions are reduced due to noise reduction.

Fig. 1 and 2 illustrate a comparison of inductor sizes for a converter using a single inductor and an interleaved converter system using two inductors in parallel.

Fig. 1A and 2A show top and cross-sectional views of a single inductor, while fig. 1B and 2B show top and cross-sectional views of two inductors in parallel implemented in an interleaving system. The theoretical dimensional and volumetric efficiency losses associated with the construction problems of the inductor itself are apparent.

Due to the reduction in losses, the interleaving system is able to reduce the dissipated power as well as the component size. Interleaving systems are typically used for high power applications, such as above 1kW applications. This is due in part to the fact that: each component requires a smaller amount of "scrap" area for packaging, pinout, soldering space on the board, minimum distance from other components, etc., than the size of the active part of the core in the high power converter.

In contrast, for lower power (such as hundreds or tens of watts) devices, the wasted space is greater in size than the saved space, and therefore interleaving solutions are not typically selected.

Referring to fig. 3, an electromagnetic device 1 is provided having a plurality of individual inductors (10, 11) wound around or on a single core 12 of magnetic material. The multiple inductor structure includes a magnetic core 12 having lateral legs and a center leg. Unlike standard transformers in which there is a gap on the center leg, there is no gap on the center leg, but a gap (13, 14) on each lateral leg in order to avoid saturation of the core 12.

As shown, the magnetic flux path of the first inductor 15 is independent of the magnetic flux path of the other inductor 16. Thus, the two inductors (10, 11) are substantially independent or magnetically isolated from each other. The magnetic energy of the magnetic flux path of the first inductor 15 is substantially concentrated in the first lateral leg air gap 13 and the magnetic energy of the magnetic flux path of the second inductor 16 is substantially concentrated in the second opposite lateral leg air gap 14. In addition, in this configuration, the center leg 17 of the core effectively has a zero or near zero magnetic field.

Two separate inductors are obtained by winding the inductors on a portion of the core that has no gaps, provided that the windings are not on the center leg, and the direction of the windings is selected to ensure that the magnetic field is eliminated or nearly eliminated in the center leg 17.

The electromagnetic device may be configured such that the multiple coupling k between the inductors (10, 11) is close to 0.

Electromagnetic devices may be used to improve the performance of the converter. For a multiphase converter, the coupling k may be configured to be, for example, between 0 and 0.4.

The plurality of inductors are highly manufacturable using standard materials and standard manufacturing processes, and wherein the inductors are wound on standard bobbins in opposite directions.

Additional advantages include, but are not limited to: the physical size is reduced compared to standard applications for similar costs; the permeability of the desired core is reduced and thus cheaper magnets can be used; multiple inductors with multiple phases may be implemented on a single magnetic core; the inductors may each be driven in a random manner.

Referring to fig. 4, a cross-sectional view of an assembled electromagnetic device is shown, comprising a plastic part or former (41, 42) comprising two dedicated channels for winding the wire of the inductor (10, 11). The coil former comprises 8 pins, a top part 42 for fixing the wire, and a bottom part 41 for soldering the electromagnetic device to the PCB.

Fig. 5 and 6 illustrate a comparison between the standard interleaving configuration as shown in fig. 1 and 2 and the custom design as shown in fig. 4. Exhibiting a substantial reduction in physical size.

Fig. 5A and 5C show top and cross-sectional views of two inductors in parallel implemented in an interleaving system. Fig. 5B and 5D show top and cross-sectional views of an assembled electromagnetic device having two independent inductors.

Many other configurations are possible, such as but not limited to: an asymmetric winding configuration or a planar structure.

Alternatively, the windings of the inductor may be wire windings or planar windings printed on a substrate or a combination thereof, as shown in fig. 6.

The windings may be individually ground wound or implemented on a printed circuit board (planar inductor), the core may have a common or custom shape including, but not limited to, toroidal, EFD, E, POT, P, PQ, RQ, etc. Fig. 6 shows an example of a transformer based on E-core and I-core (wire winding as shown in fig. 6A, 6B and 6C, PCB plane in fig. 6D, hybrid plane and wire winding as shown in fig. 6G and 6H) and toroidal core (fig. 6E and 6F) where k is close to 0.

Additional modifications are provided in section II below.

The proposed multiple inductor architecture may include two or more inductors on a single core. Referring to fig. 7, an exemplary electromagnetic device is shown that includes three independent inductors (70, 71, 72), wherein the three lateral legs each include a gap and the center piece 74 does not include a gap. Two lateral gaps (75, 76) are visible.

Multiple independent inductor architectures or electromagnetic devices may be used in any circuit using multiple phases in parallel in order to reduce average current.

The electromagnetic device may be used in multiple interleaving systems and/or push-pull systems such as: boost converters, buck-boost converters, and resonant (interleaved LLC, class E, or class F) converters.

More specifically, the proposed electromagnetic device may be implemented as an output inductor of a buck converter or an input inductor of a boost or class E converter, or an inductor of each interleaved or push-pull converter.

The electromagnetic device may also be used in PFC (power factor correction) converters based on one of the converters listed above, wherein the primary feedback variable is therefore not the output voltage or current applied to the load, but the input current, which is in phase with the input voltage in the PFC converter.

Multiple independent inductor systems may also be used in interleaved DC/AC converters (e.g., inverters for renewable energy applications, wireless chargers, or hybrid and electric vehicle applications).

In a 300W interleaved PFC boost design, a single 0.5-1$ or less weakly coupled inductor may be substituted for the two 0.7$ inductors. Additional minor BOM cost improvements (about 0.1-0.3$) can be obtained by replacing the FET with a lower capable FET, for example, with higher channel resistance or output capacitance.

1.2 weakly coupled transformer

An electromagnetic device in which a plurality of inductors arranged on the same core are configured to have weak mutual coupling k will now be described.

The electromagnetic device or weakly coupled transformer may be implemented as part of a PFC interleaved boost application or may be implemented as part of an LLC application. Weakly coupled transformers may also be applied to any architecture using interleaved or multi-phase converters.

In comparison, in a generic DC/DC interleaved or push-pull converter, the main single inductor is split into two inductors with lower current ratings and lower dimensions. The disadvantage of this solution is that it increases the total cost due to component multiplication and the total magnetic area is under-used at low power. The same problem can occur in LLC applications where the need for resonant inductors and transformers results in two bulky magnetic components. A single core solution may increase both the magnetic cost and the core area usage.

Referring to fig. 8, a weakly coupled transformer 80 is provided that includes a primary winding 81 and a secondary winding 82 disposed on or near a single core having a center leg and two lateral legs. The weak coupling between the two inductors is created by introducing an air gap 84 on the center leg of the magnetic core 83 and air gaps (85, 86) on the lateral legs. As shown, the center air gap on the center leg is smaller than the lateral air gap.

The main idea is to enhance the dynamic performance of the FET of the interleaved converter by creating a weak coupling between the inductors. By deliberately designing a bad transformer or a weakly coupled inductor, the first inductor can be driven as a separate inductor, while some coupling is still due to the second inductor and isolation still needs to be made. This weak coupling can reduce the hard switching voltage in the FET, eventually forcing zero voltage switching in the FET without changing the operating principle of the interleaved converter based on an independent inductor. Thus, a number of advantages are realized: compared to standard interleaving solutions, a single magnetic component is used instead of two, and a higher efficiency is achieved due to the lower switching voltage. In contrast to an ideal transformer that exhibits perfect coupling between different windings (k=1), a weakly coupled transformer may refer to a transformer that is not optimized to have a coupling factor k close to 1. More generally, a transformer with k <0.95 may thus be considered as weakly coupled.

Similarly, as described above, the windings of the inductor may be wire windings or planar windings printed on a substrate or a combination thereof.

An example of use in which a weakly coupled transformer is applied to an interleaved boost converter will now be described.

Having a weak coupling between the two inductors increases the slope of the current during the discharge phase of the converter (i.e. for a boost converter FET in an off condition). The increased slope allows for higher current to be generated during the recovery time of the diode that results in a better (less lossy) on condition of the low-side MOSFET due to the discharge of the parasitic capacitance of the low-side FET.

Even if this condition globally worsens both the input peak-to-peak current and the reverse recovery loss of the boost diode (which can be minimized by replacing the diode with another FET), the inventors can improve efficiency due to lower hard switching voltages.

An increase in the k-factor allows for a greater improvement in hard switching conditions but with higher currents in the magnets. Thus, a trade-off between k=0 (minimum magnetic loss) and 0< k <1 (minimum loss in FET) can be chosen in order to maximize global loss.

In the absence of coupling (k=0), the hard switching condition (v_drain) and the inductor current are shown in fig. 9 and 10, respectively.

Considering the case of 300W interleaved boost converters, a mutual coupling k of less than 0.4 (such as about 0.35) has been chosen to significantly reduce hard switching losses without introducing other significant drawbacks. For k of about 0.35, the hard switching condition (v_drain) and the inductor current are shown in fig. 11 and 12, respectively.

In general, the mutual coupling k may be selected based on a plurality of parameters such as: specific technology for magnets and FETs, input/output voltages and currents. Thus, each circuit may result in a different optimal mutual coupling k.

The weakly coupled transformer may be implemented with the following circuits, but is not limited to:

● A multiphase interleaved DC/DC converter;

● A plurality of output DC/DC converters;

● A current multiplying rectifier.

PFC converters comprising weakly coupled transformers achieve 99% efficiency; the schematic diagram is shown in fig. 13, and the weakly coupled transformer (k < 1) is modeled as two inductors in series in each branch: self inductance (k=0) L1 and L2 and mutual inductance (k=1) L3 and L4.

By comparison, current PFCs can only achieve very high efficiency with bulky inductors. Standard PFCs for the same application will typically yield an efficiency of about 97% to 98%, but with greater physical dimensions and about two costs.

Thus, small size, high efficiency, and low cost can be achieved singly or in combination.

Applications include any architecture including a power conversion stage for fixing or altering a voltage, such as but not limited to:

● An application delivering between 30W and 300W of power;

● Delivering applications up to 1 kW;

● Delivering applications up to 50 kW;

● Power sources for TV, high power laptop computers, home appliances, electric and hybrid vehicles;

● All power sources including isolation barriers;

● A switched mode power supply;

● A power converter;

● A silicon chip (including a timing circuit and a microprocessor) including a controller;

● Electronic devices tailored for specific applications;

● Very high power applications, including fixed output and fixed input and including a primary stage tuned to an operating point;

● USB power delivery: input change from 90V to 265V in the united states and the european union-energy change (square of voltage) -output change from 5V to 20V-laptop-multiple output-other solutions-PFC across the barrier.

● If the load is fixed or if there is a wide variation in the load, the topology changes.

● Fixed case: 2 sequential converters specifically tuned to the operating point.

Coupling interleaving

Referring to fig. 13, a circuit topology is provided that includes an interleaved boost converter implementing two branches of a weakly coupled transformer. The weakly coupled transformer consists of two weakly coupled inductors l_a and l_b.

Referring to fig. 14, the waveform refers to the branches 131 of the interleaved boost converter (see fig. 13). Two interleaved branches are indicated as 131 and 132.

● For symmetry reasons, in branch 132 the waveforms are identical and the phase shift is equal to 180 °.

● Multiphase converters with n branches have similar behavior, and therefore the following explanation can be extended for multiphase converters with n branches weakly coupled together.

● For simplicity, the weak coupling (k < 1) is modeled as two inductors in series in each branch: self inductance (k=0) L1 and L2, and mutual inductance (k=1) L3 and L4. The physical inductor is only 2 windings with a weak mutual inductance (0 < k < 1) that has substantially the same behavior as the model consisting of the inductors L1 and L2 and the ideal transformer comprising L3 and L4.

The inventors have appreciated that the phase in each branch of the loop is described as shown in fig. 14:

phase A

The two MOSFETs are turned off. The previously loaded L1 inductor discharges with the following slope:

Δi/Δt= (V input-V output)/L1;

this is curve I (L3) -note that the current in L1 and L3 are the same.

Phase B

When inductor L2 (self-inductance of branch 132) discharges, the drain node of branch 132 begins to oscillate due to the L-C resonator (where C is the parasitic capacitance of the component-i.e., the MOSFET), and this oscillation is reflected on the drain of branch a by the "transformer" (mutual inductance) consisting of L4 and L3. The current on L1 begins to change its slope.

Phase C

MOSFET 134 of branch 132 is turned on to charge inductor l_b (shown in fig. 14 as a series of self inductance L2 and mutual inductance L4).

A portion of the input voltage falls on L4 and the voltage is reflected on L3. Since the turn ratio in the "transformer" given by L3 and L4 is about 1:1, the voltage on L3 is the same as L4.

Since the voltage drop Vout_Drain A must remain equal to Vout and the voltage drop added over L3 is equal to the voltage drop over L4, the voltage drop over L1 is equal to Vout+Vout_L4. Depending on K, this voltage may take different values (since K affects how much voltage falls on L4). In practice, v_l4=k×v input may be shown; thus, the slope of the current will be equal to:

Δi/Δt= (V input-V output + k V input)/L1

Thus, the higher the coupling k between the two branches, the faster the slope, which is reasonable because the self-inductance of each branch is smaller.

Phase D

Branch 132 of MOSFET 133 is open.

The voltage on L4 drops to 0 and inductor L1 discharges at the same slope as phase a. It is interesting to emphasize that L4 is physically absent, since it is part of l_b, so this theoretical voltage equal to 0 is given by the superposition of effects.

Phase E

Inductor L1 is fully discharged and drain a begins to oscillate. Again-in reality there is no inductor L1, L1 and L3 are models of the same inductor l_a weakly coupled to l_b.

This provides a transition to a lower voltage level, since the coupling between the L3 and L4 portions of the voltage "falls" on the transformer (launches to the other branch) and the drain node on branch 131 drops to a voltage level below the uncoupled condition.

Phase F

When the drain voltage reaches the valley, MOSFET 133 of branch 131 turns on.

The input voltage charges the previously discharged L1 and L3. Thus, the slope of the current is equal to:

Δi/Δt=v input/(l1+l3)

Everything is also repeated periodically on the other branch.

The introduction of coupling between the two normally decoupled inductors means that the drain node tends to discharge more than the decoupled version, and therefore the lower voltage is in phase E during oscillation when MOSFET 133 is on.

Therefore, losses are smaller due to the hard switching on the MOSFET compared to a standard uncoupled multiphase converter.

The choice of such a solution is reasonable in interleaving applications where the current on the MOSFET is relatively low and thus the static losses are lower than those of hard switches. In other words, the choice of k is a trade-off between a lower on hard switching transition (which requires a higher k) and a lower current (which requires a lower k). As an example, starting from a 300W standard interleaved (k=0) design with 500mW total FET loss, a weakly coupled inductor (k=0.2) may reduce FET loss to 350mW, and more coupled inductors (k=0.4) may reduce loss up to 250 mW.

In PFC stages with interleaved topology standard european input voltage and "low" power target, this solution shows a greater efficiency (99% in case of low cost MOSFETs) compared to the decoupling solution, since the current is in any case lower due to the multi-branch converter and due to the high voltage input.

Referring to fig. 15-17, different views of a 3D CAD model of a plastic part are provided. The plastic part or former is configured to wrap around the wire of a weakly coupled transformer, where the mutual coupling k is about 0.35, which can be used in PFC outputting about 300W of power. The overall dimensions of the plastic part are about 22mm x 34mm.

Fig. 17 shows a cross-sectional view of a bobbin for winding an inductor. The coil former includes 8 pins, each of which has a top part for securing the wire and a bottom part for soldering the inductor to the PCB. The wire is wound around the coil former in two dedicated channels. The coil former is designed for a transformer based on an E-core (called E-core) and an I-core (called I-core): the wire is wound around the bobbin, then the I-core is inserted inside it, and then the block is glued to the E-core. Fig. 18 shows a finished assembled weakly coupled transformer.

Chapter II improved LLC converter

LLC resonant converters belong to a huge family of resonant converters. These are typically switching converters that include tank circuits that actively affect the input to output power transfer. LLC resonant converters are circuits based on so-called "resonant inverters", i.e. converting a DC voltage into a low harmonic content AC voltage (ideally a sinusoidal voltage) and providing AC power to a load. For this purpose, switching networks are generally used to generate a square-wave voltage, which is applied to a resonant tank tuned to its basic components. The slot will respond primarily to the component and in a negligible manner to higher order harmonics so that its voltage and/or current will be approximately sinusoidal. Then, the resonant inverter is followed by a rectifier and output filter stage; the whole system acts as a DC-DC resonant converter. In most cases, the rectifier block is coupled to the resonant inverter through a transformer to ensure isolation required by safety regulations. The rectifier block may be configured as a bridge rectifier (preferable when high voltage/low current is required for output) or a center-tapped full-wave rectifier (preferable when low voltage/high current is required for output). Depending on the configuration of the tank circuit, the low pass filter is typically made of capacitors only or of an L-C smoothing filter.

Different types of resonant inverters can be established, depending on the type of switching network and the characteristics of the resonant tank, such as the number of reactive elements and their configuration. When two inductors and one capacitor are used in series and a load is connected in parallel to one L, a so-called LLC inverter associated with an LLC DC-DC converter is obtained. One of the possible existing configurations of a half-bridge implementation with a switching network and a full-wave rectifier is shown in fig. 19.

Fig. 20 shows a graph illustrating the output voltage of the V-output in dB of fig. 21, illustrating the change in resonant frequency of the circuit of fig. 21, defined as:

fr1＝1/(2π√(Ls·Cr))

since the resonant tank is made of three reactive elements (Cr, ls and L primary), there is also another resonant frequency associated with the circuit with respect to the condition that the secondary winding is open, where the tank circuit turns from LLC to LC, since Ls and L primary can be unified in a single inductor:

fre2=1/(2pi/(ls+Lprimary) ·Cr)

Of course, this is fr1> fr2. Then, when the load changes, the actual resonance frequency fr0 of the LLC circuit becomes a function of the load moving within the range fr 1. Ltoreq.fr0. Ltoreq.fr2. At no load, fr0=fr2. When the load increases, fr0 moves toward fr 1. This implies that for f > fr1 the input impedance of the loaded resonant tank is inductive and for frequency f < fr2 the input impedance is capacitive. For fr2< f < fr1, the impedance may be inductive or capacitive, depending on the load resistance RL. There is a threshold R circuit such that if RL < R circuit the impedance will be capacitive, if RL > R circuit it will instead be inductive. For any resonant tank configuration, it can be shown that:

R circuit= v (z0.z0inf)

Where Z0 and Z0inf are resonant tank output impedances with short and open source inputs, respectively.

The change in LLC output voltage with both frequency and load is shown in fig. 20, obtained with AC analysis of the circuit of fig. 21.

LLC resonant converters are typically configured to operate in areas where the input impedance of the resonant tank is of inductive nature. This means that the impedance increases with frequency, which implies that the power transfer can be controlled by varying the operating frequency of the converter. In this way, reduced power demands from the load require a frequency increase, while increased power demands require a frequency decrease.

Referring to fig. 19, consider the following case: wherein the half-bridge driver switches both power switches (implementing but not limited to a power GaN HEMT or a power MOSFET) on and off in symmetrical phase opposition, i.e. at exactly the same time. This is often a "50% duty cycle" operation, even though the conduction time of any power switch is in fact slightly shorter than 50% of the switching period, because a small dead time is interposed between the opening of any switch and the opening of the complementary switch. The effect of this dead time is critical to the operation of the converter and will also be elucidated in the next section. At this point it will be ignored and the voltage applied to the resonant tank will be considered a square wave with 50% duty cycle swinging from 0 all the way to vin.

In the previous paragraph, the impedance of the tank circuit is mentioned. Impedance is a concept related to a linear circuit under sinusoidal excitation, whereas in the circuit of fig. 19, the excitation voltage is a square wave. However, due to the selective nature of the resonant tank, most of the power handling properties of resonant converters are associated with the fundamental components of the fourier expansion of the voltages and currents in the circuit.

The input square wave excitation has a DC component equal to vin/2. In the LLC resonant tank, the resonant capacitor Cr in series with the voltage source exhibits an average voltage also equal to vin/2 in steady state conditions, since the average voltage over the inductor must be zero. Thus, cr serves the dual function of a resonant capacitor and a DC blocking capacitor.

A number of improvements from the LLC resonant converters described above will now be described. The following techniques may also be applied more generally to other power converters including LC resonance.

LLC resonant converter with split resonant capacitor configuration

An LLC resonant converter is shown in fig. 22, which includes a first switch 220 (or high or upper MOSFET) connected between a DC voltage input 221 and a half-bridge or switching node vsw 222, and a second switch 223 (or low MOSFET) connected between the half-bridge node 222 and a ground input 224.

To improve performance, cr is split using two capacitors C1 and C2, as shown in fig. 22. The two capacitors (C1 and C2) are dynamically connected in parallel so that the capacitance of the total resonant tank is still Cr. Although the sum of C1+C2 is equal to Cr, C1 and C2 may or may not have the same value. This new configuration is useful, especially at higher power levels, to reduce the current stress in each capacitor. In addition, it makes the input current to the converter look like the input current of a full bridge converter and results in a significant reduction in both input differential mode noise and stress of the input capacitor (ideally in parallel with the input voltage). The proposed design can thus improve the performance of the converter in terms of efficiency, since a given capacitor, which can sustain the maximum peak voltage, is stressed at lower currents. In other words, for a given target efficiency, a lower cost capacitor may be used. In any case, the differential noise of the proposed converter is lower than a conventional LLC.

LLC resonant converter comprising two clamping diodes

In fig. 22, diodes D1 and D2 have been added in addition to the new split capacitor configuration. These diodes clamp the voltages of C1 and C2 between 0 and V input, thereby also acting as hardware LLC tank peak current limiters, thus providing fast cycle-by-cycle over-power protection. D1 and D2 do not have any kind of effect in regular operation, since they only intervene when overpower occurs and the efficiency is not affected. Such inexpensive and simple hardware protection cannot be achieved in the case of a single resonant capacitor, where the over-power protection must be achieved with some voltage and/or current measurement (which may affect efficiency) and software algorithms.

LLC converter with improved electromagnetic components

The system represented in fig. 19 appears to be bulky with its three magnetic components. In order to reduce size and cost without adversely affecting the characteristics of the converter, the resonant inductor and the transformer may be integrated into a single physical magnetic device, which may be modeled as two inductors weakly coupled together, as shown in fig. 22, a transformer having a coupling factor k below 1. From an electrical point of view, the leakage inductance of the weakly coupled transformer acts as previously covered by the external inductor Ls.

Such magnetic integration provides several advantages such as reduced volume and size, reduced cost (requiring only one magnetic component instead of 2, fewer raw materials), and higher efficiency (only one core being magnetized). For this reason, a high leakage flux structure is required, which is contrary to the best practice of conventional transformer design with the aim of minimizing leakage inductance. Techniques already shown in section I above may also be used.

The LLC converter also provides safety insulation. Various implementations of a weakly coupled transformer including safe insulation will now be described as examples. The following weakly coupled transformers may be implemented as part of an LLC power converter as well as any other topology where high leakage inductance may be required, such as parallel resonant converters or dual active bridge architectures.

To obtain a repeatable value, it is possible to place the windings on separate core legs (as shown in fig. 23) or side by side on the same legs (as shown in fig. 24).

Fig. 23 shows an implementation of a weakly coupled transformer comprising a single core with two lateral legs (231, 232) and one central leg 230 connected to each other using end members (233, 234). The center leg includes an air gap 235. The transformer further comprises a primary winding 236 and a secondary winding 237 each arranged on a lateral leg.

Fig. 24 shows another implementation of a weakly coupled transformer comprising a single core with two lateral legs (241, 242) and one central leg 240 connected to each other using end members (243, 244). The center leg includes an air gap 245. The transformer further comprises a primary winding 246 and a secondary winding 247, each arranged on a section of the central leg 240.

A more compact alternative design is that shown in fig. 25 for a secondary winding having a tapped configuration. The primary winding 250 and the secondary windings 251, 252 are concentric, but the two secondary windings (251, 252) rest side by side. The primary winding is arranged around a center leg 253, wherein the center leg comprises an air gap 254. Each secondary winding (251, 252) is arranged on the inside of two lateral legs (251, 252) and overlaps a portion of the primary winding 250. This solution uses the bobbin window very efficiently (e.g., the primary winding 250 may occupy the entire bobbin height). Furthermore, in this way only one or a few primary and secondary layers may be used in order to minimize power loss due to proximity effects. These optimizations allow for very good power densities and using thin shaped ferrite cores (such as but not limited to EFD types), extremely thin transformers can be built relative to standard solutions.

As an example, the weakly coupled transformer shown in fig. 25 may be implemented as part of the architecture shown in fig. 22 with two secondary inductors or windings (L3 and L4).

Thus, a 210W LLC weakly coupled transformer may be thinner than 13mm due to the EFD30 core and custom thin bobbin.

By placing the windings inside the core as shown for example in fig. 24 and 25, an improvement of the robustness of EMI (electromagnetic interference) can also be achieved.

In order to further improve the reproducibility of the leakage inductance value, it is necessary to reduce the gap tolerance between the primary winding and the secondary winding. An alternative configuration is now described in which the primary winding is first fixed and then a controlled arrangement of the secondary winding is achieved based on the position or arrangement of the fixed primary winding. Controlling the arrangement of the secondary winding includes controlling a spacing between one or more turns of the secondary winding and/or controlling a spacing between the primary winding and the secondary winding. The primary winding and the secondary winding are then configured to have a specific weak mutual coupling k determined by selecting different pitches.

Examples are shown in fig. 26A-26F, each of which includes a primary winding 260 disposed around a bobbin 261. The center leg of the core is then inserted inside the opening 262 of the bobbin.

A virtual wire winding 263 made of a conductive or non-conductive material may be placed between the two secondary windings, as shown in fig. 26A.

Virtual wire windings 265 may also be placed between each turn of the primary winding and/or the secondary winding, as shown in fig. 26B, in order to obtain a desired value of the distance between wires placed in different layers and/or within the same layer.

As shown in fig. 26B, an insulating tape 266 may be used to provide a controlled separation distance between the primary and secondary windings.

Alternatively, it is also possible to use an insulated wire consisting of one or more copper strands covered by a calibrated insulation layer (which exceeds the minimum thickness required by safety regulations), so that the distance between the conductive wires is set by the thickness of the insulation layer.

Coil former housings placed around the primary layer (or layers) may be used to define the position of the secondary layer. The bobbin may present a physical barrier defining the bobbin windows of these windings, such as, but not limited to, one or more plastic teeth separating the windings with or without a more precise wire seat. Some possible solutions are shown in fig. 26. The proposed arrangement is able to provide a very accurate separation distance between the secondary windings and the separation distance of these secondary windings from the primary winding.

The coil former may also be used to provide accurate spacing between one or more PCBs and/or one or more wire windings. Due to the spacer made of plastic or other material, the coupling factor between windings in a planar transformer can be controlled by designing different windings on different PCBs 280 and separating them by a calibration gap (fig. 27A). Because of both the PCB 280 stack and the 2D shape of the windings, the windings can also be implemented on the same PCB, calibrating the coupling factor (fig. 27B and 28).

Alternatively, hybrid planar and wound transformers may also be used. A great advantage of this solution is the opportunity to realize all primary (or secondary) windings in the inner layer of the PCB and wire-wind the remaining windings without the use of insulated wires or strips, since insulation can be ensured by the insulating outer layer of the PCB, including but not limited to glass fibers. Examples of transformers implemented with single or double layer PCBs are provided for simplicity, but the same ideas can be extended to multi-layer PCBs. In addition, the copper traces on the outer layer may also be covered by an additional insulating PCB layer.

Method for increasing the timer resolution of an LLC converter

Standard LLC control techniques are based on symmetrical and complementary signals driving the high and low FETs, which means that the two signals have a duty cycle equal to 50% minus the dead time value and they are generated with a 180 ° phase shift, as shown in fig. 29A, where the control signal for the high MOSFET, the control signal for the low MOSFET and the resulting period are provided.

Control of the power is then obtained by varying the frequency (or period) of the signal rather than its duty cycle (as compared to other conventional DC/s like buck and boost).

When the LLC is controlled with a timer that clocks at a fixed frequency (e.g., using a timer embedded in a microcontroller), the resolution on the drive signal is limited by the clock resolution (T), so a 32MHz frequency-clocked timer will generate a signal with a resolution of t=1/f=1/32 mhz=32.25 ns. In the case of two symmetrical signals, the resolution of the total waveform is equal to 2T (fig. 29A, 29B, 30A, and 30B).

Reducing the clock frequency of the timer allows for a cheaper controller to be used and helps to reduce the power consumption of the controller from one hand, but from another hand the frequency adjustment will be coarser, which results in a poor output voltage adjustment.

Two techniques for increasing the resolution of the timer are now listed.

1) Asymmetric signal

As shown in fig. 29B, standard control typically increases or decreases by a multiple of both the "on" length and the "off" length of the two signals, and has an overall 2T resolution.

Resolution may also be halved by generating two symmetrical signals (e.g., by firmware), as shown in fig. 29C, to increase or decrease the "on" length of only one signal and the "off" length of the complementary signal.

In a single cycle, the two signals will have slightly different duty cycles, but after each cycle or after a limited number of cycles, an equal average duty cycle can be obtained, exchanging the two asymmetric signals (fig. 29D).

2) Controlling jitter

In the event that the timer resolution is too coarse, it may not be possible to generate an ideal signal that can provide zero error adjustment. Thus, the standard control algorithm will therefore bounce between coupling of signals that provide too much power (error < 0) or too little power (error > 0). This may occur each time a control routine is executed, thus generating a significant output ripple, as shown in fig. 30A.

The proposed control dithering technique may be applied to any LLC power converter and more generally to any other power converter topology.

One way to reduce such ripple is to increase the control frequency so that it bounces more frequently between the two operating points.

The "non-violent" accurate adjustment is obtained to generate a signal pattern to cause the device to function above the control routine frequency around the ideal operating point (fig. 30B-30D).

The control system may consist of a digital timer for generating the control signals for the FETs and a digital controller responsible for configuring the digital timer period, duty cycle, etc. In this case, the controller generates a signal pattern consisting of a series of timer configurations so that after each switching cycle the timer can take the configuration of the next cycle. Alternatively, starting from a given configuration, the timer may internally generate the signal pattern by slightly modifying the configuration from cycle to cycle, i.e. adding or subtracting a given value from the period and/or duty cycle of the signal.

Soft start technique

As described above, the LLC is typically controlled to generate several signals with a (almost) fixed 50% duty cycle for each of the two primary side FETs, and a variable frequency/period shared by the two signals, and a 180 ° phase shift between the two signals. The frequency of the signal is lower (higher) at higher (lower) output loads.

Conventional soft start management will start at high frequency and decrease it until the soft start is over and steady state is reached.

Due to hardware, firmware, or other types of limitations, it may not be possible to generate the output signal at the desired high frequency (i.e., if the converter is functioning at a high frequency at full load, the higher frequency of soft start may be too high for a low cost controller), thus different approaches are proposed.

Starting with the hardware completely disconnected (no electrical and/or magnetic energy in any capacitor and/or inductor and/or transformer), the high-side FET is responsible for powering the whole system starting from the input voltage.

Unbalanced signal (Soft start and light load conditions)

During soft start, the two FETs are driven with an unbalanced signal, with a (very) high duty cycle for the low-side FET and a (very) low duty cycle for the high-side FET.

Then, the duty cycle of the high-side FET is gradually increased while the low-side duty cycle is decreased until the two duty cycles match. This may be done at a fixed frequency or at a fixed low-side FET conduction time (increasing the high-side FET conduction time) or with other techniques.

Referring to the LLC in fig. 22, fig. 31 and 32 show a standard start-up routine (fig. 31A and 32A), where starting at a high frequency and 50% duty cycle, and where ton and toff are slowly increased to increase power and an unbalanced low duty cycle method (fig. 31B and 32B), where ton starts at a higher value and toff is slowly increased.

Each curve represents a fixed ton and toff configuration. In case a, each curve is obtained, increasing both ton and toff by 100ns (from 100ns to 500 ns). In case B, toff is fixed and toff is increased by 100ns steps (from 100ns to 500 ns).

This example helps understand that when using a low resolution timer (in this example, a resolution of 100 ns), the method used in case B can be used to achieve smoother and more accurate soft start.

The efficiency of LLC is not good enough in light loads due to the need to drive FET with high frequency drive signals. In contrast, the proposed unbalanced duty FET drive technique also serves to maximize converter efficiency under light loads due to the reduced operating frequency and reduced amount of reactive current of the converter under light load conditions.

Then, when the high duty cycle and the low duty cycle match, conventional frequency control techniques may be used.

Secondary side drive technique

In many applications, FETs controlled by an ideal diode controller are often a good choice for the rectifying stage of a power converter. In the case of a primary-side controlled converter, the advantage of this solution is that synchronous rectification is achieved without the need to generate a signal on the primary side and send it to the secondary side through an isolation barrier.

Because of the need to ensure noise tolerance and avoid spurious switching pulses, ideal diodes are typically configured with minimum and maximum conduction times. This may not be compatible with the very high frequency and/or low conduction time algorithms used during soft start.

Thus, a safe soft start routine is proposed that enables the controller of the rectifier to remain idle until the soft start is over. The circuit in fig. 33 is an example of a circuit capable of generating an active low enable signal for an ideal diode controller. In order to command M1 and M2 as ideal diodes, two conditions are required: the output voltage must be above a threshold (in order to properly supply the controller) and the enable signal must be below the threshold (which is an active low signal). At circuit start-up, C7 is initially not charged, so the enable signal is initially equal to the output voltage: the ideal diode does not turn on the FET because the output voltage is too low to supply the controller. During this phase, D3 and D4 are able to rectify the secondary winding voltage, so the output voltage slowly increases towards the output nominal voltage. D3 and D4 may be body diodes embedded on M2 and M1 or may be external diodes used only during soft start. During the D4 (D3) on time, D6 (D5) turns on, providing a path for slowly charging C7 and pulling the enable voltage toward 0V. After a certain amount of time (typically longer than the soft start duration), the output voltage is significantly high and the active low enable voltage is significantly low: the ideal diode will begin to operate normally.

In fig. 34, an analog active low "enable" signal is shown. For better understanding, a "rect_enable" logic active high signal is also shown, which is high when "enable" is below a certain threshold and the output voltage is above another threshold.

Chapter III Wireless charger

3.1 background on Wireless charging

Referring to fig. 35, a conventional wireless charging circuit is provided in which power is transferred from a wireless charger to a receiver device by inductive coupling. A wireless charger generally includes:

● AC/DC isolating adapter (input 110V-240V, output DC up to 48V);

● DC/DC power conversion (optional);

● A DC/AC power amplifier for coil driving.

The basic degrees of freedom of a wireless power system are:

● The switching frequency (fsw) of the DC/AC power amplifier;

● Input voltage DC of DC/AC power amplifier

● A coupling factor (k) between the transmission coil and the receiving coil.

The power delivered to the receiver device may be controlled by:

● Fixed frequency: the DC/AC stage of the transmitter switch is switched at a fixed frequency that is higher than the natural frequency of the LC tank. The control is accomplished by varying the input voltage of the DC/AC stage using a DC/DC converter. This is typically the preferred method in the automotive field (to reduce EMI problems and simplify authentication of the device) and when one of the apple-specific quick charge algorithms is required (one of the apple quick charge algorithms is based on a switching frequency of about 127 kHz).

● Variable frequency: in this case, a DC/DC converter is not required, as any DC constant input voltage may be used. The control is done by adjusting the switching frequency of the DC/AC stage. Specifically, the switching frequency is increased so as to reduce the power transmitted to the load (the distance between the natural resonance of the LC tank and the switching frequency is increased), and the switching frequency is decreased so as to increase the output power (in this case the switching frequency is closer to the natural resonance of the LC tank).

The wireless power transmitter may be configured using a variety of topologies such as:

● Class D topology: as shown in fig. 36, this is mainly suitable for controlling the resonant tank in half-bridge and full-bridge configurations. The main drawbacks include radiation Emission (EMI), which can be partially solved by using LC filter stages. Furthermore, such architecture requires high-side driving that can be implemented by bootstrap circuits or the like. In addition, zero voltage switching may be implemented for only a narrow range of loads.

● Class E topology: as shown in fig. 37, this is a well-adopted topology for wireless power transfer and typically has high frequency applications and low k (k < 0.5), such as the airfiel standard at 6.78 MHz. Advantages of class E topologies include low emission, only one low-side MOSFET, extremely high efficiency at full load due to the ability to achieve ZVS operation. However, it typically requires fixed input, output and drive operations, otherwise ZVS is difficult to achieve. In addition, there is a high peak voltage on the drain and any load variation makes ZVS operation difficult or impossible.

● Push-pull class E topology (see fig. 38 and 39). However, the architecture includes a large number of capacitors in parallel with the main capacitor, and in turn uses a large number of MOSFETs for regulating the reactive load located in the large number of capacitors in parallel with the main capacitor. The resonant tank C2/L4 (FIG. 38) is tuned at a natural frequency of 100 kHz. The frequency transfer is at 127.7 kHz. As mentioned above, one significant disadvantage of this implementation is that ZVS is easily lost with load variations for standard class E amplifiers. Therefore, the tank capacitance (such as C1-C3-C7-C8 in FIG. 38) must be dynamically switched in order to remain under ZVS conditions.

3.2 integration of uninsulated AC/DC and Wireless charging (insulation)

Referring to fig. 40, a single isolation device integrating an AC/DC converter and a wireless charger is provided.

The wireless charger includes a single insulated non-conductive (i.e., plastic) housing that houses the AC/DC converter and the wireless transmitter circuitry. Thus, there is no need for an insulated AC/DC adapter that is typically provided in an additional insulated housing. And because the AC/DC conversion function is integrated into the transmitter coil circuit housing, the AC/DC converter may therefore be uninsulated.

In addition, the need for high loss DC cables between the conventional AC/DC isolation adapter and the wireless transmitter housing is also eliminated. Instead, the proposed design includes an insulating housing capable of supporting the interface to directly receive the AC input voltage.

Because a safety transformer is not required, the uninsulated AC/DC converter is also smaller and more efficient than an insulated converter. Thus, there is no need to convert electrical energy to magnetic energy and vice versa.

The elimination of the AC/DC adapter also eliminates the need for a low voltage, high current (and therefore high loss) cable between the adapter and the wireless power transmitter, thus further reducing losses and allowing longer cables to be used.

As an example, the flyback stage may become an uninsulated buck step-down converter due to the lack of a transformer that results in higher losses compared to standard inductors, thereby improving efficiency. The AC/DC module includes both an input stage filter and a diode bridge, while the "DC/DC buck" is a synchronous buck converter that can function in a forced conduction mode to achieve ZVS and reduce losses in the active device.

A second advantage is to freely drive coils with higher voltages than usual without the DC limit of the standard AC/DC adapter of 50V given by the low voltage operation due to safety regulations.

The topology is particularly applicable to long range (k < 0.5) wireless charging solutions, where high range requires high input voltages, but also to short range transmitters (k > 0.5), where the DC/AC converter may be a standard converter such as class D or class E.

There are fewer serial stages, fewer components, higher efficiency, and greater output power than existing architectures.

3.3 Integrated AC/DC and Wireless charger in Single stage

Referring to fig. 41, the integrated AC/DC converter and the wireless power transmitter are provided as a single stage. As shown, the bridge rectifier is directly connected to the drive circuit, thus eliminating the need for a DC/DC buck converter.

The architecture is particularly suitable for long range (k < 0.5) wireless charging solutions, where high range requires high input voltages, but also for short range transmitters (k > 0.5). For example, the coil drive circuit may be a standard coil drive circuit (such as class D or class E) or other converter capable of being directly supplied with an AC voltage and properly exciting the transmission coil.

Coil drive circuitry regulates coil voltage and power delivery using 2 methods:

● Changing the frequency of the driving circuit

● Changing the duty cycle of a driving circuit

3.4 coil drive topology for short range (k > 0.5)

The proposed short-range topology is set forth in fig. 42.

In contrast to class E coil drive circuits, the proposed architecture eliminates LC series resonance, instead of parallel and series resonant circuits, where all inductive elements L1, L3, L2, L6 and coil L4 resonate.

Unlike series connection of a resonant capacitor with a coil of standard class E topology, a multi-resonant system is not proposed:

● L1 and L3 may be sized large enough to act as choke inductors (which means as constant current sources where the current ripple is low compared to the average current over the inductor). For wireless power transfer applications compatible with the Qi standard, where the operating frequency is 100kHz-300kHz, this means that L1 and L3 should be greater than 100uH.

● L2 and L6 are calculated to resonate with the capacitor and with the transmission coil.

● Another possibility is to reduce the size of L1 and L3 and let L1 and L3 resonate together with L2 and L6.

● Finally, another possibility is to remove L2 and L6 and let L1 act as both the current source and the inductive resonator.

● In any case, the series capacitor can be eliminated compared to a standard Qi transmitter

● In contrast, a capacitor is added in parallel with the coil in order to have the circuit have a resonant current path regardless of the load, so that zero voltage switching can be maintained for each load condition and the dependency of ZVS on the load is significantly reduced

For example, a particular implementation of a wireless charger compatible with the Qi wireless charging standard includes the following components:

● L2 is a 12uH wireless charging coil and delivers energy to the output according to the coupling factor and output impedance

● L4, L6, C2 create a series resonance that is tuned slightly above the switching frequency that causes the system to tune under different loads

● C2, L2 also series-resonates according to leakage inductance given by the load

● In this example, L1, L3 are chosen to be high enough (> = 22 uH) to avoid negative current return to the source. They will have a DC current (real power) and a current ripple. The choice is a trade-off between DC loss and AC loss.

● Negative currents in L1 and L3 are also possible (and because of the push-pull implementation these currents almost cancel each other) as shown in fig. 43. In this case, L1 and L3 act as both current generator and resonator

● The switching frequency is 127.7kHz in this example

Fig. 44 shows the line trends for push-pull, switching node, coil voltage and coil current from top to bottom.

The advantages of the architecture are the following:

● A sine wave shape that results in very low radiation and performs EMI;

● Since the MOSFET drain voltage is a negative derivative at the zero voltage crossing, the tuning of the circuit is robust to significant load and coil coupling variations, created by adding a resonant current path in parallel with the coil (capacitor C2) and sizing the circuit so as not to compensate the power absorbed by the resonant network with the power absorbed from the current source at the zero voltage crossing instant: having more power absorbed by the resonant network at this instant results in reverse conduction of the MOSFET (negative current from source to drain) so that the change in load moves around the zero current instant only during T on, without affecting the zero voltage condition.

● Capacitor C2 is large enough to ensure this negative derivative/negative current even under light load conditions so that ZVS is always ensured.

● This results in a small increase in reactive current and more robustness to load variations and coil coupling than standard class E architectures, which are zero derivative zero voltage crossings. Therefore, the natural frequency of the system is not adjusted using a switched capacitor in parallel with the FET, and the cost is significantly reduced and the performance is improved.

● As a function of the proposed multi-resonant network, the peak voltage at the drain can be significantly reduced compared to standard class E. Specifically, the lower the inductors L1 and L3, the lower the peak voltage (and the higher the reactive current in the circuit). This degree of freedom can therefore be used to select the best trade-off between ensuring zero voltage switching at each load condition and lower current at the off-instants of the MOSFETs.

● Standard class E amplifiers function at a fixed 50% duty cycle and require a variable input voltage to control the power delivered to the load. In contrast, in the proposed solution, when variable frequencies are allowed, the following technique can be employed:

the tswitch of the circuit depends on the detection of a zero voltage crossing. When zero voltage is detected, the MOS is turned on

Calculating ton in order to keep the reactive energy in the system to a minimum in order to minimize the drawbacks from the negative derivative at the zero voltage crossing. A possible way to control T-on in order to keep the non-functional amount to a minimum is

■ The feedback loop on the output power required by the load is closed. If the load requires less power, T on decreases. If the load requires more power, T on increases.

■ The feedback loop on the peak voltage at the drain of the MOSFET is turned off. Since the peak voltage is related to the energy stored in the system, if the load requires less power, the peak voltage increases and ton may correspondingly decrease.

The currents in L2 and L6 are the same as the opposite sign of the AC component and L1 and L3. Thus, these inductors may be wound on the same core as the weakly coupled inductors or the individual inductors may be wound on the same core (as described in section i. improved performance of the converter).

Alternative implementations may include one or more of the following:

● C2 is removed and the system is retuned. This reduces reactive current but also makes the system less robust to load variations;

● Substitution of C2 only by C1 and C3

● Only C8 replaces C1 and C3, which has a size that is half the size of C1 and C3. This replacement is possible because C1 and C3 can be modeled as two capacitors connecting the two drains of the MOSFETs in series, and the node between the capacitors is connected to ground voltage. Thus, C8 is an equivalent single capacitor (half the size of series C1 and C3) connecting the two drains of the MOSFETs.

3.5 coil drive topology for long range (k < 0.5)

A proposed architecture for long-range applications is shown in fig. 45. The circuit replaces the series capacitance seen in the standard class E topology with a parallel coil capacitor, forming a multi-resonant system. L1, L3 and C1 resonate almost independently of L4, so the circuit is very robust to load variations or coupling variations.

In this implementation, the switching frequency may be about 128kHz and L4 is equal to 12uH.

Fig. 46 shows the line trends for push-pull, switching nodes, coil voltage differences, and coil currents from top to bottom.

Alternatively, C1 may be replaced by a pair of capacitors connected between the switching node and ground.

3.6 alternative topology for Wireless Power

The main idea is to combine the above ideas with an insulated forward AC/DC converter described in more detail in the following' section iv. An insulated forward AC/DC converter includes a weakly coupled transformer having primary and secondary windings arranged in a forward configuration.

Since the transmitter and receiver coils used in wireless power systems are characterized by a coupling coefficient k that is significantly lower than 1, they can also be considered to form a weakly coupled transformer. This topology is particularly convenient for wireless charging of vehicles, as the primary side inductance can be high enough to avoid high currents, taking into account the AC grid voltage.

Various resulting architectures are shown in fig. 47. Fig. 47A includes an AC input, while fig. 47B and 47C include a DC input. The rectifier stage may be implemented with a variety of topologies such as: full bridge, voltage multiplier, or push-pull.

Fig. 47D and 47E show different configurations, wherein the capacitor may be divided into two capacitors in series and reference the node between the input voltage and the primary winding of the transformer.

A single higher voltage capacitor may be used to achieve a higher energy density, for example in the case where the capacitor is intended to function as energy storage.

Alternatively, two lower voltage capacitors may also be used to provide a lower power density. The solution will then be cheaper with a lower Equivalent Series Resistance (ESR), so this is a good choice without requiring a large energy storage on the primary side, especially in case the voltage of the capacitor should resonate or handle high ripple currents.

The circuit operation mode is explained in detail in 'section iv. Insulating converter', and the circuit protection shown in fig. 47F to 47I is added. They refer to an AC power supply with a surge diode (fig. 47F), a DC power supply with a standard arrangement of surge diodes (fig. 47G), a DC power supply with an alternating arrangement of surge diodes (fig. 47H), and an AC power supply with a clamp diode (fig. 47I), respectively.

3.6 Capacity-free architecture

With reference to the topology described above, in the case of AC power, the idea is to eliminate the input capacitor used in the conventional topology, since it is not required in the architecture shown in fig. 47.

Thus, the storage of energy is done on the secondary side. This provides an advantage in situations where power factor correction (i.e. wireless power transmitters with input power higher than 75W) is required, as the inductive input helps to let the architecture control as PFC.

Similar to the wireless converter, the secondary side is typically a battery powered device (i.e., smart phone, tablet, phone, notebook, electric car, vacuum cleaner, etc.), and thus the battery of the device can be used as an energy store. Thus, the output of the receiver may be configured as a battery charger (i.e., the commonly used CC-CV algorithm or other battery charging algorithm), thereby completely eliminating the need for a bulky capacitor in the converter and thus significantly reducing the size of the wireless power charger.

3.7 Single stage bridgeless and Capacity free Wireless Power transfer

In addition to the topology shown in fig. 47, a single-stage bridgeless and capacitor-free primary side is also achieved in case of storage on the secondary side, employing the same architecture as the bridgeless forward AC/DC converter in the 'section iv. Isolated converter'.

Bridge removal is particularly convenient for efficiency improvement in cases where the transmit power is greater than 200W.

3.8 communication between secondary and primary side

In order to ensure optimal performance of the controller and to avoid bulky passive components, low delay communication between the secondary side and the primary side is preferred.

Standard communication protocols can be used (as in standard Qi, based on a change in impedance of the secondary side or based on frequency modulation). However, communication channels based on close coupling may also ensure lower delays as low as 10 ns. The following may be used:

● Capacitive communication (see, e.g., PCT/IB 2019/054668).

● As capacitive data coupling either a capacitive pad is used or both parasitic capacitances between the transmit coil and the receive coil are used.

● Near the antenna.

● Inductive data coupling.

As an effect of the high bandwidth low delay communication, the control adjustment may also be performed in a synchronous manner, meaning that the on signal sent to the primary side provides proper real-time rectification.

3.9 Long-range packet demodulation

The use of a long distance (i.e., 10mm to 50 mm) results in a very high resonant voltage on the coil driving part. In the wireless power alliance standard, qi packets are modulated by a receiver to a transmitter over a power channel. The modulation is called ASK (amplitude shift keying) and the receiver changes its own impedance and affects the coil voltage reflected to the transmitter. Each transition is read as either a "0" or a "1" depending on the length of time.

The higher the transmitter coil voltage, the more difficult it is for the microcontroller to detect the data packets, as they are compressed with the rest of the waveform. The problem is solved with the addition of a simple analog circuit that "cuts off" a portion of the waveform that is not of interest to the inventors, allowing them to greatly reduce the scaling factor applied to the full bridge topology as shown below.

Referring to fig. 48, a circuit is shown including a resonant class D inverter and with the addition of a sensing network. Alternatively, other transmission circuits may be used to drive the transmitter coil L1, such as but not limited to a class E topology or any other topology.

The sensing network is configured to distinguish very small voltage signals modulated over very large voltage waveforms, such as a few volts over a few hundred volts. The sensing network includes a voltage divider configured to be tuned to be able to read a desired voltage value. The sensing network includes a clipping circuit and is configured to accurately extract small variations in voltage from a very high voltage signal (as shown in fig. 49, which fig. 49 provides a graph of the measured voltage at the resonant node between L1 and C1 of fig. 48) and clip the output voltage when it is high (as shown in fig. 50, which fig. 50 provides a graph of the measured output of the voltage divider made up of R1 and R2 of fig. 48). Fig. 51 shows the final voltage collected between C2 and R3 after being switched off by the internal diode of the connected microcontroller.

The sensing network is located at a node between L1 and C1. Resistors R1 and R2 form a voltage divider. The diode D1 chops the scaled voltage together with the voltage source Va. The series connection of C2 and R3 filters the DC component of the resulting waveform and the ADC input pin of the microcontroller is connected between C2 and R3. Finally, the internal clamp diode of the microcontroller cuts off the negative part of the signal.

The voltage of the original waveform is 540Vpp. Using only resistive voltage dividers (r1=36 kOhm and r2=1 kOhm), the inventors will get greater than 14Vpp. Instead, with this approach, the voltage at the end of the circuit is less than 6V (the values listed do not reflect the values used in the circuit).

Meanwhile, since the relevant portion of the demodulation waveform is a portion close to the peak, it has not lost any information. Furthermore, va may be a variable voltage source obtained, for example, with a digital output of a microcontroller and a simple RC circuit. The inventors can thus adjust the chopping voltage based on the amplitude of the coil waveform, thereby ensuring that the inventors get the best performance from the demodulation network.

3.10 distance identification for long range system calibration

Charging the device at a variable distance, such as between 10mm and 30mm, requires selecting the power level of the sounding pulse signal used to initiate communication between the transmitter and the receiver. The use of weak signals may result in the device not being identifiable at long ranges, while the use of stronger signals may damage the receiver as it approaches.

Several solutions have been studied, such as scanning the sounding pulse voltage from low to high until a phone is detected. However, this solution is not completely safe, as the user can put the phone to the right of the receiver when it scans the high voltage sounding pulse, thus potentially damaging its device.

The proposed calibration system exploits the variation of the system quality factor when a metal object is placed over the coil.

Calibration may be performed by the end user after the wireless charging system has been under or fixed relative to a furniture such as a table. At this point, the inventors have to place a wide sheet of metal (provided with the charger) over the table and transmitter device.

The sheet is large enough that it does not affect the calibration process even if it is not perfectly aligned with the emitter device. At this point, by starting the tuning procedure (i.e., holding the button down for a certain amount of time-3 seconds), the user begins to measure the Q factor, which is then compared to some known value to determine the thickness of the table. The Q factor is reduced by the sheet metal, so the thickness of the surface can be determined quickly and automatically. At this point, a different configuration may be automatically selected, allowing for safe charging.

Finally, the anti-release system (i.e., the button is held down by the surface) ensures that when the charger is removed from under the table, calibration is automatically lost and charging is prevented until the next calibration.

This can be accomplished by storing a minimum amount of energy (small battery, super capacitor, bistable electromechanical device … …) and the system can remain calibrated once a power outage occurs.

Method for calibrating the distance of a wireless charger

On or close to the receiver, known reactive elements (i.e. coils, ferrites or reactive networks consisting of capacitors and inductors) can be used to measure the distance.

The measurement is based on a change in the natural resonant frequency of the transmitting resonant network due to the presence of another known reactive element. The closer the reactive elements are, the greater the change in the resonant behavior of the transmitting reactive network. The distance calibration setup is shown in fig. 52, while the working principle is reported in fig. 53.

As shown in fig. 52 and 53, the wireless charger includes a transmitter coil 520 positioned at a fixed distance from a wireless repeater 521.

The wireless charger is configured to measure the separation distance between the transmitter coil 520 and the wireless repeater 521. The wireless repeater 521 is then configured to optimize the power delivered to the receiving coil (not shown).

The wireless repeater 521 includes a second inductor 524 in series with a first inductor 523 and a first inductor 523 (not visible) of a series resonant capacitor 525 shaped substantially similar to the transmitter coil 520. The second inductor 524 is shaped to reshape the magnetic field to deliver maximum power to a receiving coil (not shown).

The first inductor 523 and the second inductor 524 of the wireless repeater are represented in fig. 53C as inductor L2.

To calibrate the wireless charger, the resonant frequency of the wireless charger is determined by measuring the impedance value at node V resonance as a function of frequency with and without the wireless charger (as shown in FIGS. 53B and 53D).

This approach has the advantage of simplicity while avoiding the need for the receiver to sense the distance between the transmitter coil and the receiver coil. In addition, no energy storage element is required to memorize the data, as the distance can be measured continuously. Finally, the L-C network may have dual components: a known element for sensing distance and a matching element located between the transmitter and the receiver to improve the performance of wireless power transfer.

Section IV. insulation converter

Bridgeless converters and forward converters have been used. However, they are unusual in that they require many additional components and increase stress on the active components compared to flyback converters.

4.1 insulation converter topology

An insulated converter is now described, comprising: a weakly coupled transformer comprising primary and secondary windings arranged in a forward configuration.

The proposed improved architecture is shown in fig. 54 and 55, which relates to an isolated converter (including PFC function and an insulated regulator, or only an insulated regulator or insulated PFC), and the storage element C2 is located on the secondary side circuit and the storage element C10 is located on the primary side. The transformer includes a primary winding L2 and a secondary winding L3 arranged in a forward configuration.

The transformer may be used as, but is not limited to:

● Isolating the PFC.

● An isolated converter with single or multiple outputs with or without PFC.

● Storage of the battery on the secondary side and storage on the primary side at high voltage is used.

The primary side circuit is a bridgeless circuit, and M1 and M2 act as diodes (at 50 Hz). By removing two diodes, the power loss is halved compared to a standard circuit comprising a bridge (since two diodes are required compared to four diodes). Of course, M1 and M2 may be replaced by standard diodes. In addition, standard diode bridges can be used for low current converters, where the performance differences are smaller compared to the bridgeless solution.

The architecture is bridgeless to improve power efficiency and reduce bill of materials. Thus:

● The drain of the upper switch (i.e., the drain of the upper MOSFET) is connected to the 2 nd terminal of the input source through a diode, and the anode of the diode is connected to the drain of the upper MOSFET.

● The source of the lower switch (i.e., the source of the lower MOSFET) is connected to the 2 nd terminal of the input voltage source through a diode, and the cathode of the diode is connected to the source of the switch

● MOSFETs driven as ideal diodes may also be used to replace the diodes and further improve efficiency.

The secondary side rectifying circuit may be configured as a voltage multiplier circuit as shown in fig. 54 or as a full bridge circuit as shown in fig. 55.

On the primary side, M3 and M4 are fast switching MOSFETs with high switching frequencies (such as 1MHz or 500 KHz). A capacitor C10 on the primary side circuit is connected in parallel with the switching leg comprising M3 and M4.

The two inductors L2 and L3 are arranged on the same core and have a mutual coupling k smaller than 1. In the example provided in the slides below, k is selected to be equal to 0.8 to 0.95. Therefore, the transformer including the primary side winding L2 and the secondary side winding L3 is intentionally not an ideal transformer. The two inductors L2 and L3 are also arranged in a forward configuration such that the current flowing into L2 has the same direction.

The presented isolated converter provides a number of important advantages, such as but not limited to:

● The forward configuration of the transformer minimizes the amount of magnetic energy that needs to be stored, thereby reducing the physical size of the transformer. Compared to a standard forward converter, the proposed architecture does not require an auxiliary reset winding or a reset diode, thereby reducing the number of components and simplifying the architecture. In addition, the architecture is a zero voltage switch compared to a standard forward converter, thereby significantly improving the efficiency of the converter.

● L2 and L3 are weakly coupled. Thus, although most of the energy is transferred to the load (similar to a standard forward converter), a small amount of energy is stored in L2 due to k < 1. This energy is used to ensure zero voltage switching transitions in the primary side MOSFET.

In the following description, the positive half wave of a sinusoidal AC input from the grid (50-60 Hz90-260 VAC) is considered. During this half wave, M1 is on and M2 is on.

● During phase 1 (fig. 56), primary side switch M4 is turned on. Since L2 and L3 have windings of the same polarity (forward mode), energy is transferred to the secondary side and rectified by D1 (in the case of a full bridge rectifier, the energy is rectified by D1 and D4). Meanwhile, since the coupling between L2 and L3 is smaller than 1, a small amount of energy is stored in L2 (similar to the charging phase of the boost converter). The longer the time ton during which the switch is on, the more energy is transferred to the load and stored in L2.

● When switch M4 is open (fig. 57), L2 acts as a current generator (because a small amount of energy is stored in L2). In this stage, the primary side circuit acts as a boost converter, discharging L2 and charging C10 through switch M3, which switch M3 is driven as a diode (similar to the diode of the boost converter). The voltage on C10 thus rises to a voltage higher than the grid voltage and the current in L2 gradually drops to 0A. Additionally and simultaneously, in phase 2, energy is still transferred to the secondary side due to the coupling between L2 and L3, while D1 is still conducting.

● The voltage at C10 may be determined or selected based on the mutual coupling of the inductors. The lower the coupling, the higher the voltage on C10 will be because more energy is stored in L2 during phase 1 (fig. 57). In addition, the converter may be designed to ensure that after an initial transient, the voltage at C10 is nearly constant and above the grid, acting as a storage element, or the voltage at C10 may be highly variable, oscillating from a minimum value to a maximum value.

● When L2 is fully discharged, switch M3 remains open (fig. 58) (this is the difference between the proposed converter and the boost converter). Thus, the current is reversed in L3 (phase 3). Therefore, as described above, there is no need to reset the diode. In this stage, energy is still transferred to the load through diode D2 (through D2 and D3 in the case of a full bridge rectifier, in fig. 55).

● When switch M3 is open, the current in L2 is still not zero and flows through M4 to the grid (fig. 59). Thus, the parasitic capacitance of M4 has been fully discharged and the voltage on the M4 drain drops to 0V, ensuring zero voltage switching operation.

● When the voltage on M4 is 0V, a new switching cycle may be started.

Unlike the above description, when the AC input is reversed (negative input half wave), the same cycle occurs, with M2 replacing M1, and with M3 and M4 driven in a complementary fashion (fig. 60-63).

4.2 bridgeless isolated converter for insulated PFC

The proposed converter can be used as an isolated PFC as shown in fig. 64.

Energy may be stored on the secondary side:

● At the output capacitor of the secondary side rectifier

● At the battery or supercapacitor, the battery or supercapacitor is the output of the secondary side rectifier.

Energy may be stored at a high voltage on the primary side on a C10 high voltage capacitor. The lower the coupling between L2 and L3, the higher the energy stored on C10. If a converter is used as the PFC, another DC/DC converter (or multiple DC/DC converters) in series may be required to supply each load in order to easily achieve both PFC and output load regulation. However, it is also possible to control the input current with one degree of freedom (i.e. T-on of M4) in order to achieve power factor correction, and to control the output voltage with another degree of freedom (i.e. T-off of M3) to achieve a single stage solution

Additional degrees of freedom may also be added without increasing the number of active devices. In particular, the delay of the off-instants of the secondary sides FETD1 and D2 can be used to reduce the ratio between the active energy delivered to the load and the reactive energy in the converter, thus regulating the output voltage very quickly and efficiently, avoiding additional converters in series.

4.3 bridgeless insulated converter for use as a converter without PFC

In cases where PFC is not required, the freedom of the converter of fig. 55 may be used, for example, to ensure output voltage/current regulation without the need for an additional DC/DC converter in series. Of course, in this case, the input current is out of phase with the input voltage.

In both configurations (with or without PFC), the converter has several advantages, such as one or more of the following:

● Compared to standard high power converters with PFC (typically done with pfc+llc+dc/DC converters or pfc+flyback converters), there are only 1 or 2 stages in series, thus improving efficiency.

● The circuit has better performance under light load than a standard LLC based circuit

● The main magnetic component is much smaller in size than a flyback converter (similar to a standard forward converter).

● The circuit has a smaller number of components than a standard forward converter.

● The circuit has a reduced number of components compared to other bridgeless configurations.

● The circuit is zero voltage switching resulting in very high efficiency.

● The circuit does not resonate as an LLC or class E converter and therefore is much more light duty efficient.

4.4 Battery or supercapacitor as storage element

When a battery or supercapacitor is used as a storage element on the secondary side, there are some significant advantages compared to a standard capacitor:

● The energy density (J/cm 3) is much higher in the cell than in the standard capacitor. This means that the same amount of energy can be stored in a smaller size, thereby significantly reducing the size of the converter.

● If the size of the battery is high enough (i.e. consider a 3.7V lithium battery, more than thousands of mAh-i.e. 10.000 mAh), the converter will function at the same time as the AC/DC converter and as the charger baby, creating a volume and cost saving hybrid device compared to 2 separate accessories (AC/DC adapter + charger baby) or compared to accessories integrating standard AC/DC circuits + standard charger baby circuits inside.

4.5 Primary side C10 capacitor used as storage element

Thus, by reducing the mutual coupling, such as for example with a mutual coupling k of about 0.5, less energy will be stored on the secondary side and further more voltage will be stored on the storage element C10 on the primary side. In this case, there are some advantages compared to storing energy on the secondary side:

● This approach has the advantage of high voltage energy storage (thus reducing storage capacitance, as the energy follows the rule e=1/2×c×v ζ2). This results in a reduction in the size of the storage capacitor compared to secondary side storage.

● Another advantage of such storage is the fact that-compared to conventional adapters without PFC-the same high voltage C10 capacitor can be used to store energy at high voltage, regardless of the input AC voltage. In contrast, in standard adapters, the input capacitor is quite inefficient because it must withstand the high voltage of the European Union grid (and therefore less efficient in terms of F/cm 3), and at the same time must have a large capacitance and low resistance to store energy at low voltage and higher current when the U.S. grid is connected (and therefore the input capacitor is very bulky). In contrast, using C10 as a storage element means that the C10 will be regulated as the output of a standard boost converter, so that energy can be stored at high voltage in a very efficient way regardless of the input voltage (as in a standard boost PFC-but with only 1 or 2 stages in series instead of 3).

4.6 light load conditions

As the load increases, the duty cycle increases. In contrast, for light load conditions, the duty cycle is reduced.

When light loading occurs, it may be problematic to reduce the duty cycle of M1 too much. Controlling the duty cycle for light load conditions is often very complex or very expensive, as it requires the use of expensive timers.

The proposed solution for reducing the duty cycle is to switch off the high side mosfet m2 on the primary side when the current in the transmit coil is 0A and the voltage in the capacitor is at its maximum (the instant before the current is reversed from the capacitor to the input source).

The system can then remain off for a long period of time and then restart under ZVS conditions when a new cycle is required.

Advantages of this solution include:

● High efficiency is achieved.

● ZVS is achieved even under light load conditions. As a comparison this is not possible for a standard burst mode controller.

● A zero current condition is also achieved.

● The off-time may be changed in a continuous manner.

Alternatively, this may also be applied to other non-isolated topologies, where the load is directly coupled in parallel to the C10.

4.7 Surge diode

Depending on the technology used to implement the electronic switch, the topology may be embedded in a body diode (such as a silicon FET) or any other mechanism that allows current to flow from the source to the drain, even with a low drive signal (i.e., a gallium nitride FET).

When a DC or AC voltage is first applied to the input of the circuit, the body diode may provide a current path from the input voltage to the initial discharge capacitor. The capacitor may then begin to charge quickly with very high current, which in turn may overstress the switch. To protect the switch, a surge diode is added to the circuit to carry surge currents that might otherwise damage the switch.

Several solutions are presented: in the case of an AC input, both FETs must be protected, while in the case of a DC input, only one switch must be protected, since the other switch does not provide a capacitor charging current path.

To protect the low-side FET, the anode of the diode must be connected to the source of the FET and the cathode can be connected to one of the two terminals of the primary side winding (fig. 47F and 47H).

To protect the high-side or upper MOSFET, the cathode of the diode must be connected to the drain of the FET and the anode may be connected to one of the two terminals of the primary side winding (fig. 47F and 47G).

4.8 clamping diode

During start-up, light load conditions, reversible and irreversible fault conditions or for other reasons, the high frequency switching FETs may all be turned off for an undefined time (up to a few seconds). In addition, if there is an embedded body diode or similar behavior of the FET, the body diode then behaves like a voltage multiplying rectifier of the AC input voltage. Under such conditions, the voltage across the capacitor will be equal to twice the input voltage.

In many countries, the upper tolerance of the 230VAC nominal rail is about 265VAC, which means a peak voltage of 373V, and twice the peak voltage is 747V.

If the FET and the output capacitor can all individually hold this voltage, no additional protection is required. If this is not the case, some voltage clamping (such as zener diodes, transient voltage suppressors or MOVs) may be required.

A single clamp diode may be connected in parallel with the half bridge (fig. 47I, option "c"), or both clamp diodes may clamp the voltage of each FET with the same options described for the inrush current limiting diodes (fig. 47I, option "a" and option "b").

Appendix-main features

In this section, the inventors have disclosed various concepts and features into the following categories or core techniques:

section I improved converter performance

Chapter II improved LLC converter

Chapter III Wireless charging

Section IV. bridgeless isolated converter

It is noted that different concepts or methods and features may be combined with each other. For simplicity, the inventors have organized features related to certain higher-level features or concepts; however, this is generally only a preferred implementation, and skilled practitioners will understand that the features should not be construed as limited to the particular context in which they are introduced, but may be deployed independently.

Section I: improved converter performance

1.1 multiple independent inductors with k less than 0.4 on a single core

Concept a-multiple independent inductors with windings arranged on a single core

An electromagnetic device comprising a plurality of inductors, each having a winding disposed near or on a single core, wherein the device is configured such that the plurality of inductors are substantially independent or magnetically isolated from each other.

Optional features:

● The magnetic flux path of the inductor is independent of the magnetic flux path of the remaining inductor disposed on the single core.

● The apparatus is configured such that the mutual coupling k between the plurality of inductors is zero.

● The apparatus is configured such that the plurality of couplings k between the inductors is less than 0.1.

● The apparatus is configured such that the plurality of couplings k between the inductors is less than 0.4.

● Each inductor is electromagnetically coupled to the single core while avoiding saturation of the core.

● For a multiphase converter, k=0 to 0.4.

● The windings of the inductor are wire windings.

● The windings of the inductor are planar windings printed on a substrate.

● The primary winding and the secondary winding are planar windings printed on the same substrate.

● The primary winding is printed on one side of the substrate and the secondary winding is printed on the other side of the substrate.

● Windings are printed on the inner layer of the substrate.

● The windings of the inductor are a combination of wire-wound groups and planar windings.

● Advantages include, but are not limited to: the physical size is reduced compared to standard applications for similar cost, ease of manufacture; the permeability of the desired core is reduced and thus cheaper magnets can be used; multiple inductors with multiple phases may be implemented on a single magnetic core; the inductors may each be driven in a random manner.

Concept B-multiple independent inductors with windings arranged on a single core with multiple air gaps

An electromagnetic device comprising a plurality of inductors having windings disposed adjacent to or on a single core, wherein the device is configured such that the plurality of inductors are substantially independent or magnetically isolated from each other, and wherein the single core comprises a plurality of air gaps, each air gap associated with an inductor such that magnetic energy of the magnetic flux path of each inductor is substantially concentrated within the air gap associated with the inductor.

Optional features:

● The inductors are wound on different portions of the single core so as to avoid wire-to-wire magnetic coupling.

● The single core includes a center leg that does not include an air gap therein.

● The center leg effectively includes a zero or near zero magnetic field.

● The center leg effectively includes a zero or near zero magnetic field regardless of the excitation signal driving each inductor.

● The excitation signals driving the plurality of inductors are asymmetric.

● The inductor excites the center leg with opposing magnetic fields such that the resulting magnetic field is zero or near zero.

Concept C-specific Structure of two independent inductors on a Single core

An electromagnetic device, comprising:

(i) A single core having two lateral legs and a central leg connected to each other with end members, wherein the lateral legs include an air gap,

(ii) A first inductor and a second inductor, each inductor having a winding disposed about the end member;

and wherein the first winding and the second winding are substantially independent or magnetically isolated from each other.

Optional features:

● The magnetic energy of the magnetic flux path of the first inductor is substantially concentrated in a first lateral leg air gap and the magnetic energy of the magnetic flux path of the second inductor is substantially concentrated in a second opposing lateral leg air gap.

● The first winding and the second winding are wound in opposite directions.

● The center leg of the core does not include an air gap.

● The center leg effectively includes a zero or near zero magnetic field.

● The center leg effectively includes a zero or near zero magnetic field regardless of the excitation wave input in each leg.

● The inductor excites the center leg with opposing magnetic fields such that the resulting magnetic field is zero or near zero.

● The winding of the first inductor is disposed on a portion of the end member between the first lateral leg and the center leg, and the winding of the second inductor is disposed on a portion of the end member between the center leg and the second opposing lateral leg.

● The magnetic flux path of the first inductor is independent of the magnetic flux path of the second inductor.

Concept D-specific structure of multiple independent inductors on a single core

An electromagnetic device, comprising:

(i) A single core having a plurality of lateral legs and a central leg connected to each other with end members, wherein the lateral legs include air gaps,

(ii) A plurality of inductors having windings arranged around the end member;

and wherein the apparatus is configured such that the plurality of inductors are substantially independent or magnetically isolated from each other.

Optional features:

● The center leg of the core does not include an air gap.

● The center leg effectively includes a zero magnetic field.

● Each inductor is electromagnetically coupled to the single core while avoiding saturation of the core.

● The plurality of inductors are wound on the single core in the same direction and driven with the same input signal with different phases (the phase difference between subsequent windings is equal to 360/n degrees, where n is the number of inductors).

● When the mutual coupling k between the plurality of inductors is very low, such as less than 0.1, the plurality of inductors are wound in a random manner.

● The excitation signals driving the plurality of inductors are random independent signals.

● The inductor excites the center leg with opposing magnetic fields such that the resulting magnetic field is zero or near zero.

Concept E-converter comprising multiple independent inductors on a single core

A power converter comprising an electromagnetic device comprising a plurality of inductors having windings disposed near or on a single core, wherein the device is configured such that the plurality of inductors are substantially independent or magnetically isolated from each other.

Optional features:

● The converter is a boost converter and the electromagnetic device corresponds to an input inductor.

● The converter is a class E converter or a class F converter and the electromagnetic means corresponds to the input inductor.

● The converter is a PFC and the electromagnetic means corresponds to the input inductor.

● The converter is a buck converter and the electromagnetic means corresponds to an output inductor.

● The converter is any other converter such as LLC, LCC, cuk or Sepic converter, forward converter or asymmetric half-bridge flyback converter.

● The converter is an interleaved power converter.

● The converter is a multiphase interleaved converter and the number of independent inductors on the single core corresponds to the number of phases of the converter.

● The converter is a multiphase interleaved converter having n different phases, where each phase is offset 360/n degrees in the drive signal.

● The converter delivers up to 1 kw.

● The transducer delivers up to 500 watts.

● The transducer delivers up to 300 watts.

● PFC applications deliver about 200 watts to 300 watts.

● PFC applications have >75W input (i.e., due to regulatory requirements)

● The converter does not comprise any cooling elements.

● The converter does not include a fan.

● The converter does not include a heat sink.

● The inductor may generate opposing magnetic fields on the center leg, allowing for a reduction in core size.

● The cost of MOSFETs and diodes is reduced compared to standard MOSFETs because lower current values are required.

● Efficiency is improved by implementing multiple inductors with multiple phases and lower current values required for each phase.

● For power converters operating at about several hundred KHz, the permeability of a single core is less than 100, such as 40 (compared to conventional solutions using a permeability of a single core greater than 100 or even greater than 1000 for similar frequency range operation).

Conception F-multiphase power converter comprising multiple independent inductors on a single core

A multiphase power converter comprising an electromagnetic device comprising a plurality of inductors having windings disposed near or on a single core, wherein the electromagnetic device is configured such that the plurality of inductors are substantially independent or magnetically isolated from each other, and wherein the multiphase power converter is integrated on a single core.

Concept G-method of manufacturing an electromagnetic device comprising a plurality of independent inductors

A method of manufacturing an electromagnetic device comprising a plurality of inductors, each having a winding disposed adjacent to or on a single core, the method comprising winding the plurality of inductors using a conventional bobbin,

and wherein the electromagnetic device is configured such that the plurality of inductors are substantially independent or magnetically isolated from each other.

Optional features:

● The plurality of inductors are wound on the same conventional bobbin in opposite directions.

1.2. Weak coupling transformer

Specific structure of conception A-weakly coupled inductor

An electromagnetic device, comprising:

(i) A single core having two lateral legs and one central leg connected to each other with end members, wherein the central leg and the two lateral legs each include an air gap, the air gap of the central gap being smaller than the lateral leg air gap;

(ii) A plurality of inductors having windings arranged around the end member; and wherein the plurality of inductors are configured to have a weak mutual coupling k, wherein k is determined by selecting a ratio between the center gap and the lateral gaps.

Optional features:

● The center gap is smaller than the lateral gaps.

● Reducing the center gap reduces the coupling between the plurality of inductors.

● The center gap is half as large as the lateral gap (to achieve a k of about 0.5).

● The center gap is four times smaller than the lateral gap (to achieve a k of about 0.35).

● k is less than 0.5.

● k is less than 0.4.

● k is greater than 0.5 and less than 0.95

● The primary side is driven in a resonant or quasi-resonant manner.

● The inductor is configured to operate in a discontinuous current mode.

● k depends on the magnetic current or hard switching voltage conditions.

● For resonant applications, k=0.5 to 0.8.

● The windings of the inductor are wire windings.

● The windings of the inductor are planar windings printed on a substrate.

● The primary winding and the secondary winding are planar windings printed on the same substrate.

● The primary winding is printed on one side of the substrate and the secondary winding is printed on the other side of the substrate.

● Windings are printed on the inner layer of the substrate.

● The windings of the inductor are a combination of wire-wound groups and planar windings.

Concept B-PFC converter including weakly coupled inductor

PFC converter comprising an electromagnetic device, wherein the electromagnetic device comprises a plurality of inductors having windings arranged near or on a single magnetic core, wherein the electromagnetic device is configured such that the inductors have a weak mutual coupling k.

Optional features:

● The inductors are configured such that the mutual coupling k is up to 0.35.

● The inductors are configured such that the mutual coupling k is up to 0.5.

● The value of the mutual coupling is selected based on one or more of the following: parameters of the core, type of MOSFET, required input voltage and required output voltage.

● The current required to drive the inductor is reduced by the interleaved polyphase operation, which in turn reduces the required MOSFET and diode sizes.

● K is chosen to be the optimum value to minimize global losses (mainly to minimize inductor losses (which means K- > 0) and MOSFET losses (which means K > 0), the higher the parasitic capacitance and input voltage of the MOSFET, the higher the K required).

● The mechanism is to transfer energy from one branch to the other through weak coupling, thus reducing the voltage on the MOSFET drain before turning on the MOSFET. This increases losses in the inductor due to higher RMS currents and is why a compromise is required.

● The weak coupling k between the plurality of inductors is determined to enable drain discharge of an active MOSFET used in the PFC converter.

● The MOSFET is selected based on parasitic capacitance (in standard applications, parasitic capacitance is generally considered to limit the operation of the PFC converter, including its operating frequency or switching speed, the parasitic capacitance of the MOSFET is used herein to increase the overall efficiency of the PFC converter while reducing overall cost, as cheaper MOSFETs may be used.)

● PFC achieves 99% efficiency for a particular PFC converter application using the following parameters: k is about 0.35 and the EU grid (230 VAC rectification) has an output power of about 300W.

● Improved efficiency is achieved for high frequency operation with a core having a smaller size and lower cost than standard cores for similar operating frequencies (as an example, PFC converters delivering 300W power are implemented using standard cores conventionally used in applications delivering about 30W of power, thus saving the size and cost of the magnet). Compared with a standard multi-branch interleaving converter; the weak coupling between the inductors can increase the overall efficiency of the converter (requiring lower turn-on voltages).

Use case application:

● Power supplies for TV, high power laptop, household appliances.

● A switched mode power supply.

● USB power delivery.

Chapter II improved LLC converter

Concept a-LLC resonant converter with split resonant capacitor configuration and weakly coupled transformer

An LLC resonant converter comprising:

(i) A first switch connected between the DC input and the half-bridge node;

(ii) A second switch connected between the half-bridge node and a ground input;

(iii) A resonant tank, comprising:

a resonant inductor, said resonant inductor being connected to said half-bridge node,

a first resonant capacitor connected to the positive terminal of the transformer and the DC input,

a second resonant capacitor connected to the negative terminal of the transformer and the DC input;

wherein the transformer is connected to the resonant inductor and comprises a primary winding and a secondary winding, the secondary winding being connected to a rectifying circuit for providing a rectified DC voltage to an output load, and wherein the transformer is configured as a weakly coupled transformer.

Optional features:

● The windings are wire windings.

● The windings are planar windings printed on the substrate.

● The primary winding and the secondary winding are planar windings printed on the same substrate.

● The primary winding is printed on one side of the substrate and the secondary winding is printed on the other side of the substrate.

● Windings are printed on the inner layer of the substrate.

● The windings are a combination of wire windings and planar windings.

Concept B-LLC resonant converter further comprising two clamp diodes

The LLC resonant converter as defined above further comprises two clamping diodes.

Optional features:

● Each clamping diode is connected in parallel to one of the resonant capacitors.

● The LLC resonant converter includes a plurality of resonant capacitors.

● The LLC resonant converter includes a plurality of clamp diodes.

Concept C-technique for improving timer resolution

A method of operating a switching signal of an LLC power converter, wherein the LLC power converter includes a first switch connected between a DC input voltage and a half-bridge node, and a second switch connected between the half-bridge node and a ground input;

the method comprises generating at a control subsystem a first switching signal and a second switching signal for controlling the first switch and the second switch of the LLC power converter,

wherein the first and second switching signals are generated as asymmetric signals by increasing or decreasing the on-time of one of the switching signals by a specific duration, and wherein the duration is substantially similar to the maximum resolution of the control subsystem.

Optional features:

● The first switching signal and the second switching signal are generated as asymmetric signals by increasing or decreasing an on-time of one of the switching signals by a specific duration.

● At each cycle, the control subsystem generates the switching signal based on a previous cycle or cycles.

● The control subsystem generates a pattern of switching signals such that the average duty cycles of the two switching signals are substantially similar.

● At a cycle, the on-time of one of the switching cycles is increased or decreased by the specific duration, and at a subsequent cycle, the on-time of the other switching cycle is increased or decreased.

● The control subsystem can change the manner in which each switching signal is generated at each cycle or after a certain number of cycles such that the average duty cycle of the two switching signals is substantially similar.

● The control subsystem is implemented as firmware by a digital signal processor.

● The control subsystem comprises a digital timer for generating the switching signal and a digital controller for configuring parameters of the digital timer, such as a period or a duty cycle.

● The pattern of the switching signals generated is preprogrammed at the control subsystem.

● The pattern of the switching signal is generated by the digital timer.

● Defining "NMN" as the on-time of a nominal 50% duty cycle, and "INC" as the asymmetrically increased on-time, some possible examples of sequences of HIGH-LOW signal on-times would be:

1. INC-NOM/repeat.

2. INC-NOM/NOM-INC/repeat.

3. NOM-NOM/INC-INC/NOM-NOM/INC-INC/repeat.

4. INC-NOM/NOM-NOM/INC-NOM/NOM-NOM/repeat.

5. INC-NOM/NOM-NOM/NOM-INC/NOM-NOM/repeat.

Concept D-method of controlling an LLC converter during soft start

A method of controlling a switching signal of an LLC power converter during soft start, wherein the LLC power converter includes a first switch connected between a DC input voltage and a half-bridge node, and a second switch connected between the half-bridge node and a ground input;

the method comprises generating at a control subsystem a first switching signal and a second switching signal for controlling the first switch and the second switch of the LLC power converter,

and wherein during the soft start, the first switching signal has a significantly lower duty cycle than the second switching signal.

Optional features:

● The first switching signal and the second switching signal are generated as asymmetric signals.

● The off-time of the first switching signal is significantly higher compared to the on-time of the first switching signal.

● The first switch is configured to energize the LLC power converter starting from the DC input voltage.

● Soft start refers to gradually turning on the power converter to avoid stressing the components by abrupt current or voltage surges.

Concept E-secondary side controller to keep the rectifier idle until soft start is over

A method of controlling a switching signal of an LLC power converter during soft start, wherein the LLC power converter includes a first switch connected between a DC input voltage and a half-bridge node; and a second switch connected between the half-bridge node and a ground input; and a transformer comprising a primary winding and a secondary winding, the secondary winding being connected to a rectifying circuit for providing a rectified DC voltage to an output load;

the method comprises generating at a primary side control subsystem a first switching signal and a second switching signal for controlling the first switch and the second switch of the LLC power converter during said soft start,

and wherein the method further comprises the steps of: the secondary side rectifier is kept idle at the secondary side control subsystem until the soft start is over.

Optional features:

● The secondary side control subsystem is configured to keep the rectifying circuit off for a predetermined duration.

● The predetermined duration corresponds to a duration of the soft start.

● The rectifier circuit includes a switch or ideal diode and the secondary side control subsystem is configured to keep the switch or ideal diode of the rectifier circuit off for a predetermined duration during the soft start.

● When the rectifying switch or the ideal diode is turned off, they behave as simple diodes, allowing current to flow in only one direction.

● Additional diodes are added in parallel to one or more rectifier switches to provide a path for the rectifier current when the switches are open.

● The secondary side control circuit is implemented as firmware.

● The secondary side control circuit is implemented as hardware.

● The secondary side control subsystem consists of a first (or large) capacitor connected to the output voltage and the active low enable signal.

● The active low enable signal is connected to a second (or small) capacitor with a resistor.

● The anodes of the two low current diodes are connected to a small capacitor.

● The cathode is connected to the drains of two FETs that function as ideal diodes.

● The large capacitor is initially uncharged, so the enable signal is equal to the output voltage.

● The ideal diode controller requires both a high output voltage and a low enable signal in order to command the FETs so they are initially in an idle condition.

● When the body diode (or external diode) of the rectifier FET turns on under the influence of the secondary winding, the low current diode turns on and provides a current path capable of charging the capacitor and pulling down an active low enable signal.

● After a certain period of time, i.e. similar to the duration of the soft start routine, the output voltage is significantly high and the active low enable signal is significantly low: this causes the ideal diode controller to start commanding the FET.

Concept F-method of controlling an LLC converter under light load conditions

A method of controlling a switching signal of an LLC power converter, wherein the LLC power converter includes a first switch connected between a DC input voltage and a half-bridge node, and a second switch connected between the half-bridge node and a ground terminal;

the method comprises generating at a control subsystem a first switching signal and a second switching signal for controlling the first switch and the second switch of the LLC power converter,

wherein the first switching signal has a significantly lower duty cycle than the second switching signal,

And wherein the duty cycle of the first switching signal is optimized for light load conditions at the output load of the LLC power converter.

Optional features:

● The off time of the first switching signal is preconfigured such that low power is delivered at the output load of the LLC power converter.

● The off-time of the first switching signal is fixed.

● The on-time of the first switching cycle is increased to deliver more power at the output load.

● A light load condition refers to a load less than 10% of the peak load. Peak load refers to the highest load (in terms of power) that the power converter is configured to be able to deliver to the load.

The inventors now describe many implementations in which the LLC converter includes a transformer configured as a weakly coupled transformer. The weakly coupled transformer may also be implemented using the features described in section I above or by any of the following concepts G to J.

Concept G-weakly coupled transformers (examples shown in fig. 23)

The weakly coupled transformer includes:

(i) A single core having two lateral legs and a central leg connected to each other with end members, wherein the central leg comprises an air gap,

(ii) Two inductors having windings (wire windings and/or planes), each inductor being arranged around one of the lateral legs;

and wherein the two inductors are configured to have a weak mutual coupling k, wherein k is determined by selecting the center gap.

Concept H-weakly coupled transformers (examples shown in FIG. 24)

The weakly coupled transformer includes:

(i) A single core having two lateral legs and a central leg connected to each other with end members, wherein the central leg comprises an air gap,

(ii) Two inductors having windings (wire windings and/or planes), each inductor being arranged around the air gap on one side of the center leg;

and wherein the two inductors are configured to have a weak mutual coupling k, wherein k is determined by selecting the center gap.

Concept I-weakly coupled transformers (examples shown in fig. 25)

The weakly coupled transformer includes:

(i) A single core having two lateral legs and a central leg connected to each other with end members, wherein the central leg comprises an air gap,

(ii) A center winding disposed on the center leg,

(iii) A first outer winding and a second outer winding, the first outer winding and the second outer winding being arranged on the lateral leg;

and wherein the first outer winding and the second outer winding are configured to have a weak mutual coupling k, wherein k is determined by selecting the center gap.

Concept J-coil former for weakly coupled transformers (examples shown in fig. 23)

A weakly coupled transformer comprising: a single core, a primary winding and a secondary winding arranged around the single core; and wherein the spacing between one or more turns of the secondary winding is calibrated or controlled such that the primary winding and the secondary winding have a particular weak mutual coupling k determined by selecting different spacings.

Optional features:

● The virtual wire windings are used to calibrate the different pitches.

● The dummy lines are non-conductive.

● Coil formers are used to calibrate different spacings

● The spacing between the primary and secondary windings is also calibrated.

● A bobbin case between the primary winding and the secondary winding is used.

● An insulating tape between the primary winding and the secondary winding is used.

● The windings are insulated wires.

● Providing accurate spacing between one or more PCBs and/or one or more wire windings.

Chapter III Wireless charger

Wireless charger conceiving a-integrated AC/DC converter (insulation)

An apparatus for wireless charging includes an insulated housing including an AC/DC converter and a wireless charger.

Optional features:

● The housing includes an interface to receive an AC input voltage (the wireless charger is directly powered by the grid and does not include any input stage).

● The wireless charger includes a transmitter coil optimized based on different wireless protocols.

● The wireless charger is optimized by adjusting an operating frequency and/or duty cycle of the wireless charger.

● The wireless charger is optimized for long range wireless protocols with k less than 0.5.

● The wireless charger is optimized for short range wireless protocols with k greater than 0.5.

● The housing includes a flat top surface.

● The AC/DC converter includes a bridge rectifier to rectify an AC input voltage and a DC-to-DC converter.

● The AC/DC converter is implemented with class D, class E, half-bridge, full-bridge or any other converter topology.

● The wireless charger does not require an isolated AC/DC adapter to connect to the AC/DC converter.

● The wireless charger is capable of ZVS.

● The wireless charger is capable of ZVS for a variety of load conditions.

● The housing includes a planar surface for charging.

● The wireless charger is configured to deliver wireless power to a receiver device.

Concept B-single stage circuit comprising a wireless charger and an AC/DC converter (insulated)

An apparatus for wireless charging comprising an insulated housing comprising an AC/DC converter and a wireless charger, wherein the AC/DC converter and the wireless charger are integrated into a single stage circuit.

Optional features:

● The bridge rectifier is directly connected to the transmitter coil drive circuit.

● The coil drive circuit optimizes the wireless charger by adjusting the operating frequency and/or duty cycle of the wireless charger.

● The device is implemented using only one integrated circuit.

Conception C-wireless charger

A wireless charger includes a transmitter coil and a drive circuit configured to drive the transmitter coil, the drive circuit including an input or choke inductor and a capacitor disposed in parallel with the transmitter coil, and wherein the drive circuit resonant frequency is tuned by adjusting the choke inductor and the capacitor,

And wherein the drive circuit comprises two branches implemented in a push-pull configuration.

Optional features:

● The drive circuit optimizes the wireless charger by adjusting the operating frequency and/or duty cycle of the wireless charger.

● The drive circuit includes a plurality of branches implemented in a push-pull configuration.

● When the drive circuit includes only one branch, a DC blocking capacitor is added along the current path of the transmitter coil.

● Each branch of the drive circuit comprises only one switch or MOSFET.

● Each switch can be made of several parallel MOSFETs driven by the same control signal

● The switch is a BJT, MOSFET (Si or SiC) or GaNHEMT.

● One terminal of the inductor is connected to the input voltage, the other terminal of the inductor is connected to a drain of the MOSFET, a source of the MOSFET is connected to ground, and the drain of the MOSFET is connected to the capacitor and the transmitting coil which are made in parallel.

● The wireless charger is configured to deliver up to 65 watts to the load.

● The wireless charger is configured to deliver power to a load over a distance of less than 10 mm.

● The wireless charger is configured to operate at ZVS due to resonance of reactive components of the drive circuit.

Conception E-wireless charger

The wireless charger comprises a transmitter coil and a driving circuit based on a half-bridge topology, wherein

A first node of the transmitter coil is connected to an input source, such as an AC or DC input, and a second node of the transmitter coil is connected to two switches (such as an upper MOSFET and a lower MOSFET) via a half-bridge node,

and wherein a capacitor is connected between the drain terminal of the upper MOSFET and the source terminal of the lower MOSFET.

Optional features:

● The primary side half-bridge circuit is configured to provide a desired switching frequency.

● When the input source is an AC input, the drain terminal of the upper MOSFET is connected to the input source through a diode and the source terminal of the upper MOSFET is connected to the second node of the transmit coil.

● The source terminal of the lower MOSFET is connected to the input source through a diode; the drain terminal of the lower MOSFET is connected to a second node of the transmit coil.

● The AC input is rectified by an input bridge rectifier, followed by a large capacitor, so that the input source of the converter is quasi-DC.

● The AC input is rectified by an input bridge rectifier, followed by a small (or no) capacitor, so that the input source of the converter is a rectified sine.

● When the input source is a DC input, its positive terminal is connected to one node of the transmit coil and its negative terminal is connected to the source terminal of the lower MOSFET.

● When the input source is a DC input, the negative terminal of the DC input is connected to one node of the transmit coil and its negative terminal is connected to the drain terminal of the upper MOSFET.

● The capacitor is connected to the drain of the upper MOSFET and one terminal of the transmit coil, and the other capacitor is connected to the source of the lower MOSFET and the same terminal of the transmit coil.

● The bridgeless isolated converter is directly connected to the AC input voltage to provide a single stage wireless charger.

● The wireless charger operates at a frequency above the LC resonator resonant frequency and is capable of ZVS. When the transmit coil current is flowing toward the half-bridge node, the lower MOSFET is always off, so during the dead time, the leakage inductance of the coil pushes the current to the node, increasing its voltage until the voltage across the high-side MOSFET is zero. The high side MOSFET is then turned on at ZVS. When the transmit coil is drawing current from the half-bridge node, the high-side FET is turned off, so the node reaches zero volts, and the low-side MOSFET is turned on at ZVS.

● The receiving coil is connected to a rectifier, such as a voltage multiplication circuit or a full bridge circuit or any other rectifier circuit.

● An in-rush current diode is placed between the input terminal and the source of the low side MOSFET to limit MOSFET stress due to the start-up charging of the capacitor.

● An in-rush current diode is placed between the input terminal and the drain of the high side MOSFET to limit MOSFET stress due to the start-up charging of the capacitor.

● An in-rush current diode is placed between the half-bridge node and the source of the low-side MOSFET to limit MOSFET stress due to the start-up charging of the capacitor.

● An in-rush current diode is placed between the half-bridge node and the drain of the high-side MOSFET to limit MOSFET stress due to the start-up charging of the capacitor.

● A voltage clamping device (zener diode, transient voltage suppressor, metal oxide varistor or the like) is placed between the input terminal and the source of the low side MOSFET.

● A voltage clamping device (zener diode, transient voltage suppressor, metal oxide varistor or the like) is placed between the input terminal and the drain of the high side MOSFET.

● A voltage clamping device (zener diode, transient voltage suppressor, metal oxide varistor or the like) is placed between the half bridge and the source of the low side MOSFET.

● A voltage clamping device (zener diode, transient voltage suppressor, metal oxide varistor or the like) is placed between the half-bridge node and the drain of the high-side MOSFET.

● One or more diodes provide both inrush current protection and voltage clamp protection.

Concept F-communication between secondary side and primary side

Any of the devices or wireless chargers as described above, wherein the communication between the secondary side and the primary side is based on one of: parasitic capacitance between the transmitter and receiver as capacitive data coupling, near the antenna, or on a coupled signal inductor.

Optional features:

● The proximity antenna is an NFC (near field communication) antenna.

● Inductive data coupling based on weakly coupled signal coils.

● High bandwidth (i.e., 10 Mbps) low latency (i.e., 10 ns)

● Synchronous control of active switches on both the receiver side and the transmitter side.

Concept G-sense circuit for reading very low voltages

The sensing network is configured to distinguish very small voltage signals modulated by very large voltage waveforms, wherein the sensing network is connected at a resonant node of the LC wireless transmitter and comprises: (i) A voltage divider connected to a resonant node of the LC; (ii) A first diode and a variable voltage source connected in series to an output of the voltage divider; and (iii) a high pass RC filter connected to the output of the voltage divider with a resistor connected to ground; (iV) a second diode connected to the output node of the high pass RC filter.

Optional features:

● The second diode is connected to the ADC pin of the microcontroller.

● The variable voltage source is obtained using the digital output of the microcontroller and an RC circuit.

● The variable voltage source is sized to set the clamping threshold based on the magnitude of the voltage across the voltage divider.

● The cathode of the first diode is connected to the output of the voltage divider.

● The anode of the first diode is connected to the output of the voltage divider.

● The second diode is integrated in the microcontroller.

● The second diode is an external diode.

Concept H-distance identification for long range system calibration

A method for calibrating a distance between a wireless charger and a receiver device, the wireless charger comprising a transmitter coil and the receiver device comprising a receiver coil, the method comprising:

placing the transmitter coil at a fixed position;

placing a calibration subsystem at a fixed distance from the fixed transmitter coil;

measuring a parameter at the transmitter coil and calibrating the wireless charger for the fixed distance;

and wherein the calibration is automatically lost when the transmitter coil moves.

Optional features:

● The parameters include a quality factor, an impedance value, or a resonant frequency.

● The calibration subsystem includes a sheet of metal.

● The wireless charger is configured to deliver up to 20 watts to the load.

● The wireless charger is configured to deliver power to the load over a distance of up to 35mm or 50 mm.

Concept I-distance identification for long range system calibration including wireless repeater

A method for calibrating an operable distance between a wireless charger and a receiver device, the wireless charger comprising a transmitter coil and the receiver device comprising a receiver coil, the method comprising:

placing the transmit coil at a fixed location;

placing a wireless repeater at a fixed distance from the fixed transmitter coil;

measuring a parameter at the transmitter coil and calibrating the wireless charger for the fixed distance;

and wherein the wireless repeater comprises a first inductor optimized to receive power from the transmitter coil and a second inductor optimized to transmit power to a receiving device.

Optional features:

● The wireless repeater includes a series resonant capacitor.

● The first inductor of the wireless repeater is shaped to substantially match the shape of the transmitter coil.

● The second inductor of the wireless repeater is shaped to substantially match the shape of a receiving coil of the receiving device.

● The parameter includes a resonant frequency.

● Parameters are first measured in the absence of a wireless repeater.

Section IV. insulation converter

Conception A-insulation converter

An insulated converter, comprising: a transformer comprising a primary winding and a secondary winding arranged in a forward configuration;

wherein a first node of the primary winding is connected to an input source, such as an AC or DC input, and a second node of the primary winding is connected to an upper switch and a lower switch via a half bridge node or a switching node;

and wherein the primary winding and the secondary winding are arranged on the same core and are configured to have a weak mutual coupling k.

Optional features:

● The transformer is a weakly coupled transformer as implemented by any of the features listed above.

● The half-bridge circuit is configured to provide a desired switching frequency.

● When the input source is an AC input, the drain terminal of the upper MOSFET is connected to the input source through a diode and the source terminal of the upper MOSFET is connected to the second node of the primary winding.

● The source terminal of the lower MOSFET is connected to the input source through a diode; the drain terminal of the lower MOSFET is connected to the second node of the primary winding.

● The AC input is rectified by an input bridge rectifier, followed by a large capacitor, so the input source of the converter is quasi-DC.

● The AC input is rectified by an input bridge rectifier, followed by a small (or no) capacitor, so the input source of the converter is a rectified sine.

● When the input source is a DC input, its positive terminal is connected to one node of the primary winding and its negative terminal is connected to the source terminal of the lower MOSFET.

● When the input source is a DC input, its negative terminal is connected to one node of the primary winding and its positive terminal is connected to the drain terminal of the upper MOSFET.

● A capacitor is connected to the drain terminal of the upper MOSFET and the source terminal of the lower MOSFET.

● The capacitor is connected to the drain terminal of the upper MOSFET and one terminal of the primary winding, and the other capacitor is connected to the same terminal of the primary winding and the source terminal of the lower MOSFET.

● The bridgeless isolated converter is directly connected to the AC input voltage to provide a single stage wireless charger.

● The converter functions in a forced continuous conduction mode and is capable of ZVS. While primary winding current is flowing toward the half bridge switching node, the lower MOSFET is always off, so during dead time, the leakage inductance of the winding pushes current to the node, increasing its voltage until the voltage across the high side MOSFET is zero. The high side MOSFET is then turned on at ZVS. When the primary winding is drawing current from the switching node, the high-side FET turns off, so the node reaches zero volts, and the low-side MOSFET turns on at ZVS.

● The secondary winding is connected to a rectifier, such as a voltage multiplier circuit or a full bridge circuit or any other rectifier circuit.

● The windings are wire windings.

● The windings are planar windings printed on the substrate.

● The primary winding and the secondary winding are planar windings printed on the same substrate.

● The primary winding is printed on one side of the substrate and the secondary winding is printed on the other side of the substrate.

● The primary winding is printed on an inner layer of the substrate.

● The secondary winding is printed on an inner layer of the substrate.

● The windings are a combination of wire windings and planar windings.

● An in-rush current diode is placed between the input terminal and the source terminal of the low side MOSFET to limit MOSFET stress due to the start-up charging of the capacitor.

● An in-rush current diode is placed between the input terminal and the drain terminal of the high side MOSFET to limit MOSFET stress due to the start-up charging of the capacitor.

● An in-rush current diode is placed between the switching node and the source terminal of the low side MOSFET to limit MOSFET stress due to the start-up charging of the capacitor.

● An in-rush current diode is placed between the switching node and the drain terminal of the high side MOSFET to limit MOSFET stress due to the start-up charging of the capacitor.

● A voltage clamping device (zener diode, transient voltage suppressor, metal oxide varistor or the like) is placed between the input terminal and the source terminal of the low-side MOSFET.

● A voltage clamping device (zener diode, transient voltage suppressor, metal oxide varistor or the like) is placed between the input terminal and the drain of the high side MOSFET.

● A voltage clamping device (zener diode, transient voltage suppressor, metal oxide varistor or the like) is placed between the switching node and the source terminal of the low-side MOSFET.

● A voltage clamping device (zener diode, transient voltage suppressor, metal oxide varistor or the like) is placed between the switching node and the drain terminal of the high-side MOSFET.

● One or more diodes provide both inrush current protection and voltage clamp protection

Concept B-insulating converter used as PFC

PFC includes an insulated converter including a transformer including a primary winding and a secondary winding arranged in a forward configuration;

Wherein a first node of the primary winding is connected to an input source, such as an AC or DC input, and a second node of the primary winding is connected to an upper switch and a lower switch via a half bridge node or a switching node;

wherein the primary winding and the secondary winding are arranged on the same core and are configured to have a weak mutual coupling k.

And wherein power factor correction is obtained by controlling the converter so as to absorb a current having almost the same waveform and phase as the input voltage and a low harmonic content.

Optional features:

● The input source is AC or rectified AC.

Concept C-Battery or supercapacitor for use as a storage element

An isolated converter, comprising: a transformer comprising a primary winding and a secondary winding arranged in a forward configuration;

wherein a first node of the primary winding is connected to an input source, such as an AC or DC input, and a second node of the primary winding is connected to an upper switch and a lower switch via a half bridge node or a switching node;

and wherein the primary winding and the secondary winding are arranged on the same core and configured to have a weak mutual coupling k, and wherein the storage element is located after the rectifier circuit on the secondary side.

Optional features:

● The storage element includes one or more batteries.

● The storage element is a supercapacitor.

Concept D-primary side capacitor for use as a memory element

An isolated converter, comprising: a transformer comprising a primary winding and a secondary winding arranged in a forward configuration;

wherein a first node of the primary winding is connected to an input source, such as an AC or DC input, and a second node of the primary winding is connected to an upper switch and a lower switch via a half bridge node or a switching node;

and wherein the primary winding and the secondary winding are arranged on the same core and are configured to have a weak mutual coupling k, and wherein one or more primary side capacitors are used as storage elements.

Optional features:

● The mutual coupling k is about 0.5.

● The mutual coupling k is about 0.9.

● The storage capacitor is connected to the drain terminal of the upper MOSFET and the source of the lower MOSFET.

● The storage capacitor is connected to the drain of the upper MOSFET and one terminal of the primary winding, and the second storage capacitor is connected to the same terminal of the primary winding and the source of the lower MOSFET.

● The converter is supplied with an AC voltage or a rectified AC voltage and operates as a PFC. When the power absorbed from the input is higher (or lower) than the power delivered to the load, the excess (or required) power is stored in (or taken out of) the storage capacitor.

● In the event of a temporary input voltage drop, the converter is able to supply power to the load by extracting it from the previously charged storage capacitor.

Concept E-light load conditions

An insulated converter, comprising:

a transformer comprising a primary winding and a secondary winding arranged in a forward configuration;

wherein a first node of the primary winding is connected to an input source, such as an AC or DC input, and a second node of the primary winding is connected to an upper switch and a lower switch via a half bridge node or a switching node;

a load located at the secondary side circuit;

wherein the off-time of the isolated converter is adapted or changed in a continuous manner.

Optional features:

● The primary winding and the secondary winding are arranged on the same core and are configured to have a weak mutual coupling k.

● The primary winding is connected to two switching MOSFETs, an upper MOSFET and a lower MOSFET, and wherein the duty cycle on the primary side is reduced by controlling the upper MOSFET.

● When the voltage at the capacitor is equal to its maximum value, the upper MOSFET is turned off.

● The converter is configured to restart at zero volt or zero current conditions after a certain amount of time, with both the upper MOSFET and the lower MOSFET turned off.

● The converter is capable of achieving more than 90% efficiency under light load conditions.

● A light load condition refers to a load less than 10% of the peak load.

Concept F-rectifier MOSFET with delayed turn-off

An insulated converter, comprising:

a transformer comprising a primary winding and a secondary winding arranged in a forward configuration;

wherein a first node of the primary winding is connected to an input source, such as an AC or DC input, and a second node of the primary winding is connected to an upper switch and a lower switch via a half bridge node or a switching node;

wherein the secondary side circuit comprises a rectifying circuit and a load;

and wherein the off-time of the rectifying circuit is delayed so as to reflect that energy received on the secondary side circuit is returned to a portion of the primary side circuit via coupling between the primary winding and the secondary winding.

Optional features:

● The off-time of the rectifying circuit is delayed by a specific duration, which is determined in order to regulate the output voltage or current at the load.

● The delay is implemented by a closed loop controller, such as a proportional, integral and derivative (PID) controller.

● The delay is increased to reduce the output voltage or the current at the load.

● The delay is determined or calculated by a digital controller.

● The delay is implemented using analog circuitry.

● One or more of the rectifier switches are open with a delay and one or more switches are open without a delay.

● The converter is configured to provide PFC and output power regulation.

● Delayed disconnection techniques are used with additional degrees of freedom and are easy

Annotating

It is to be understood that the above-described arrangements are only illustrative of the application of the principles of the present invention. Many modifications and alternative arrangements may be devised without departing from the spirit and scope of the present invention. While the present invention has been illustrated in the drawings and fully described above with particularity and detail in connection with what is presently deemed to be the most practical and preferred examples of the invention, it will be apparent to those of ordinary skill in the art that numerous modifications can be made without departing from the principles and concepts of the invention as set forth herein.

Claims (136)

1. An electromagnetic device comprising a plurality of inductors, each having a winding disposed near or on a single core, wherein the device is configured such that the plurality of inductors are substantially independent or magnetically isolated from each other.

2. The electromagnetic apparatus of claim 1, wherein a magnetic flux path of an inductor is independent of a magnetic flux path of a remaining inductor disposed on the single core.

3. The electromagnetic device of any preceding claim, wherein the electromagnetic device is configured such that mutual coupling k between a plurality of inductors is zero.

4. The electromagnetic device of any preceding claim, wherein the electromagnetic device is configured such that a plurality of couplings k between the inductors is less than 0.1.

5. The electromagnetic device of any preceding claim, wherein the electromagnetic device is configured such that a plurality of couplings k between the inductors is less than 0.4.

6. The electromagnetic apparatus of any preceding claim, wherein each inductor is electromagnetically coupled to the single core while avoiding saturation of the core.

7. The electromagnetic device of any preceding claim, wherein the windings of the inductor are wire windings or planar windings printed on a substrate or a combination thereof.

8. An electromagnetic device according to any preceding claim, wherein the windings of the inductor are planar windings printed on the same substrate.

9. The electromagnetic apparatus of any preceding claim, wherein the single core comprises a plurality of air gaps, each air gap being associated with an inductor such that magnetic energy of a magnetic flux path of each inductor is substantially concentrated within the air gap associated with the inductor.

10. The electromagnetic device of any preceding claim, wherein the inductors are wound on different portions of the single core to avoid wire-to-wire magnetic coupling.

11. The electromagnetic apparatus of any preceding claim, wherein the single core comprises a central leg that does not include an air gap.

12. The electromagnetic apparatus of any preceding claim, wherein the central leg is effective to comprise a zero or near zero magnetic field.

13. The electromagnetic apparatus of any preceding claim, wherein the center leg effectively comprises a zero or near zero magnetic field regardless of an excitation signal driving each inductor.

14. The electromagnetic device of any preceding claim, wherein the single core comprises two lateral legs and a central leg connected to each other by an end member, wherein the lateral legs comprise an air gap, and wherein the plurality of inductors have windings arranged around the end member.

15. An electromagnetic device according to any preceding claim, wherein the device comprises two inductors and the magnetic energy of the magnetic flux path of a first inductor is substantially concentrated in a first lateral leg air gap and the magnetic energy of the magnetic flux path of a second inductor is substantially concentrated in a second opposing lateral leg air gap.

16. An electromagnetic device according to any preceding claim, wherein the first winding and the second winding are wound in opposite directions.

17. The electromagnetic device of any preceding claim, wherein the winding of the first inductor is disposed on a portion of the end member between the first lateral leg and the central leg, and the winding of the second inductor is disposed on a portion of the end member between the central leg and the second opposing lateral leg.

18. A power converter comprising an electromagnetic device as defined in claims 1 to 17.

19. The power converter of claim 18, wherein the converter is a boost converter and the electromagnetic device corresponds to an input inductor of the boost converter.

20. The power converter of claims 18-20, wherein the converter is a class E converter or a class F converter, and the electromagnetic device corresponds to an input inductor of the class E converter or the class F converter.

21. The power converter of claims 18-20 wherein the converter is a PFC and the electromagnetic device corresponds to an input inductor of the PFC.

22. The power converter of claims 18-20, wherein the converter is a buck converter and the electromagnetic device corresponds to an output inductor of the buck converter.

23. The power converter of claims 18 to 22, wherein the converter is a LLC, LCC, cuk or Sepic converter, a forward converter, or an asymmetric half-bridge flyback converter.

24. The power converter of claims 18 to 23, wherein the converter is an interleaved power converter.

25. The power converter of claims 18-24 wherein the converter is a multiphase interleaved converter and the number of independent inductors on the single core corresponds to the number of phases of the converter.

26. A power converter according to claims 18 to 25, wherein the converter does not comprise any cooling element, such as a fan or a heat sink.

27. The power converter of claims 18-26, wherein the converter is a multi-phase power converter, and wherein the multi-phase power converter is integrated on a single chip.

28. A method of manufacturing the electromagnetic device of claims 1-17, the method comprising winding the plurality of inductors using a conventional bobbin, and wherein the plurality of inductors are wound on the same conventional bobbin in opposite directions.

29. An electromagnetic device, comprising:

(i) A single core having two lateral legs and a central leg connected to each other by an end member, wherein the central leg and the two lateral legs each include an air gap, the air gap of the central gap being less than the air gap of the lateral legs;

(ii) A plurality of inductors having windings arranged around the end member;

and wherein the plurality of inductors are configured to have a weak mutual coupling k, wherein k is determined by selecting a ratio between the center gap and the two lateral gaps.

30. The electromagnetic apparatus of claim 29, wherein the center gap is smaller than the lateral gaps.

31. The electromagnetic apparatus of claims 29-30, wherein reducing the center gap reduces coupling between the plurality of inductors.

32. The electromagnetic apparatus of claims 29-31, wherein the center gap is half as large as the lateral gap.

33. The electromagnetic apparatus of claims 29-32, wherein the center gap is four times smaller than the lateral gaps.

34. The electromagnetic apparatus of claims 29-33, wherein the mutual coupling k is less than 0.5.

35. The electromagnetic apparatus of claims 29-34, wherein the mutual coupling k is less than 0.35.

36. The electromagnetic apparatus of claims 29-35, wherein the mutual coupling k is greater than 0.5 and less than 0.95.

37. The electromagnetic device of claims 29-36, wherein the winding of the inductor is a wire winding, a planar winding printed on a substrate, or a combination thereof.

38. The electromagnetic apparatus of claims 29-37, wherein the windings of the inductor are planar windings printed on a same substrate.

39. A PFC converter comprising an electromagnetic device according to any of claims 29 to 38.

40. The PFC converter of claim 39, wherein the mutual coupling k is selected based on one or more of: parameters of the core, type of MOSFET, required input voltage and required output voltage.

41. The PFC converter of claims 39-40, wherein the current required to drive the plurality of inductors is reduced by implementing an interleaved polyphase operation.

42. The PFC converter as defined in claims 39-41, wherein the mutual coupling k is selected such that global losses are minimized.

43. The PFC converter as defined in claims 39-42, wherein the mutual coupling k is determined to enable drain discharge of an active MOSFET used in the PFC converter.

44. The PFC converter of claims 39-43, wherein k is less than 0.4 and the PFC achieves 99% efficiency at an output power of about 300W.

45. An LLC resonant converter, comprising:

(i) A first switch connected between the DC input and the half-bridge node;

(ii) A second switch connected between the half-bridge node and a ground input;

(iii) A resonant tank, the resonant tank comprising: a resonant inductor connected to the half-bridge node; a first resonant capacitor connected to the positive terminal of the DC input and the transformer; a second resonant capacitor connected to the negative terminal of the transformer and the DC input;

Wherein the transformer is connected to the resonant inductor and the transformer comprises a primary winding and a secondary winding, the secondary winding being connected to a rectifying circuit for providing a rectified DC voltage to an output load;

and wherein the transformer is configured as a weakly coupled transformer.

46. The converter of claim 45, further comprising two clamp diodes.

47. The converter of claim 46 wherein each clamping diode is connected in parallel to one of the resonant capacitors.

48. A method of operating a switching signal of an LLC power converter, wherein the LLC power converter includes a first switch connected between a DC input voltage and a half-bridge node, and a second switch connected between the half-bridge node and a ground input;

the method comprises generating at a control subsystem a first switching signal and a second switching signal for controlling the first switch and the second switch of the LLC power converter,

wherein the first switching signal and the second switching signal are generated as asymmetric signals by increasing or decreasing an on-time of one of the switching signals by a specific duration;

And wherein the duration is substantially similar to a maximum resolution of the control subsystem.

49. The method of claim 48, wherein the first and second switching signals are generated as asymmetric signals by increasing or decreasing the on-time of one of the switching signals by a particular duration.

50. The method of claims 48-49, wherein at each cycle, the control subsystem generates the switching signal based on a previous cycle or cycles.

51. The method of claims 48 to 50, wherein the control subsystem generates a pattern of switching signals such that the average duty cycles of the two switching signals are substantially similar.

52. The method of claims 48 to 51 wherein at a cycle the on-time of one of the switching cycles is increased or decreased by the specific duration and at a subsequent cycle the on-time of the other switching cycle is increased or decreased.

53. The method of claims 48 to 52, wherein the control subsystem is capable of changing the manner in which each switching signal is generated at each cycle or after a certain number of cycles such that the average duty cycle of the two switching signals is substantially similar.

54. The method of claims 48 to 53 wherein the control subsystem is implemented as firmware by a digital signal processor.

55. A method according to claims 48 to 54, wherein the control subsystem comprises a digital timer for generating the switching signal and a digital controller for configuring parameters of the digital timer, such as a period or a duty cycle.

56. The method of claims 48 to 55, wherein the pattern of switching signals generated is preprogrammed at the control subsystem.

57. The method of claims 48 to 56, wherein the pattern of switching signals is generated by the digital timer.

58. A method of controlling a switching signal of an LLC power converter during soft start, wherein the LLC power converter includes a first switch connected between a DC input voltage and a half-bridge node, and a second switch connected between the half-bridge node and a ground input;

the method comprises generating at a control subsystem a first switching signal and a second switching signal for controlling the first switch and the second switch of the LLC power converter,

and wherein during the soft start, the first switching signal has a significantly lower duty cycle than the second switching signal.

59. The method of claim 58, wherein the first switching signal and the second switching signal are generated as asymmetric signals.

60. The method of claims 58 to 59 wherein an off time of the first switching signal is substantially higher than an on time of the first switching signal.

61. The method of claims 58 to 60, wherein the first switch is configured to power up the LLC power converter starting from the DC input voltage.

62. A method of controlling a switching signal of an LLC power converter during soft start, wherein the LLC power converter includes a first switch connected between a DC input voltage and a half-bridge node; and a second switch connected between the half-bridge node and a ground input; and a transformer including a primary winding and a secondary winding, the secondary winding being connected to a rectifying circuit for providing a rectified DC voltage to an output load.

The method comprises generating at a primary side control subsystem a first switching signal and a second switching signal for controlling the first switch and the second switch of the LLC power converter during said soft start,

And wherein the method further comprises the steps of: the secondary side rectifier is kept idle at the secondary side control subsystem until the soft start is over.

63. The method of claim 62, wherein the secondary side control subsystem is configured to keep the rectifying circuit off for a predetermined duration.

64. The method of claims 62-63, wherein the predetermined duration corresponds to a duration of the soft start.

65. The method of claims 62-64 wherein the rectifying circuit comprises a switch or ideal diode and the secondary side control subsystem is configured to keep the switch or ideal diode of the rectifying circuit off for a predetermined duration during the soft start.

66. The method of claims 62-65, wherein when the rectifying switch or the ideal diode is open, the rectifying switch and the ideal diode appear as simple diodes that allow current to flow in only one direction.

67. The method of claims 62 to 66 wherein additional diodes are added in parallel to one or more rectifier switches to provide a path for the rectifier current when the switches are open.

68. The method of claims 62-67, wherein the secondary side control subsystem is implemented as firmware.

69. The method of claims 62 to 68, wherein the secondary side control subsystem is implemented in hardware.

70. The method of claims 62-69, wherein the secondary side control subsystem includes a first capacitor connected to an output voltage and an active low enable signal, and wherein the active low enable signal is connected to a second capacitor through a resistor, and wherein the first capacitor is initially not charged such that the enable signal is equal to the output voltage.

71. The method of claims 62-70, wherein the rectifying circuit comprises a switch or ideal diode, and wherein when the output voltage is significantly high and active low enable voltage is significantly low: the ideal diode begins to operate and commands the MOSFET.

72. A method of controlling a switching signal of an LLC power converter, wherein the LLC power converter includes a first switch connected between a DC input voltage and a half-bridge node, and a second switch connected between the half-bridge node and a ground terminal;

the method includes generating, at a control subsystem, first and second switching signals for controlling the first and second switches of the LLC power converter;

Wherein the first switching signal has a significantly lower duty cycle than the second switching signal;

and wherein the duty cycle of the first switching signal is optimized for light load conditions at the output load of the LLC power converter.

73. The method of claim 72, wherein an off time of the first switching signal is preconfigured such that low power is delivered at the output load of the LLC power converter.

74. The method of claims 72-73, wherein an off time of the first switching signal is fixed.

75. The method of claims 72-74, wherein an on-time of the first switching cycle is increased to deliver more power at the output load.

76. An apparatus for wireless charging includes an insulated housing including an AC/DC converter and a wireless charger.

77. The apparatus of claim 76, wherein the housing includes an interface to receive an AC input voltage.

78. The device of claims 76-77, wherein the wireless charger comprises a transmitter coil optimized based on different wireless protocols.

79. The device of claims 76-78, wherein the wireless charger is optimized by adjusting an operating frequency and/or duty cycle of the wireless charger.

80. The apparatus of claims 76-79, wherein the wireless charger is optimized for long range wireless protocols with a mutual coupling k less than 0.5.

81. The apparatus of claims 76-80, wherein the wireless charger is optimized for short range wireless protocols with a mutual coupling k greater than 0.5.

82. The apparatus of claims 76-81, wherein the AC/DC converter comprises a bridge rectifier to rectify an AC input voltage and a DC-to-DC converter.

83. The apparatus of claims 76-82, wherein the AC/DC converter is implemented with a class D, class E, half-bridge, full-bridge, or any other converter topology.

84. The apparatus of claims 76-83, wherein the wireless charger does not require an isolated AC/DC adapter to connect to the AC/DC converter.

85. The apparatus of claims 76-84 wherein the wireless charger is capable of ZVS.

86. The device of claims 76-85, wherein the wireless charger is capable of ZVS for a variety of load conditions.

87. The device of claims 76-86, wherein the housing comprises a flat surface for charging.

88. The device of claims 76-87, wherein the AC/DC converter and the wireless charger are integrated as a single stage circuit.

89. The apparatus of claims 76-88 wherein the bridge rectifier is directly connected to a transmitter coil drive circuit.

90. The device of claims 76-89, wherein the coil drive circuit optimizes the wireless charger by adjusting an operating frequency and/or duty cycle of the wireless charger.

91. The apparatus of claims 76-90, wherein the apparatus is implemented using only one integrated circuit.

92. The apparatus of claims 76-91, wherein the wireless charger comprises a transmitter coil and a drive circuit configured to drive the transmitter coil; wherein the drive circuit comprises two branches, an input or choke inductor, implemented in a push-pull configuration, and a capacitor disposed in parallel with the transmit coil; and wherein the drive circuit resonant frequency is tuned by adjusting the choke inductor and the capacitor.

93. The apparatus of claims 76 to 92 wherein a DC blocking capacitor is added along the current path of the transmitter coil when the drive circuit includes only one branch.

94. The apparatus of claims 76-93, wherein each branch of the drive circuit comprises only one switch or MOSFET.

95. The device of claims 76 to 94, wherein each switch can be made of a plurality of parallel MOSFETs driven with the same control signal.

96. The device of claims 76-95, wherein one terminal of the inductor is connected to the input voltage, the other terminal of the inductor is connected to a drain of the MOSFET, a source of the MOSFET is connected to ground, and the drain of the MOSFET is connected to the capacitor and the transmit coil made in parallel.

97. The device of claims 76-96, wherein the wireless charger is configured to deliver up to 65 watts to a load.

98. The apparatus of claims 76-97, wherein the wireless charger is configured to deliver power to a load over a distance of less than 10 mm.

99. The apparatus of claims 76-98, wherein the wireless charger is configured to operate at ZVS due to resonance of a reactive component of the drive circuit.

100. The device of claims 76-99, the wireless charger comprising a half-bridge topology based transmitter coil and a drive circuit;

wherein a first node of the transmitter coil is connected to an input source, such as an AC or DC input, and a second node of the transmitter coil is connected to two switches (such as an upper MOSFET and a lower MOSFET) via a half-bridge node;

And wherein a capacitor is connected between the drain terminal of the upper MOSFET and the source terminal of the lower MOSFET.

101. The apparatus of claims 76-100, wherein communication between a secondary side and a primary side is based on one of: parasitic capacitance between the transmitter and receiver as capacitive data coupling, near the antenna, or on a coupled signal inductor.

102. The apparatus of claims 76-101, wherein the apparatus further comprises: a sensing network configured to distinguish very small voltage signals modulated on very large voltage waveforms, wherein the sensing network is connected at a resonant node of an LC wireless transmitter and comprises: (i) A voltage divider connected to a resonant node of the LC; (ii) A first diode and a variable voltage source connected in series to an output of the voltage divider; (iii) A high-pass RC filter connected to the output of the voltage divider through a resistor connected to ground; and (iv) a second diode connected to the output node of the high pass RC filter.

103. A method for calibrating a distance between a wireless charger and a receiver device, the wireless charger comprising a transmitter coil and the receiver device comprising a receiving coil, the method comprising:

(i) Placing the transmitter coil at a fixed position;

(ii) Placing the calibration subsystem at a fixed distance from the fixed transmitter coil; and

(iii) Measuring parameters at the transmitter coil and calibrating the wireless charger for a fixed distance;

and wherein the calibration is automatically lost when the transmitter coil moves.

104. The method of claim 103, wherein the parameter comprises a quality factor, an impedance value, or a resonant frequency.

105. The method of claims 103-104, wherein the calibration subsystem comprises a sheet of metal.

106. The method of claims 103-105, wherein the wireless charger is configured to deliver up to 20 watts to a load.

107. The method of claims 103-106, wherein the wireless charger is configured to deliver power to a load over a distance of up to 35mm or 50 mm.

108. The method of claims 103-107, wherein the calibration subsystem is a wireless repeater, and wherein the wireless repeater comprises a first inductor optimized to receive power from the transmitter coil and a second inductor optimized to transmit power to a receiving device.

109. The method of claims 103-108, wherein the wireless repeater comprises a series resonant capacitor.

110. The method of claims 103-109, wherein a shape of the first inductor of the wireless repeater is formed to substantially match a shape of the transmitter coil.

111. The method of claims 103-110, wherein a shape of the second inductor of the wireless repeater is formed to substantially match a shape of a receive coil of the receiving device.

112. The method of claims 103-111, wherein the parameter is measured first in the absence of the calibration subsystem.

113. An insulated converter, comprising: a transformer comprising a primary winding and a secondary winding arranged in a forward configuration;

wherein a first node of the primary winding is connected to an input source, such as an AC or DC input, and a second node of the primary winding is connected to an upper switch and a lower switch via a half bridge node or a switching node;

and wherein the primary winding and the secondary winding are arranged on the same core and are configured to have a weak mutual coupling k.

114. The converter of claim 113 wherein the transformer is a weakly coupled transformer as implemented by any preceding claim.

115. The converter of claims 113-114, wherein the half-bridge circuit is configured to provide a desired switching frequency.

116. The converter of claims 113-115, wherein the converter is implemented as a PFC (power factor correction) converter, and wherein the PFC is obtained by controlling the converter to absorb current having nearly the same waveform and phase as an input voltage and low harmonic content.

117. The converter of claims 113-116, wherein the primary winding and the secondary winding are arranged on a same core and configured to have a weak mutual coupling k, and wherein the storage element is located on a secondary side after a rectifier circuit.

118. The converter of claims 113-117, wherein the primary winding and the secondary winding are arranged on a same core and configured to have a weak mutual coupling k, and wherein one or more primary side capacitors are used as storage elements.

119. The converter of claims 113-118, wherein the mutual coupling k is about 0.5.

120. The converter of claims 113-119, wherein the mutual coupling k is about 0.9.

121. The converter of claims 113-120, wherein the storage capacitor is connected to the drain of the upper MOSFET and the source of the lower MOSFET.

122. The converter of claims 113-121 wherein a storage capacitor is connected to the drain of the upper MOSFET and one terminal of the primary winding, and a second storage capacitor is connected to the same terminal of the primary winding and the source of the lower MOSFET.

123. The converter of claims 113-122, wherein the converter is supplied with an AC voltage or a rectified AC voltage and operates as a PFC.

124. The converter of claims 113 to 123, wherein the converter is capable of providing power to a load by extracting power from a previously charged storage capacitor in the event of a temporary input voltage drop.

125. The converter of claims 113 to 124 wherein the off-time of the isolated converter is adapted or changed in a continuous manner.

126. The converter of claims 113-125 wherein the primary winding is connected to two switching MOSFETs, an upper MOSFET and a lower MOSFET, and wherein the duty cycle on the primary side is reduced by controlling the upper MOSFET.

127. The converter of claims 113-126 wherein the upper MOSFET is turned off when the voltage at the capacitor is equal to its maximum value.

128. The converter of claims 113-127 wherein the converter is configured to restart at zero volts or zero current conditions after an amount of time, wherein both the upper MOSFET and the lower MOSFET are off.

129. The converter of claims 113-128 wherein the converter is capable of achieving greater than 90% efficiency under light load conditions.

130. The converter of claims 113 to 129, wherein the secondary side circuit comprises a rectifying circuit and a load; and wherein the off-time of the rectifying circuit is delayed to reflect the portion of the primary side circuit to which energy received on the secondary side circuit is returned via coupling between the primary winding and the secondary winding.

131. The converter of claims 113-130 wherein an off-time of the rectifying circuit is delayed by a particular duration determined to be used to regulate an output voltage or current at the load.

132. The converter of claims 113 to 131, wherein the delay is implemented by a closed loop controller, such as a proportional, integral and derivative (PID) controller.

133. The converter of claims 113-132, wherein the delay increases to reduce an output voltage or current at the load.

134. The converter of claims 112-133, wherein the delay is generated by a digital controller.

135. The converter of claims 112-134, wherein the delay is implemented using analog circuitry.

136. The converter of claims 112-135 wherein one or more of the rectifying switches are open with a delay and one or more switches are open without a delay.