CN116073647A - DC converter and control method thereof - Google Patents

DC converter and control method thereof Download PDF

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Publication number
CN116073647A
CN116073647A CN202310206826.2A CN202310206826A CN116073647A CN 116073647 A CN116073647 A CN 116073647A CN 202310206826 A CN202310206826 A CN 202310206826A CN 116073647 A CN116073647 A CN 116073647A
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China
Prior art keywords
voltage
capacitor
switching tube
input
switching
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CN202310206826.2A
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Chinese (zh)
Inventor
何志兴
管仁锋
卫琪琪
罗安
李宗鉴
王靖
肖雨
覃俊杰
方凌清
侯仁杰
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Hunan University
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Hunan University
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Priority to CN202310206826.2A priority Critical patent/CN116073647A/en
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/01—Resonant DC/DC converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
    • H02M1/0074—Plural converter units whose inputs are connected in series
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0083—Converters characterised by their input or output configuration
    • H02M1/009—Converters characterised by their input or output configuration having two or more independently controlled outputs
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33507—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters
    • H02M3/33523—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters with galvanic isolation between input and output of both the power stage and the feedback loop
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E60/00—Enabling technologies; Technologies with a potential or indirect contribution to GHG emissions mitigation
    • Y02E60/60—Arrangements for transfer of electric power between AC networks or generators via a high voltage DC link [HVCD]

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

The invention discloses a direct current converter and a control method thereof, wherein the direct current converter comprises a plurality of cascaded power units; the power unit comprises at least two input capacitors connected in series; each input capacitor is connected with at least one switch tube in parallel; the connection midpoint of the first switching tube and the second switching tube is connected with one end of the first impedance matching inductor, and the connection midpoint of the (t-1) th switching tube and the (t) th switching tube is connected with one end of the second impedance matching inductor; the other end of the first impedance matching inductor is connected with the other end of the second impedance matching inductor through a voltage equalizing capacitor branch; the voltage-sharing capacitor branch circuit comprises at least two voltage-sharing capacitors connected in series; the connecting midpoint of two adjacent voltage-sharing capacitors is connected with one end of the resonance capacitor; the other end of the resonant capacitor is connected with one end of a primary winding of the high-frequency transformer, and the other end of the primary winding of the high-frequency transformer is connected with the connecting midpoint of the (m) th input capacitor and the (m+1) th input capacitor through a resonant inductor. The invention has the advantages of automatic input voltage equalization and automatic current equalization of the power device.

Description

DC converter and control method thereof
Technical Field
The invention relates to a power electronic circuit technology, in particular to a direct current converter and a control method thereof.
Background
In different power occasions such as a medium-voltage direct-current distribution network, a renewable energy direct-current system, a submarine power transmission and distribution system, a ship comprehensive power system and the like, the medium-voltage direct-current converter can realize electric energy conversion from a medium-voltage direct-current bus to a low-voltage direct-current bus and electric isolation, so that the medium-voltage direct-current converter becomes a research hot spot, and the input-series-output parallel-connection direct-current converter with the advantages of structural modularization, high current output capacity and the like is widely applied. However, when the traditional medium-voltage direct-current converter supplies power to different loads, the requirements of different voltage classes can be met through multistage voltage conversion, the circuit structure is complex and power balance is difficult to realize, and meanwhile, the efficiency of the system is reduced due to multistage processing of power; in addition, in the occasion with higher reliability requirements such as a submarine power transmission and distribution system, short-circuit faults often occur at the low voltage side, although the fault current can be removed by adopting a protection method adopted by the traditional medium voltage direct current converter, voltages of a plurality of capacitors and inductors are required to be sampled and error protection is easy to occur, meanwhile, due to different load characteristics of output ports, when one or more output ports have short-circuit faults, the converter is difficult to quickly isolate the failed output port, so that the multi-port output system is broken down. The resistance of the line for submarine remote power transmission is large, so that the terminal voltage drops, and the input voltage range fluctuates greatly. On the premise of medium voltage input, a wide input range direct current converter which can realize power balance, multi-port output and has a short circuit fault isolation protection function is designed to improve the efficiency and the power supply reliability of the system, and is a design difficulty of the current medium voltage direct current converter.
The conventional switched capacitor type structure (for example CN109302072 a) transmits energy from Cink-1 to Cbk-1 step by step and then to Cink, and according to this rule, the energy is transmitted from the first capacitor to the last capacitor to realize voltage equalizing of the input capacitor, and in the voltage equalizing process, the conventional switched capacitor type structure must undergo multi-step voltage equalizing energy transmission, which leads to increased loss. The traditional switched capacitor converter can not realize the current balance of a switching tube, has the problem of uneven thermal stress caused by large current impact and inconsistent current, and is difficult to be applied to high-power occasions. Meanwhile, the existing structure or control method (CN 109302072A) can only realize voltage closed-loop control of one output port, other ports are indirectly controlled, the control precision is not high, and the method is difficult to be applied to occasions with high requirements on the precision of the output voltage. And the traditional control method is frequency conversion or phase shift control, and wide-range voltage regulation is difficult to realize in the resonant converter.
Disclosure of Invention
The invention aims to solve the technical problem of providing a direct current converter and a control method thereof, which aims to overcome the defects of the prior art, does not need an additional voltage equalizing process and reduces energy transmission loss.
In order to solve the technical problems, the invention adopts the following technical scheme: a dc converter comprising a plurality of cascaded power cells; the power unit comprises at least two input capacitors connected in series; each input capacitor is connected with at least one switching tube in parallel; all the switching tubes are connected in series; the connection midpoint of the first switching tube and the second switching tube is connected with one end of the first impedance matching inductor, the connection midpoint of the (t-1) th switching tube and the (t) th switching tube is connected with one end of the second impedance matching inductor, and t is the number of the switching tubes; the other end of the first impedance matching inductor is connected with the other end of the second impedance matching inductor through a voltage equalizing capacitor branch; the voltage-sharing capacitor branch circuit comprises at least two voltage-sharing capacitors connected in series; the connecting midpoint of two adjacent voltage-sharing capacitors is connected with one end of the resonance capacitor; the other end of the resonant capacitor is connected with one end of a primary winding of the high-frequency transformer, and the other end of the primary winding of the high-frequency transformer is connected with the connecting midpoint of the (m) th input capacitor and the (m+1) th input capacitor through a resonant inductor; wherein m=k/2, k is the number of input capacitances, and k is an even number; the secondary side of the high-frequency transformer is connected with a rectifying circuit; the rectifying circuit is connected with the multi-port output voltage stabilizing circuit.
The direct current converter realizes direct transmission of high-frequency alternating current, the power transmitted by each capacitor is equal, the equal capacitor voltage can be obtained according to the ampere-second balance of the capacitors, the capacitor voltage balance is realized from the angle of equal energy transmitted by the capacitors, no additional voltage equalizing process is needed, and the energy transmission loss is greatly reduced. The current of all the switching tubes is completely consistent, the current of the load is equally divided, and the current stress of the switching devices is greatly reduced while the series connection bearing high voltage is realized.
In the present invention, k=2, t=4; each input capacitor is connected with two switching tubes in parallel.
In the present invention, k=4, t=4; each of the input capacitors is connected in parallel with one of the switching tubes. The midpoint of the connection between the first input capacitor and the second input capacitor is connected with one end of a first inductor, and the other end of the first inductor is connected with the midpoint of the connection between the first switching tube and the second switching tube; the connection midpoint of the third input capacitor and the fourth input capacitor is connected with one end of a second inductor, and the other end of the second inductor is connected with the connection midpoint of the third switching tube and the fourth switching tube.
Each voltage equalizing capacitor is connected with a clamping diode in parallel; one end of the resonance capacitor is connected with the connecting midpoint of the two adjacent voltage-sharing capacitors, the other end of the resonance capacitor is connected with the connecting midpoint of the two adjacent clamping diodes, and the connecting midpoint of the two adjacent clamping diodes is connected with one end of the primary winding of the high-frequency transformer.
The voltage-sharing capacitor of the existing direct-current converter is divided into the first voltage-sharing capacitor and the second voltage-sharing capacitor, and the clamping of the short-circuit current of each port can be realized by combining the clamping circuit (the clamping diode), so that even if one or more output ports are short-circuited, the converter can still work normally, the normal output of other non-failure ports is maintained, the high reliability of multiple ports is ensured, and the short-circuit operation can be naturally realized without any sampling and detection control.
The rectifying circuit adopts one of a half-bridge rectifying circuit, a full-bridge rectifying circuit and a synchronous rectifying circuit.
The multi-port output voltage stabilizing circuit comprises a filter capacitor; or the multi-port output voltage stabilizing circuit comprises a filter capacitor and a voltage stabilizing converter connected with the filter capacitor in parallel.
The input capacitors of all the power units are connected in series, the switching tubes are connected in series, and the voltage equalizing capacitors are connected in series.
The inductance values of all the impedance matching inductors are equal.
The invention also provides a control method of the direct current converter, which comprises the following steps:
when the input voltage is lower than a set threshold value, the first and t switching tubes of all the power units are synchronously turned on, and the other switching tubes of all the power units are synchronously turned on;
when the input voltage rises to exceed a set threshold value, the first switching tubes and the second switching tubes of all the power units are complementarily conducted, and the duty ratio is smaller than 0.25; the phase of the t-1 switching tube in each power unit leads the phase of the first switching tube in the power unit by 90 degrees, and the t switching tube and the t-1 switching tube of each power unit are complementarily conducted;
and/or the number of the groups of groups,
when the output voltage V of the ith power unit oi Lower than the expected value V of the output voltage of the power unit oi_ref When the power unit is in the power state, the first and the t-th switching tubes of all the power units are synchronously turned on, and the other switching tubes of all the power units are synchronously turned on; when outputting voltage V oi Higher than the expected value V of the output voltage oi_ref When the power supply is in the power supply state, the first switching tubes and the second switching tubes of all the power units are complementarily conducted, and the duty ratio is smaller than 0.25; phase superb of t-1 th switch tube in each power unitThe phase of the first switching tube in the power unit is 90 degrees, and the t-th switching tube and the t-1-th switching tube of each power unit are complementarily conducted; i is more than or equal to 1 and less than or equal to n;
if V oi Deviation V oi_ref The switching pulse of the t-1 th switching tube in each power unit is synchronized when the first set value is exceeded, and the duty ratio of the t-1 th switching tube in each power unit is adjusted; the voltage range of the two ends of each voltage-sharing capacitor is 2Vin/4 (2n (2n+2)) -3.5 Vin/4 (2n (2n+2)), vin is the sum of the input voltages of all the power units, and n is the number of the power units;
if V oi Deviation V oi_ref The duty ratio of the first switching tube and the last switching tube in all switching tubes is adjusted to charge all voltage-sharing capacitors when the second set value is exceeded; the voltage range of the two ends of each voltage-sharing capacitor is 3.5Vin/4 (2n (2n+2)) -4 Vin/4 (2n (2n+2));
wherein the second set value is greater than the first set value.
The invention provides a resonant cavity alternating voltage amplitude multi-degree-of-freedom control method with a plurality of output ports being independently controllable and being capable of adapting to wide-range input voltage changes, which has the voltage regulating capability of 1-2 (2 n-1) times ultra-wide input voltage, has 2n-1 voltage regulating degrees of freedom and can realize independent control of 2n-1 output ports. The voltage of each output port can be formed by combining a plurality of switch states, different output ports select different switch state combinations, independent direct closed-loop control of the voltage of each output port is realized, the degree of freedom of voltage regulation is 2n-1, and independent control of 2n-1 output ports can be realized. The multi-stage structure is characterized in that a step-up or step-down circuit is formed by combining a plurality of groups of capacitors and a plurality of groups of inductors through different switch states, and step-up or step-down is performed among the multi-stage structure, so that the upper limit and the lower limit of alternating voltage amplitude of the resonant cavity can be further expanded, and the output voltage can adapt to 1-2 (2 n-1) times ultra-wide input voltage change.
Compared with the prior art, the invention has the following beneficial effects:
(1) The self-equalizing current-equalizing switch unit realizes power balance; the voltage-sharing capacitor and the impedance matching inductor in the partial circuit form a high-frequency alternating current branch, and a loop for supplying power to a load is provided for capacitors with different potentials, so that the self voltage-sharing of an input capacitor and the self current-sharing of a power device are realized, the voltage-sharing capacitor value is not required, and even if the capacitor values are different, the self voltage-sharing can be realized.
(2) The invention realizes multi-port output through cascading of a plurality of power units; compared with the traditional medium-voltage direct-current converter, the direct-current converter can output voltages with different voltage levels without multistage processing on the premise of realizing power balance, so that the power supply requirements of different loads are met, the efficiency of a system is improved, and the circuit structure and the control method are simpler.
(3) The invention has high reliability. The overcurrent clamping circuit in the direct current converter can play a role in short-circuit protection rapidly when one or more output ports have short-circuit faults, so that the normal operation of other power units is not influenced, the reliability of the system is improved, and the direct current converter is suitable for high-reliability occasions.
(4) The invention can realize the voltage stabilization of the ultra-wide input voltage range, and a plurality of output ports are independently controllable. The control method is that the amplitude of the resonant cavity alternating voltage is controlled in multiple degrees of freedom, the resonant cavity alternating voltage has ultra-wide 1-2 n times of input voltage regulating capacity (n is the number of self-equalizing current equalizing switch units), the degree of freedom of regulating voltage is 2n-1, and independent control of a plurality of output ports can be realized.
Drawings
Fig. 1 is a basic operation schematic diagram of a dc converter according to embodiment 1 of the present invention.
Fig. 2 (a) and fig. 2 (b) are two circuit forms of the self-equalizing current equalizing switch unit in the dc converter of embodiment 1 of the present invention.
Fig. 3 is a circuit configuration diagram of the dc converter in embodiment 1 of the present invention when n=2.
Fig. 4 is a schematic diagram of a control voltage regulation range with multiple degrees of freedom of ac voltage amplitude of a resonant cavity according to embodiment 2 of the present invention.
Fig. 5 is a control block diagram of the control of the ac voltage amplitude multiple degrees of freedom for 2n-1 output ports according to embodiment 2 of the present invention.
Fig. 6 is a current balance equivalent circuit of the MOSFET of embodiment 2 of the present invention.
Detailed Description
For the purpose of making the objects, technical solutions and advantages of the embodiments of the present invention more apparent, the technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention, and it is apparent that the described embodiments are some embodiments of the present invention, but not all embodiments of the present invention. All other embodiments, which can be made by those skilled in the art based on the embodiments of the invention without making any inventive effort, are intended to be within the scope of the invention.
The terms "first," "second," and the like, herein do not denote any order, quantity, or importance, but rather are used to distinguish one element from another. The terms "a," "an," and other similar words are not intended to mean that there is only one of the things, but rather that the description is directed to only one of the things, 2, which may have one or more. In this document, the terms "comprise," "include," and other similar words are intended to denote a logical relationship, but not to be construed as implying a spatial structural relationship. For example, "a includes B" is intended to mean that logically B belongs to a, and not that spatially B is located inside a. In addition, the terms "comprising," "including," and other similar terms should be construed as open-ended, rather than closed-ended. For example, "a includes B" is intended to mean that B belongs to a, but B does not necessarily constitute all of a, and a may also include other elements such as C, D, E.
Example 1
Fig. 1 is a basic operation schematic diagram of a power self-balancing medium voltage dc-dc input multi-port output dc converter according to embodiment 1 of the present invention.
As shown in fig. 1, the dc converter of this embodiment includes a plurality of power units, where the power units include a self-equalizing and current-equalizing switch unit, an overcurrent clamping circuit, a resonant circuit, a high-frequency transformer, a rectifying circuit, and a multi-port output voltage stabilizing circuit, which are sequentially connected, and the entire dc converter has n power units, where n self-equalizing and current-equalizing switch units are cascaded. Wherein:
the self-equalizing current-equalizing switch unit at least comprises two input capacitors connected in series, four power switches connected in series, two impedance matching inductors and two equalizing capacitors.
The overcurrent clamping circuit comprises at least one diode or an active clamping circuit, one end of the overcurrent clamping circuit is connected to a connection point of one voltage-sharing capacitor and one impedance matching inductor in the self-voltage-sharing current-sharing switch unit, and the other end of the overcurrent clamping circuit is connected to a connection point of a resonant capacitor and a high-frequency transformer in the resonant circuit.
The resonant circuit is composed of inductance capacitance series-parallel connection, and comprises a series resonant circuit, an LLC resonant circuit and an LCC resonant circuit.
The primary side of the high-frequency transformer is connected in series in the resonant circuit, and the secondary side is connected with the rectifying circuit.
Rectifying circuits come in a variety of circuit forms including half-bridge rectifying circuits formed from diodes, full-bridge rectifying circuits, full-wave rectifying circuits, and synchronous rectifying circuits formed from fully-controlled devices.
The multi-port output voltage stabilizing circuit can be composed of a filter capacitor or a filter capacitor parallel voltage stabilizing converter.
A power unit includes a self-equalizing current-equalizing switch unit, an overcurrent clamping circuit, a resonant circuit, a high-frequency transformer, a rectifying circuit and a multi-port output voltage-stabilizing circuit.
Fig. 2 (a) and fig. 2 (b) are two circuit forms of the self-equalizing current equalizing switch unit in the present embodiment, which are mainly different in the connection manner of the input capacitor and the power switch.
As shown in fig. 2 (a), the first circuit form of the self-equalizing current equalizing switch unit is: each input capacitor is connected in parallel with two power switches connected in series, namely an input capacitor C in11 And power switch M 11 And M 21 Parallel connection C in21 And M is as follows 31 And M 41 The midpoints of the two power switches connected in parallel with the same input capacitor are respectively connected with one end of the two equalizing inductors, namely the power switch M 11 And M 21 Is the midpoint b of (2) 11 And inductance L b11 Is connected with one end of the power switch M 31 And M 41 Is the midpoint b of (2) 21 And inductance L b21 Two ends of the two voltage-equalizing capacitors connected in series are respectively connected with the other ends of the two voltage-equalizing inductors, namely the inductor L b11 Is arranged at the other end c of 11 And inductance L b21 Is arranged at the other end c of 21 Voltage-equalizing capacitor C connected in series b11 And C b21 Is provided.
As shown in fig. 2 (b), the second circuit form of the self-equalizing current equalizing switch unit is: splitting each input capacitor into two input capacitors connected in series, then respectively connecting the two input capacitors with two power switches in parallel, and connecting an inductor, namely an inductor L, with the midpoint of the split two input capacitors and the midpoint of the two power switches connected in parallel 11 Is connected with the input capacitor C at both ends in11 And C in21 Is a midpoint a of (a) 11 Power switch M 11 And M 21 Is the midpoint b of (2) 11 Connection, inductance L 21 Is connected with the input capacitor C at both ends in31 And C in41 Is a midpoint a of (a) 21 Power switch M 31 And M 41 Is the midpoint b of (2) 21 The connection mode with the voltage equalizing capacitor and the impedance matching inductor is the same as that of fig. 2 (a).
In this embodiment, the resonant inductor is split into the power equalizing inductors Lb (Lb 11 to Lb22, lb numbers increase with increasing numbers of stages) and Ls (Ls 1 to Ls 3), and the equalizing capacitor Cb is split into Cb1 and Cb2 (Cb 11 to Cb23, cb numbers increase with increasing numbers of stages), so that Lb11 to Lb22 and Cb11 to Cb23 form a high-frequency ac branch, and a high-frequency ac channel is provided for a plurality of input capacitors Cin (Cin 11 to Cb22, cin numbers increase with increasing numbers of stages) with different potentials, and energy can be directly transmitted from the plurality of input capacitors to a plurality of loads. The impedance of the Lb and Cb alternating current loops is adjusted, so that the current of all switching tubes can be ensured to be completely consistent.
In the embodiment, the energy required by the soft switching of the switching tube is decoupled from the resonant cavity and the load, each group of switching tubes M11 and M21 is configured as a unit for naturally realizing the soft switching by utilizing the auxiliary inductor Lb11, and the soft switching problem of the switching tube is not needed to be considered in the design process. Because the soft switching effect of the conventional LLC converter (e.g., CN109302072 a) is closely related to the resonant cavity excitation inductance, dead time, load, etc., in order to implement soft switching, the transformer design difficulty is increased, and the converter bus voltage utilization rate is low due to the excessive dead time. Especially under the condition that a plurality of ports output, the load power of a plurality of ports is inconsistent, and the traditional converter can cause the soft switching effect of a plurality of switching tubes to have difference, even partial soft switching failure, has huge hidden danger. The embodiment realizes the decoupling of the soft switching effect and the load power, and is more suitable for occasions with multi-port output.
Fig. 3 is a schematic circuit diagram of the present embodiment when n=2.
As shown in fig. 3, when n=2, at point c 21 And c 12 2 series voltage-equalizing capacitors C are added between b12 And C b22 It can be connected with the input capacitor C in21 And C in12 High-frequency switch M 31 、M 41 、M 12 And M 22 Impedance matching inductance L b21 And L b12 A self-equalizing current-equalizing switch unit is formed, so that 3 power units can be formed at this time. Each self-equalizing current-equalizing switch unit in fig. 3 adopts the circuit form shown in fig. 2 (a), and the midpoint of the parallel branch of the input capacitor and the power switch in each power unit is connected with the input end of the subsequent resonant circuit.
As shown in fig. 3, taking a diode as an example of the overcurrent clamping circuit, one end of the diode is connected to a connection point of one voltage-sharing capacitor and one impedance matching inductor in the self-voltage-sharing current-sharing switch unit, and the other end of the diode is connected to a connection point of a resonant capacitor and a high-frequency transformer in the resonant circuit.
As shown in fig. 3, the resonant circuit is exemplified by an LLC resonant circuit, which is connected in series with the primary side of the high-frequency transformer; the rectifying circuit takes a full-bridge rectifying circuit as an example, and is connected with the secondary side of the high-frequency transformer in series; the multi-port output voltage stabilizing circuit takes a filter capacitor as an example, and is connected with the output side of the rectifying circuit.
Example 2
Fig. 4 is a graph of a resonant cavity ac voltage amplitude multiple degree of freedom control voltage regulation range of the dc converter according to embodiment 2 of the present invention.
As shown in fig. 4, the control method of the dc converter is that the ac voltage amplitude of the resonant cavity is controlled with multiple degrees of freedom. At output voltage V o2 Control as an example, control M 31 、M 41 、M 12 And M 22 The frequency multiplication low gain and high gain modes of (2) can realize the amplitude adjustment of the input alternating voltage of the resonant cavity, and the implementation details are as follows: when the input voltage is low, a high gain mode is used, i.e. M is controlled 11 、M 41 、M 12 And M 42 Synchronous on, M 21 、M 31 、M 22 And M 32 Synchronous on, at this time for V Cb11 、V Cb21 、V Cb12 、V Cb22 、V Cb13 、V Cb23 Charging, i.e. V AC1 、V AC2 And V AC3 The voltage is increased, and a high gain mode is realized; when the input voltage increases, a low gain mode is used, i.e. M is controlled 11 、M 12 Synchronous on, duty cycle less than 0.25, M 21 And M is as follows 11 Complementation, M 31 、M 32 Phase respectively leading M 11 And M 12 90 degrees, M 41 、M 42 And M is as follows 31 、M 32 Complementary at this time V Cb11 、V Cb21 、V Cb12 、V Cb22 、V Cb13 、V Cb23 The charge voltage drops to half of the high gain mode, i.e. V AC1 、V AC2 And V AC3 The voltage is reduced and a low gain mode is achieved.
For M 11 ～M 42 All switching tubes can expand the gain range by 2-3 times, 4-6 times and the like according to the step charging control in the figure 5, and the amplitude of the input alternating voltage of the resonant cavity is adjusted by 1-n times. The two control strategies can be combined to realize the voltage regulation of the input voltage which is 1-2 n times of the ultra-wide voltage of the converter, which is far higher than that of the traditional converter, and the control strategy block diagram is shown in figure 5.
Fig. 5 is a control block diagram of performing ac voltage amplitude multiple degree of freedom control on 2n-1 output ports according to this embodiment, and the specific control process includes:
degree of freedom #1: i.e. frequency doubled low gain and high gain mode switching as depicted in fig. 4，V o1 For the output voltage of port 1, V o1_ref For the desired value of the output voltage of port 1, if output V o1 Below V o1_ref Switching to high gain mode if V o1 Higher than V o1_ref Then switching to the frequency doubling low gain mode.
Degree of freedom #2: if the input voltage is too large, V is caused o1 Deviation V o1_ref If the degree of freedom #1 is excessively large and exceeds the 1-2 multiplication beneficial range, the degree of freedom #2 is started to regulate the voltage, and the implementation details are that M is added 21 And M 32 By adjusting M 21 And M 32 Duty cycle of (2) can be set to V Cin11 、V Cin21 、V Cin12 、V Cin22 Charging and discharging are carried out simultaneously on V Cb11 、V Cb21 、V Cb12 、V Cb22 、V Cb13 、V Cb23 Charging and discharging, V Cb11 、V Cb21 、V Cb12 、V Cb22 、V Cb13 、V Cb23 The voltage range of (2) Vin/4 (2n+2)) to 3.5Vin/4 (2n+2), further extending the adjustable gain range.
Degree of freedom #3: if the input voltage variation amplitude is further increased, V is caused o1 Deviation V o1_ref Excessively large, beyond the gain range of degrees of freedom #1 and #2, the regulation of degree of freedom #3 is initiated, the implementation details of which are the addition of a single regulation M in the control cycle 11 And M 42 By modulating M 11 And M 42 Duty cycle of (2) can be set to V Cb11 、V Cb21 、V Cb12 、V Cb22 、V Cb13 、V Cb23 The charging is performed, the voltage range is 3.5Vin/4 (2n (2n+2)) to 4Vin/4 (2n (2n+2)), and the adjustable gain range is further expanded.
According to the three degrees of freedom, three-level stepped ultra-wide adjustable gain is realized, the amplitude of the alternating voltage of the resonant cavity is controlled to have 2n-1 degrees of freedom in multiple degrees of freedom, and the output voltages of 2n-1 ports are independently adjustable by adjusting the proportion of the 2n-1 degrees of freedom in the adjustment of the output voltages of different ports. Compared with the prior art, the embodiment has the remarkable effects that: the voltage regulation of the ultra-wide input voltage of the converter 1-2 n times can be realized, the voltage regulation range is far larger than that of the traditional converter, and the converter is more suitable for occasions with wide input voltage ranges.
Fig. 6 is a current balance equivalent circuit of the MOSFET of embodiment 2 of the present invention. The structure of the embodiment of the invention can realize the current balance of the switching tube.
MOSFET current balancing principle:
the current of a MOSFET can be broken down into multiple voltage sources to provide current to multiple loads. At 0-T s Stage/2, to load Z ac1 The equivalent circuit of the power supply is shown in fig. 6. v 11 -v 32 For equivalent voltage source, take C in1k ＝C in2k ＝C in ，C b1k ＝C b2k ＝C b V is then 11 -v 32 Is that
Figure BDA0004111235470000081
v 11 ＝v 31 ＝v 12 ＝v 32 ＝v in /(4k) (2)
By M 11 -M 32 The current of (2) is
Figure BDA0004111235470000082
Figure BDA0004111235470000083
i M11Z1 -i M11Z3 For flowing through M 11 To the load Z ac1 、Z ac2 And Z ac3 And (5) supplying power. Z is Z M11Z1 -Z M32Z1 To the load Z ac1 Equivalent internal resistance of the power supply. If the line direct current resistance is ignored, Z M11Z1 -Z M32Z1 Is that
Figure BDA0004111235470000091
ω s Is the switching angular frequency. Equivalent internal resistance Z M11Z2 -Z M32Z2 And Z M11Z3 -Z M32Z3 The same method can be used for calculation. M is M 11 -M 32 Is the current ratio of (1)
k iM1k ＝i M1k /(i M11 +i M31 +i M12 +i M32 ) (6)
If the impedance matches the inductance omega s L b Far greater than the bus capacitance impedance, the current ratio is
Figure BDA0004111235470000092
Flow through M 11 -M 32 The current of (2) is
Figure BDA0004111235470000093
While preferred embodiments of the present application have been described, additional variations and modifications in those embodiments may occur to those skilled in the art once they learn of the basic inventive concepts. It is therefore intended that the following claims be interpreted as including the preferred embodiments and all such alterations and modifications as fall within the scope of the application.
It will be apparent to those skilled in the art that various modifications and variations can be made in the present application without departing from the spirit or scope of the application. Thus, if such modifications and variations of the present application fall within the scope of the claims and the equivalents thereof, the present application is intended to cover such modifications and variations.

Claims (10)

1. A dc converter comprising a plurality of cascaded power cells; the power unit comprises at least two input capacitors connected in series; each input capacitor is connected with at least one switching tube in parallel; all the switching tubes are connected in series; the connection midpoint of the first switching tube and the second switching tube is connected with one end of the first impedance matching inductor, the connection midpoint of the (t-1) th switching tube and the (t) th switching tube is connected with one end of the second impedance matching inductor, and t is the number of the switching tubes; the other end of the first impedance matching inductor is connected with the other end of the second impedance matching inductor through a voltage equalizing capacitor branch; the voltage-sharing capacitor branch circuit comprises at least two voltage-sharing capacitors connected in series; the connecting midpoint of two adjacent voltage-sharing capacitors is connected with one end of the resonance capacitor; the other end of the resonant capacitor is connected with one end of a primary winding of the high-frequency transformer, and the other end of the primary winding of the high-frequency transformer is connected with the connecting midpoint of the (m) th input capacitor and the (m+1) th input capacitor through a resonant inductor; wherein m=k/2, k is the number of input capacitances, and k is an even number; the secondary side of the high-frequency transformer is connected with a rectifying circuit; the rectifying circuit is connected with the multi-port output voltage stabilizing circuit.
2. The dc converter of claim 1, wherein k = 2, t = 4; each input capacitor is connected with two switching tubes in parallel.
3. The dc converter of claim 1, wherein k = 4, t = 4; each of the input capacitors is connected in parallel with one of the switching tubes.
4. A dc converter according to claim 3, wherein a connection midpoint of the first input capacitor and the second input capacitor is connected to one end of the first inductor, and the other end of the first inductor is connected to a connection midpoint of the first switching tube and the second switching tube; the connection midpoint of the third input capacitor and the fourth input capacitor is connected with one end of a second inductor, and the other end of the second inductor is connected with the connection midpoint of the third switching tube and the fourth switching tube.
5. The dc converter of claim 1, wherein each of the voltage-sharing capacitors is connected in parallel with a clamping diode; one end of the resonance capacitor is connected with the connecting midpoint of the two adjacent voltage-sharing capacitors, the other end of the resonance capacitor is connected with the connecting midpoint of the two adjacent clamping diodes, and the connecting midpoint of the two adjacent clamping diodes is connected with one end of the primary winding of the high-frequency transformer.
6. The dc converter of claim 1, wherein the rectifying circuit is one of a half-bridge rectifying circuit, a full-bridge rectifying circuit, and a synchronous rectifying circuit.
7. The dc converter of claim 1, wherein the multi-port output voltage regulator circuit comprises a filter capacitor; or the multi-port output voltage stabilizing circuit comprises a filter capacitor and a voltage stabilizing converter connected with the filter capacitor in parallel.
8. The dc converter of claim 1, wherein the input capacitors of all power cells are connected in series, the switching transistors are connected in series, and the voltage-sharing capacitors are connected in series.
9. A dc converter according to any of claims 1-8, characterized in that the inductance values of all impedance matching inductances are equal.
10. A control method of a dc converter according to any one of claims 1 to 9, characterized in that the method comprises:
when the input voltage is lower than a set threshold value, the first and t switching tubes of all the power units are synchronously turned on, and the other switching tubes of all the power units are synchronously turned on;
when the input voltage rises to exceed a set threshold value, the first switching tubes and the second switching tubes of all the power units are complementarily conducted, and the duty ratio is smaller than 0.25; the phase of the t-1 switching tube in each power unit leads the phase of the first switching tube in the power unit by 90 degrees, and the t switching tube and the t-1 switching tube of each power unit are complementarily conducted;
and/or the number of the groups of groups,
when the output voltage V of the ith power unit oi Lower than the expected value V of the output voltage of the power unit oi_ref When allThe first and the t-th switching tubes of the power units are synchronously turned on, and the rest switching tubes of all the power units are synchronously turned on; when outputting voltage V oi Higher than the expected value V of the output voltage oi_ref When the power supply is in the power supply state, the first switching tubes and the second switching tubes of all the power units are complementarily conducted, and the duty ratio is smaller than 0.25; the phase of the t-1 st switching tube in each power unit leads the phase of the first switching tube in the power unit by 90 degrees,
the t-th switching tube and the t-1-th switching tube of each power unit are complementarily conducted; i is more than or equal to 1 and less than or equal to n;
if V oi Deviation V oi_ref The switching pulse of the t-1 th switching tube in each power unit is synchronized when the first set value is exceeded, and the duty ratio of the t-1 th switching tube in each power unit is adjusted; the voltage range of the two ends of each voltage-sharing capacitor is 2Vin/4 (2n (2n+2)) -3.5 Vin/4 (2n (2n+2)), vin is the sum of the input voltages of all the power units, and n is the number of the power units;
if V oi Deviation V oi_ref The duty ratio of the first switching tube and the last switching tube in all switching tubes is adjusted to charge all voltage-sharing capacitors when the second set value is exceeded; the voltage range of the two ends of each voltage-sharing capacitor is 3.5Vin/4 (2n (2n+2)) -4 Vin/4 (2n (2n+2));
wherein the second set value is greater than the first set value.
CN202310206826.2A 2023-03-02 2023-03-02 DC converter and control method thereof Pending CN116073647A (en)

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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN117713544A (en) * 2023-12-15 2024-03-15 浙江大学 Power converter, power converter control method, device and medium

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN117713544A (en) * 2023-12-15 2024-03-15 浙江大学 Power converter, power converter control method, device and medium

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