CN115603628A - Sensorless dynamic improvement strategy for single current regulation of permanent magnet synchronous motor - Google Patents
Sensorless dynamic improvement strategy for single current regulation of permanent magnet synchronous motor Download PDFInfo
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
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- H02P21/0085—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed
- H02P21/0089—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed using field weakening
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- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
- H02P21/18—Estimation of position or speed
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- H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
- H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
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- H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
- H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation
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Abstract
The invention discloses a sensorless dynamic promotion strategy for single current regulation of a permanent magnet synchronous motor, which specifically comprises the following steps: based on the weak magnetic control of the voltage vector angle under the square wave modulation, obtaining the corresponding relation between the voltage vector angle and the electromagnetic torque; calculating the relation between the effective adjusting range of the voltage vector angle under the flux weakening control and the single current regulator, and removing an uncontrollable interval; obtaining rotor position estimated value by taking sliding mode observer under rotating coordinate system as position observation model
Thereby forming a rotating speed ring; a single-q-axis current regulator flux weakening control link is introduced into a dq-SMO-based position-sensorless closed-loop structure to improve the current dynamic response and the estimated rotating speed tracking capability. The method of the invention adopts only one current regulator no matter the traction working condition or the braking working condition, the dynamic response of the current is good, and the switching between the traction/braking control states is eliminated.
Description
Technical Field
The invention belongs to the technical field of transmission control of alternating current motors, and particularly relates to a sensorless dynamic improvement strategy for single current regulation of a permanent magnet synchronous motor.
Background
The traction transmission unit of the motor train unit has the characteristics of high voltage, large current and wide operation speed range, and simultaneously requires that the traction motor has operation capacity under the working conditions of low switching frequency and square wave. With the rapid development of high-speed rail trains, permanent magnet synchronous motors with wide speed regulation range, high power density and low energy consumption become research hotspots in the current traction field of motor train units, and the existing traction systems of the motor train units all adopt mechanical position/speed sensors to acquire position or rotating speed signals of the motors. In the actual running process of a high-speed train, the electromagnetic environment is complex, vibration is severe, a mechanical sensor is easy to lose efficacy, further a traction system fault is caused, large torque impact is caused, critical components such as a bearing, a gear and a motor are damaged in serious conditions, and the running safety of the train is damaged. The driving technology without the position sensor can fundamentally eliminate the potential safety hazard and has the advantages of strong anti-jamming capability, high integration level, long service cycle and the like.
In the field of traction of motor train units, the power level of a converter is high, a switching device is limited by heat dissipation capacity, the highest switching frequency is usually within 500Hz, the fundamental frequency of a permanent magnet synchronous traction motor can often reach more than 300Hz, and the switching frequency during synchronous 3-frequency division modulation reaches 900Hz and is far higher than the limit value of the switching frequency. In order to reduce switching loss and prolong the service life of a high-power switching tube, single-pulse square wave modulation is usually used when a motor enters a weak magnetic region, namely the rotating speed is greater than the rated rotating speed and the terminal voltage amplitude is saturated, so that the bus voltage is fully utilized while the requirement of a power device is met, and the maximum torque output range is reached. The magnitude of the voltage vector under square-wave modulation is fixed and only the voltage vector angle can be adjusted.
At present, a method for flux weakening control of a permanent magnet synchronous motor mainly comprises dual-current regulator control and voltage vector angle control. The dual-current regulator still has the characteristics of dual-regulation targets during flux weakening control, which conflicts with the characteristics of square wave working conditions, and redundant processing needs to be carried out on the regulator so as to prevent saturation. The torque output is controlled through the voltage vector angle, so that the square wave modulation characteristic is better met, and the dynamic flux weakening current track is extremely difficult to plan under the condition of no-position closed loop only through the mode that the voltage vector angle is adjustable, and meanwhile, the dynamic performance is poor. Therefore, in the permanent magnet traction system, the single current regulator is introduced by combining the modulation characteristic under the working condition of square waves, and the method has important significance for improving the dynamic response capability of the voltage vector angle flux weakening control under the condition of no position sensor operation.
Disclosure of Invention
The invention aims to provide a sensorless dynamic improvement strategy for single current regulation of a permanent magnet synchronous motor, which realizes the improvement of sensorless flux control dynamic performance under square wave modulation by introducing a single q-axis current regulator into a square wave flux weakening area.
The technical scheme adopted by the invention is that a sensorless dynamic improvement strategy for single current regulation of a permanent magnet synchronous motor is implemented according to the following steps:
and 4, introducing a single-q-axis current regulator flux weakening control link in a dq-SMO-based position-sensorless closed-loop structure to improve the dynamic response of current and the tracking capability of the estimated rotating speed.
The present invention is also characterized in that,
in the step 1, the method specifically comprises the following steps:
step 1.1, under the working condition of high-rotating-speed square waves, establishing a permanent magnet synchronous motor stable mathematical model based on a synchronous rotating coordinate system as shown in formulas (1) and (2);
T e ＝1.5P[ψ f +(L d -L q )i d ]i q (2)；
in the formulae (1) and (2), u d Is the stator voltage direct component; u. of q Is the stator voltage quadrature component; omega e Is the synchronous electrical angular velocity; t is e Is an electromagnetic torque; p is the number of pole pairs of the motor; l is d Is the stator inductance direct component; l is a radical of an alcohol q Is the quadrature component of the stator inductance; psi f Is a motor permanent magnet flux linkage; i.e. i d Is the stator current direct component; i.e. i q Is the quadrature component of the stator current;
1.2, corresponding relation between d-axis and q-axis voltages and voltage vector amplitudes is shown in a formula (3);
in the formula (3), θ VVA Is the voltage vector angle; u. of max Is the magnitude of the voltage vector, u max ＝2u dc /π，u dc Is the dc bus voltage;
step 1.3, obtaining a corresponding relation formula of the electromagnetic torque and the voltage vector angle as shown in a formula (4) by combining the vertical type (1), the formula (2) and the formula (3);
derivation of formula (4) gives formula (5):
in the step 2, the method specifically comprises the following steps:
step 2.1, in a voltage ud-uq plane, enabling a voltage vector of 0rad to be located on a ud positive half shaft, taking a counterclockwise direction as a voltage vector rotation direction, and when the angle of the voltage vector is changed from 0 to 2 pi, connecting different voltage vector end points into a circle to obtain a voltage vector angle distribution diagram on the voltage plane;
step 2.2, performing coordinate transformation on the voltage vector angle distribution on the voltage plane to obtain a voltage vector angle distribution diagram on the current plane, wherein the effective voltage vector angle ranges are both [0, pi ], and an equation of a current limiting circle and a voltage limiting ellipse is shown in a formula (6);
and 2.3, drawing a relation curve graph of the motor torque and a derivative function thereof and a voltage vector angle by the formulas (4) and (5), wherein the positive and negative signs of the torque numerical value show that the ranges of the voltage vector angle are respectively as follows: [0, π/2) and (π/2, π ]; when the output torque is 0, the voltage vector angle is pi/2;
step 2.4, combining vertical type (1) and formula (3), obtaining i d And i q The relation with the voltage vector angle is shown as formula (7);
step 2.5, based on the range [0, pi ] of the voltage vector angle, the derivative function of the formula (7) is shown as the formula (8);
in step 3, the method specifically comprises the following steps:
step 3.1, establishing a dq sliding mode observer model as shown in a formula (9);
wherein,
and
observed values of d-axis and q-axis currents, omega, of stator respectively re The angular speed of the rotor is shown, and D is a differential operator;
V d and V q Respectively controlling input of a synovial membrane observer;
the function sgn takes the value of
Gain of the sliding mode observer;
and 3.2, carrying out low-pass filtering through a PI (proportional-integral) module to obtain the estimated angular speed of the rotor
As shown in formula (10);
wherein, K po To proportional gain, K io Is the integral gain;
estimating angular velocity for a rotor
Integral operation is carried out to obtain the estimated value of the rotor position
Thereby constituting a revolution speed ring.
In step 4, the method specifically comprises the following steps:
step 4.1, calculating a preliminary d-axis current instruction through MTPA control relation
As shown in formula (11);
step 4.2, adjusting a current instruction according to the flux-weakening control target; the method specifically comprises the following steps:
step 4.2.1 d-and q-Axis feedforward voltages supplied by the feedforward element
And
calculating the amplitude of the feedback voltage
As shown in formula (12);
step 4.2.2, voltage vector amplitude u max Performing PI adjustment on the error of the feedback voltage amplitude to obtain weak magnetic compensation current delta i d,wkfd ；
Step 4.2.3, output weak magnetic compensation current delta i through voltage closed loop d,wkfd Correcting the d-axis current command to the optimal current track as shown in a formula (13);
step 4.2.4, calculating the flux weakening control constraint through a torque formulaq-axis current command
As shown in formula (14);
4.3, obtaining a final voltage output instruction based on the voltage vector angle control of the single q-axis current regulator; the method specifically comprises the following steps:
step 4.3.1, for three-phase current i A 、i B 、i C Sampling, clark transformation, and obtaining the rotor position estimated value by the step 3
Carrying out Park conversion to obtain stator current quadrature axis component i q ；
Step 4.3.2, according to the q-axis current instruction under the weak magnetic control constraint obtained in the step 4.2
And the stator current quadrature component i obtained in step 4.3.1 q The voltage vector angle θ can be obtained from equation (15) VVA ；
Wherein, K p To proportional gain, K i Is the integral gain;
and 4.3.3, calculating d-axis and q-axis command voltages output under the single q-axis current regulator voltage vector angle flux weakening control according to the formula (3), and realizing the current dynamic response and the estimated rotating speed tracking capability improvement.
The invention has the beneficial effects that:
1) The sensorless dynamic improvement of the single current regulator of the permanent magnet synchronous motor of the motor train unit under the square wave modulation under the weak magnetic control is realized;
2) The problems of regulator regulation conflict and saturation existing in a dual-current regulator flux weakening control method are essentially eliminated;
3) No matter in traction condition or brake condition, only one current regulator is adopted, the dynamic response of the current is good, and the switching between traction/brake control states is eliminated;
drawings
FIG. 1 is a schematic block diagram of a sensorless dynamic boosting strategy for single current regulation of a PMSM according to the present invention;
FIG. 2 is a block diagram of an experimental system hardware circuit architecture used in a sensorless dynamic lifting strategy for single current regulation of a permanent magnet synchronous motor according to the present invention;
FIG. 3 is a block diagram of a flux weakening control strategy under single q-axis current regulation of a permanent magnet synchronous motor according to the present invention;
FIG. 4 is a voltage vector angle definition diagram in the method of the present invention;
FIG. 5 is a voltage vector angle distribution plot on a voltage plane in the method of the present invention;
FIG. 6 is a voltage vector angle distribution plot on the current plane in the method of the present invention;
FIG. 7 is a graph of torque and its derivative function versus voltage vector for the present invention;
FIG. 8 is a PI module in a sensorless dynamic lifting strategy for single current regulation of a PMSM according to the present invention;
FIG. 9 is a waveform of the AC/DC axis current response experiment under the constant speed load torque variation without position sensor according to the present invention;
FIG. 10 is a waveform of voltage vector angle range and speed tracking performance test under the constant speed load torque variation without position sensor according to the present invention;
FIG. 11 is a waveform of the speed tracking performance test of the present invention under constant load torque without position sensor speed change.
Detailed Description
The invention relates to a sensorless dynamic lifting strategy for single current regulation of a permanent magnet synchronous motor, which is shown in a schematic block diagram of the strategy in figure 1, wherein the strategy is based on a dq-SMO sensorless closed loop structure, and a single-q-axis current regulator flux weakening control link is designed, as shown in figure 3, the dynamic response of current and the tracking capability of estimated rotating speed are improved; the method is implemented according to the following steps:
step 1.1, under the working condition of high-rotating-speed square waves, establishing a permanent magnet synchronous motor stable mathematical model based on a synchronous rotation coordinate system, wherein the stable mathematical model is shown as formulas (1) and (2);
T e ＝1.5P[ψ f +(L d -L q )i d ]i q (2)；
in the formulae (1) and (2), u d Is the stator voltage direct component; u. of q Is the stator voltage quadrature component; omega e Is the synchronous electrical angular velocity; t is e Is an electromagnetic torque; p is the number of pole pairs of the motor; l is d Is the stator inductance direct component; l is q Is the stator inductance quadrature component; psi f Is a motor permanent magnet flux linkage; i.e. i d Is the stator current direct component; i.e. i q Is the quadrature component of the stator current;
step 1.2, as shown in fig. 4, the corresponding relation between the d-axis and q-axis voltages and the voltage vector amplitude is shown in formula (3);
in the formula (3), θ VVA Is the voltage vector angle; u. of max Is the voltage vector amplitude, u under the square wave working condition max ＝2u dc /π，u dc Is the dc bus voltage;
step 1.3, obtaining a corresponding relation formula of the electromagnetic torque and a voltage vector angle by combining the vertical type (1), the formula (2) and the formula (3), and controlling the magnitude of the output torque by adjusting the voltage vector angle, wherein the formula is shown in a formula (4);
derivation of formula (4) gives formula (5):
step 2.1, in a voltage ud-uq plane, a voltage vector of 0rad is positioned on a ud positive half shaft, a counterclockwise direction is taken as a voltage vector rotating direction, when the angle of the voltage vector is changed from 0 to 2 pi, different voltage vector end points are connected into a circle, and a voltage vector angle distribution diagram on the voltage plane can be obtained, as shown in fig. 5;
step 2.2, coordinate transformation is carried out on the voltage vector angle distribution on the voltage plane, and a voltage vector angle distribution diagram on the current plane can be obtained, as shown in fig. 6, so that the effective voltage vector angle ranges are all [0, pi ], wherein an equation of a current limiting circle and a voltage limiting ellipse is shown in a formula (6);
and 2.3, drawing a relation curve graph of the motor torque and derivative function thereof and the voltage vector angle by the formulas (4) and (5), wherein as shown in fig. 7, the positive and negative signs of the torque value can indicate that the ranges of the voltage vector angle are respectively as follows under the braking and traction states: [0, π/2) and (π/2, π ]; when the output torque is 0, the voltage vector angle is pi/2, which corresponds to equation (4);
step 2.4, combining vertical type (1) and formula (3), obtaining i d And i q The relation with the voltage vector angle is shown as formula (7);
step 2.5, according to the range [0, pi ] of the voltage vector angle]The derivative function of formula (7) is represented by formula (8) and is represented by di q /dθ VVA The characteristic that the q-axis current and the voltage vector angle are in a monotone increasing relation is known, the traction/braking working condition only needs one group of adjusting parameters, and the voltage vector angle control based on the single q-axis current regulator can be used for the traction and braking working condition in flux weakening control.
step 3.1, establishing a dq sliding mode observer model as shown in a formula (9);
wherein,
and
observed values of d-axis and q-axis currents, omega, of stator respectively re The angular speed of the rotor is shown, and D is a differential operator;
V d and V q Respectively controlling input of a synovial membrane observer;
the function sgn takes the value of
Gain of the sliding mode observer;
and 3.2, obtaining the estimated angular speed of the rotor by a PI module as shown in the figure 8 and performing low-pass filtering
As shown in formula (10);
wherein, K po To proportional gain, K io Is the integral gain;
estimating angular velocity for a rotor
Integral operation is carried out to obtain the estimated value of the rotor position
Thereby forming a rotating speed ring;
and 4, introducing a single-q-axis current regulator flux weakening control link in a dq-SMO-based position-sensorless closed-loop structure to improve the dynamic response of current and the tracking capability of the estimated rotating speed. The method specifically comprises the following steps:
step 4.1, calculating a preliminary d-axis current instruction through MTPA control relation
As shown in formula (11);
step 4.2, adjusting a current instruction according to the flux-weakening control target; the method specifically comprises the following steps:
step 4.2.1 d-axis and q-axis feedforward voltages provided by a feedforward link
And
calculating the amplitude of the feedback voltage
As shown in formula (12);
step 4.2.2, voltage vector amplitude u max Performing PI adjustment on the error of the feedback voltage amplitude to obtain weak magnetic compensation current delta i d,wkfd ；
Step 4.2.3, output weak magnetic compensation current delta i through voltage closed loop d,wkfd Correcting the d-axis current command to the optimal current track as shown in a formula (13);
step 4.2.4, calculating a q-axis current instruction under the weak magnetic control constraint through a torque formula
As shown in formula (14);
and 4.3, obtaining a final voltage output instruction based on the voltage vector angle control of the single q-axis current regulator. The method specifically comprises the following steps:
step 4.3.1, for three-phase current i A 、i B 、i C Sampling, clark transformation, and the step 3Rotor position estimation
Carrying out Park conversion to obtain stator current quadrature axis component i q ；
Step 4.3.2, according to the q-axis current instruction under the weak magnetic control constraint obtained in the step 4.2
And the stator current quadrature axis component i obtained in the step 4.3.1 q The voltage vector angle θ can be obtained from equation (15) VVA ；
Wherein, K p To proportional gain, K i Is the integral gain;
and 4.3.3, calculating d-axis and q-axis instruction voltages output under the control of the voltage vector angle flux weakening of the single q-axis current regulator according to the formula (3), and realizing the dynamic response of the current and the improvement of the tracking capability of the estimated rotating speed.
When the permanent magnet synchronous motor runs at a high-speed region, the permanent magnet synchronous motor is generally positioned in a square wave working condition, the amplitude of a voltage vector under square wave modulation is fixed, and only the voltage vector angle can be adjusted. The double-current regulator in the square wave flux weakening control is very easy to saturate under the condition of direct-current bus voltage limitation, the dynamic response capability of a current loop and a torque loop is reduced, the estimation error of a rotor position is increased, and the dynamic flux weakening current track under a position-free closed loop is extremely difficult to plan only by a mode that a voltage vector angle is adjustable. The invention provides a weak magnetic control method of a single q-axis current regulator aiming at the position-sensorless weak magnetic control of a built-in permanent magnet synchronous motor under square wave modulation, so as to improve the position-sensorless weak magnetic control performance under the working condition of square waves and improve the dynamic response of current and the tracking capability of estimated rotating speed.
The system hardware structure of the present invention is shown in fig. 2, and includes: the system comprises a rectification circuit, a filter circuit, a three-phase full-bridge inverter, an IPMSM (interior permanent magnet synchronous motor), an FPGA controller, an isolation driving circuit, a rotary transformer and a current acquisition circuit; the system adopts the rotary transformer to collect the real position signal. Fig. 9 to 11 are experimental waveforms of current dynamic response and estimated speed tracking capability of a motor in a square wave flux weakening mode under the control of the hardware system shown in fig. 2 and without position sensor control after a single q-axis current regulator is used. FIG. 9 is a graph of quadrature-direct axis current response results with no position sensor single q-axis current regulation: the rotating speed is constant at 1980r/min, the actual value of the q-axis current can well follow the change of a command value when the torque step rises and falls, and the q-axis current can respectively complete command tracking in 75ms (5 fundamental wave periods) and 60ms (4 fundamental wave periods) in two dynamic processes of 5Nm → 15Nm and 15Nm → 5Nm, and the rotating speed is close to the rated rotating speed at the moment, so that better dynamic response is realized. FIG. 10 is a graph of voltage vector angle variation range and feedback estimated speed waveform for constant speed load variation: the change of the load torque can change the magnitude of a voltage vector angle during flux weakening, and the estimated rotating speed keeps good tracking performance on a rotating speed set value. Fig. 11 is a waveform diagram of the rotation speed tracking when the load torque is constant and the rotation speed is changed: under the single q-axis current regulation, the fluctuation of the estimated rotating speed reaches 25r/min, when the given value of the rotating speed suddenly increases, the fed-back estimated rotating speed can reach a new steady-state value in about 100ms, and good tracking performance is kept for the given rotating speed in the whole process.
Claims (5)
1. A sensorless dynamic lifting strategy for single current regulation of a permanent magnet synchronous motor is characterized by being implemented according to the following steps:
step 1, based on the field weakening control of a voltage vector angle under square wave modulation, obtaining the corresponding relation between the voltage vector angle and electromagnetic torque;
step 2, calculating the relation between the effective adjusting range of the voltage vector angle under the flux weakening control and the single current regulator, and removing an uncontrollable interval;
step 3, taking a sliding mode observer under a rotating coordinate system as a position observation model to obtain a rotor position estimation value
Thereby forming a rotating speed ring;
and 4, introducing a single-q-axis current regulator flux weakening control link in a dq-SMO-based position-sensorless closed-loop structure to improve the dynamic response of current and the tracking capability of the estimated rotating speed.
2. The sensorless dynamic boosting strategy for single current regulation of the permanent magnet synchronous motor according to claim 1, wherein in the step 1, specifically:
step 1.1, under the working condition of high-rotating-speed square waves, establishing a permanent magnet synchronous motor stable mathematical model based on a synchronous rotating coordinate system as shown in formulas (1) and (2);
T e ＝1.5P[ψ f +(L d -L q )i d ]i q (2)；
in the formulae (1) and (2), u d Is the stator voltage direct component; u. of q Is the stator voltage quadrature component; omega e Is the synchronous electrical angular velocity; t is e Is an electromagnetic torque; p is the number of pole pairs of the motor; l is d Is the stator inductance direct component; l is q Is the quadrature component of the stator inductance; psi f Is a motor permanent magnet flux linkage; i.e. i d Is the stator current direct component; i.e. i q Is the stator current quadrature component;
step 1.2, corresponding relation between d-axis and q-axis voltages and voltage vector amplitudes is shown as a formula (3);
in the formula (3), θ VVA Is the voltage vector angle; u. of max Is the magnitude of the voltage vector, u max ＝2u dc /π，u dc Is the dc bus voltage;
step 1.3, obtaining a corresponding relation formula of the electromagnetic torque and the voltage vector angle as shown in a formula (4) by combining the vertical type (1), the formula (2) and the formula (3);
derivation of formula (4) gives formula (5):
3. the sensorless dynamic boosting strategy for single current regulation of the permanent magnet synchronous motor according to claim 2, wherein in the step 2, specifically:
step 2.1, in a voltage ud-uq plane, enabling a voltage vector of 0rad to be located on a ud positive half shaft, taking a counterclockwise direction as a voltage vector rotation direction, and when the angle of the voltage vector is changed from 0 to 2 pi, connecting different voltage vector end points into a circle to obtain a voltage vector angle distribution diagram on the voltage plane;
step 2.2, performing coordinate transformation on the voltage vector angle distribution on the voltage plane to obtain a voltage vector angle distribution diagram on the current plane, wherein the effective voltage vector angle ranges are both [0, pi ], and an equation of a current limiting circle and a voltage limiting ellipse is shown in a formula (6);
and 2.3, drawing a relation curve graph of the motor torque and a derivative function thereof and a voltage vector angle by the formulas (4) and (5), wherein the positive and negative signs of the torque numerical value show that the ranges of the voltage vector angle are respectively as follows: [0, π/2) and (π/2, π ]; when the output torque is 0, the voltage vector angle is pi/2;
step 2.4, combining vertical type (1) and formula (3), obtaining i d And i q The relation with the voltage vector angle is shown as formula (7);
step 2.5, based on the range [0, pi ] of the voltage vector angle, the derivative function of the formula (7) is shown as the formula (8);
4. the sensorless dynamic boost strategy for single current regulation of a permanent magnet synchronous motor according to claim 3, wherein in step 3, specifically:
step 3.1, establishing a dq sliding mode observer model as shown in a formula (9);
wherein,
and
d-axis and q-axis current observed values of the stator are respectively obtained, omega re is the angular speed of the rotor, and D is a differential operator;
V d and V q Respectively controlling input of a synovial membrane observer;
the function sgn takes the value of
Gain of the sliding mode observer;
and 3.2, carrying out low-pass filtering through a PI (proportional integral) module to obtain the estimated angular speed of the rotor
As shown in formula (10);
wherein, K po To proportional gain, K io Is the integral gain;
estimating angular velocity for a rotor
Integral operation is carried out to obtain the estimated value of the rotor position
Thereby constituting a revolution speed ring.
5. The sensorless dynamic boost strategy for single current regulation of a permanent magnet synchronous motor according to claim 4, wherein in step 4, specifically:
step 4.1, calculating a preliminary d-axis current instruction through MTPA control relation
As shown in formula (11);
step 4.2, adjusting a current instruction according to the flux-weakening control target; the method specifically comprises the following steps:
step 4.2.1 d-axis and q-axis feedforward voltages provided by a feedforward link
And
calculating the amplitude of the feedback voltage
As shown in formula (12);
step 4.2.2, voltage vector amplitude u max PI (proportional integral) adjustment is carried out on the error of the feedback voltage amplitude to obtain weak magnetic compensation current delta i d,wkfd ；
Step 4.2.3, output weak magnetic compensation current delta i through voltage closed loop d,wkfd Correcting the d-axis current command to the optimal current track as shown in a formula (13);
step 4.2.4, calculating a q-axis current instruction under the weak magnetic control constraint through a torque formula
As shown in formula (14);
4.3, obtaining a final voltage output instruction based on the voltage vector angle control of the single q-axis current regulator; the method specifically comprises the following steps:
step 4.3.1, for three-phase current i A 、i B 、i C Sampling, clark transformation, and obtaining the rotor position estimated value by the step 3
Carrying out Park conversion to obtain stator current quadrature component iq;
step 4.3.2, according to the q-axis current instruction under the weak magnetic control constraint obtained in the step 4.2
And the stator current quadrature component iq obtained in the step 4.3.1, and a voltage vector angle theta can be obtained by the formula (15) VVA ；
Wherein, K p To proportional gain, K i Is the integral gain;
and 4.3.3, calculating d-axis and q-axis command voltages output under the single q-axis current regulator voltage vector angle flux weakening control according to the formula (3), and realizing the current dynamic response and the estimated rotating speed tracking capability improvement.
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| CN117175987A (en) * | 2023-08-31 | 2023-12-05 | 西安微电机研究所有限公司 | Voltage angle control weak magnetic speed expansion method of reluctance auxiliary permanent magnet synchronous motor |
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