CN114070142B - Position-sensor-free weak magnetic control strategy for permanent magnet synchronous motor of rail transit - Google Patents

Position-sensor-free weak magnetic control strategy for permanent magnet synchronous motor of rail transit Download PDF

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CN114070142B
CN114070142B CN202111219402.7A CN202111219402A CN114070142B CN 114070142 B CN114070142 B CN 114070142B CN 202111219402 A CN202111219402 A CN 202111219402A CN 114070142 B CN114070142 B CN 114070142B
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CN114070142A (en
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张航
梁文睿
高林雨
张辉
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Xian University of Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H02P21/0007—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control using sliding mode control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0085—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed
    • H02P21/0089—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed using field weakening
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/13—Observer control, e.g. using Luenberger observers or Kalman filters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H02P25/024—Synchronous motors controlled by supply frequency
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation

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  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)
  • Control Of Motors That Do Not Use Commutators (AREA)

Abstract

The invention discloses a position-sensorless flux weakening control strategy for a permanent magnet synchronous motor of rail transit, which specifically comprises the following steps: performing Fourier series expansion on the linear voltage to obtain a corresponding relation between a modulation degree and a fundamental voltage amplitude, correcting a given voltage according to the modulation degree, then sampling three-phase current, performing Clarke transformation and PI regulation, performing inverse Park transformation on the linear voltage to obtain a stator flux linkage through calculation, and obtaining an axis observation current after the Park transformation to form a double current loop; cancellation of d-axis inductance L through d-axis current response characteristics d The influence of the error on the sliding-mode observer can be eliminated, namely the d-axis inductance L d The effect of mismatch on rotor position estimation performance. The flux-weakening control performance of a position-free sensor under specific subharmonic elimination pulse width modulation is improved through a current observer and a parameter error compensation method, and the estimation precision of the rotor position is further improved.

Description

Position-sensor-free weak magnetic control strategy for permanent magnet synchronous motor of rail transit
Technical Field
The invention belongs to the technical field of transmission control of alternating current motors, and particularly relates to a position-sensorless flux-weakening control strategy of a permanent magnet synchronous motor of rail transit.
Background
With the rapid development of high-speed rail trains, permanent magnet synchronous traction motors with wide speed regulation range, high power density and low energy consumption become research hotspots in the current rail traffic field, and the existing motor train unit train traction systems all adopt mechanical position/speed sensors to acquire position or rotating speed signals of the motors. In the actual running process of a high-speed train, the electromagnetic environment is complex, vibration is severe, failure of a mechanical sensor is easily caused, a traction system is caused to break down, large torque impact is caused, critical components such as a bearing, a gear and a motor are damaged in serious conditions, and the running safety of the train is damaged. The driving technology without the position sensor can fundamentally eliminate the potential safety hazard and has the advantages of strong anti-jamming capability, high integration level, long service cycle and the like.
In order to obtain good inverter voltage output performance at low switching frequency (< =500 Hz) and make full use of the bus voltage, asynchronous modulation, piecewise synchronous modulation, and square wave modulation may be used in combination. For Interior Permanent Magnet Synchronous Machine (IPMSM) drives, flux weakening control is typically performed in synchronous modulation divide-by-3 mode.
In this case, the mismatch of the motor parameters can cause a large error in estimating the position of the rotor, and the current loop bandwidth limitation reduces the field weakening control performance of the position-sensorless. The traditional permanent magnet synchronous motor sensorless field weakening control usually adopts two current regulators. When the inverter operates in synchronous modulation 3-division or square-wave mode, the coupling of the two current regulators will be strengthened due to the modulation degree. At this point, reducing harmonic interference may increase the current loop bandwidth. Therefore, in the permanent magnet traction system, the characteristics of weak magnetic control and optimized synchronous modulation are combined, the feedback current precision is improved, the motor parameter error is compensated, and the method has important practical significance for improving the control performance of the position-sensorless traction system.
Disclosure of Invention
The invention aims to provide a position-sensorless flux weakening control strategy for a permanent magnet synchronous motor of rail transit, which improves the flux weakening control performance of a position-sensorless under specific subharmonic elimination pulse width modulation (SHEPWM) through a current observer and a parameter error compensation method, and further improves the estimation precision of the rotor position.
The technical scheme adopted by the invention is that a position-sensorless flux weakening control strategy of a permanent magnet synchronous motor of rail transit is implemented according to the following steps:
step 1, to line voltage U dc Performing Fourier expansion to obtain corresponding relation between modulation degree and fundamental voltage amplitude, and setting d-axis voltage according to the modulation degree
Figure BDA0003312003040000021
With a given voltage of q-axis
Figure BDA0003312003040000022
Corrected to obtain d-axis linear voltage
Figure BDA0003312003040000023
And q-axis linear voltage
Figure BDA0003312003040000024
Step 2, for three-phase current i A 、i B 、i C Sampling, clarke transform and PI regulation, and d-axis linear voltage regulation
Figure BDA0003312003040000025
And q-axis linear voltage
Figure BDA0003312003040000026
Carrying out inverse Park conversion to obtain alpha axis voltage
Figure BDA0003312003040000027
And beta axis voltage
Figure BDA0003312003040000028
Then, alpha-axis stator flux linkage is obtained through calculation
Figure BDA0003312003040000029
And a beta axis stator flux linkage
Figure BDA00033120030400000210
Step 3, linking the alpha-axis stator with a magnetic flux
Figure BDA00033120030400000211
And a beta axis stator flux linkage
Figure BDA00033120030400000212
After Park transformation, a d-axis stator flux linkage is obtained
Figure BDA00033120030400000213
And q-axis stator flux linkage
Figure BDA00033120030400000214
Then d-axis observation current is obtained through calculation
Figure BDA0003312003040000031
And q-axis observed current
Figure BDA0003312003040000032
Forming a double current loop;
step 4, eliminating d-axis inductance L through d-axis current response characteristics d The influence of the error on the sliding mode observer can be eliminated, namely the d-axis inductance L can be eliminated d The effect of mismatch on rotor position estimation performance.
The present invention is also characterized in that,
in the step 1, the corresponding relation between the modulation degree M and the fundamental voltage amplitude is shown as a formula (1);
Figure BDA0003312003040000033
in the step 2, the method specifically comprises the following steps: for three-phase current i A 、i B 、i C Sampling, and performing Clarke transformation to obtain d-axis current i d And q-axis current i q For d-axis current i d And q-axis current i q Performing PI regulation, and inputting a difference value between the PI regulation and the observed current into a voltage model as a compensation value so as to eliminate an integral drift effect; to d-axis linear voltage
Figure BDA0003312003040000034
And q-axis linear voltage
Figure BDA0003312003040000035
Carrying out inverse Park conversion to obtain alpha axis voltage
Figure BDA0003312003040000036
And beta axis voltage
Figure BDA0003312003040000037
Then, according to the formula (2) and the formula (3), an alpha-axis stator flux linkage is obtained
Figure BDA0003312003040000038
And a beta axis stator flux linkage
Figure BDA0003312003040000039
Figure BDA00033120030400000310
Figure BDA00033120030400000311
Wherein R is s Is a stator resistor;
Figure BDA00033120030400000312
is the current of the alpha axis and is,
Figure BDA00033120030400000313
is the beta axis current.
In step 3, d-axis observation current
Figure BDA00033120030400000314
And q-axis observed current
Figure BDA00033120030400000315
Can be obtained from formula (4) and formula (5);
Figure BDA0003312003040000041
Figure BDA0003312003040000042
wherein psi f Is stator flux linkage, L d Is d-axis inductance, L q Is the q-axis inductance.
In the step 4, the method specifically comprises the following steps:
step 4.1, current is set through d axis
Figure BDA0003312003040000043
And obtained by step 3To d-axis observation current
Figure BDA0003312003040000044
The d-axis inductance L can be obtained from the formula (6) d The deviation coefficient λ of (a);
Figure BDA0003312003040000045
wherein, K p Compensating the proportional gain, K, for the parameter error i Compensating the integral gain for the parameter error;
step 4.2, d-axis inductance actual value L d_actual Obtained by formula (7);
L d_actual ＝(1+λ)L d (7)；
step 4.3, using the actual value L of the d-axis inductor d_actual The model of the sliding-mode observer is corrected in real time, so that the d-axis inductance L can be eliminated d The effect of mismatch on rotor position estimation performance.
In step 4.3, the real-time correction specifically comprises:
step 4.3.1, obtaining the actual value L of the d-axis inductance d_actual Obtaining a sliding-mode observer model as shown in a formula (8);
Figure BDA0003312003040000046
wherein, ω is re Is the angular speed of the rotor, D is the differential operator,
Figure BDA0003312003040000051
Figure BDA0003312003040000052
the function sgn takes the value of
Figure BDA0003312003040000053
Figure BDA0003312003040000054
Observer gain for sliding mode；
And 4.3.2, performing low-pass filtering through a PI module to obtain the estimated angular speed of the rotor
Figure BDA0003312003040000055
As shown in formula (9);
Figure BDA0003312003040000056
wherein, K po To proportional gain, K io Is the integral gain;
estimating angular velocity for a rotor
Figure BDA0003312003040000057
Integral operation is carried out to obtain the estimated value of the rotor position
Figure BDA0003312003040000058
Thereby constituting a rotation speed loop.
The invention has the beneficial effects that:
1) The parameter compensation of the built-in permanent magnet synchronous motor under SHEPWM under the weak magnetic control is realized;
2) A current observer is designed, so that the bandwidth of a current loop is increased;
3) The weak magnetic control performance of a position-free sensor in SHEPWM is improved, and the position estimation precision is improved;
drawings
FIG. 1 is a functional block diagram of an improved sensorless control strategy for an interior permanent magnet synchronous machine of the present invention;
FIG. 2 is a block diagram of an experimental system hardware circuit structure used in a position sensorless improved control strategy of an interior permanent magnet synchronous motor of the present invention;
FIG. 3 is a modulation depth linearization curve of the present invention in SHE3 mode;
FIG. 4 is a current observer in an improved control strategy for an interior permanent magnet synchronous motor without a position sensor according to the present invention;
FIG. 5 is a PI module in an improved control strategy for a position sensorless PMSM according to the present invention;
FIG. 6 is a waveform of a steady state performance comparison test of the present invention without and before using the strategy under SHE3 modulation;
fig. 7 is a graph of dynamic performance comparative test waveforms of the present invention without and before using the strategy under SHE3 modulation.
Detailed Description
The invention is described in detail below with reference to the drawings and the detailed description.
The invention discloses a position sensorless flux weakening control strategy of a permanent magnet synchronous motor of rail transit, which is shown in a schematic diagram of fig. 1, is based on a double current loop, and takes the influences of parameter mismatch and current loop bandwidth limitation into consideration, a current observer is designed to increase the current loop bandwidth, and a rotor position estimation method is improved, as shown in fig. 4, and the specific steps are as follows:
step 1, d-axis set current
Figure BDA0003312003040000061
With q axis given current
Figure BDA0003312003040000062
After being adjusted by a current regulator, the d-axis given voltage is obtained
Figure BDA0003312003040000063
With a given voltage of q-axis
Figure BDA0003312003040000064
By applying a line voltage U dc Performing Fourier decomposition to obtain a corresponding relation between a modulation degree M and a fundamental voltage amplitude, as shown in formula (1), under harmonic interference, the actually calculated modulation degree is far from a target value, a corresponding linear modulation degree output value under a real voltage proportion is shown in figure 3, and after the actual modulation degree is obtained, linear correction can be performed according to data shown in the figure, that is, the real modulation degree M is subjected to linear correction 1 According to a linear modulation degree M 2 Output, i.e. for a given voltage
Figure BDA0003312003040000071
And
Figure BDA0003312003040000072
carrying out linearization to obtain d-axis linear voltage
Figure BDA0003312003040000073
And q-axis linear voltage
Figure BDA0003312003040000074
Figure BDA0003312003040000075
Step 2, for three-phase current i A 、i B 、i C Sampling, and performing Clarke transformation to obtain d-axis current i d And q-axis current i q For d-axis current i d And q-axis current i q Performing PI regulation, and inputting a difference value between the PI regulation and the observed current into a voltage model as a compensation value so as to eliminate an integral drift effect; to d-axis linear voltage
Figure BDA0003312003040000076
And q-axis linear voltage
Figure BDA0003312003040000077
Carrying out inverse Park conversion to obtain alpha axis voltage
Figure BDA0003312003040000078
And beta axis voltage
Figure BDA0003312003040000079
Then, according to the formula (2) and the formula (3), the alpha-axis stator flux linkage is obtained
Figure BDA00033120030400000710
And a beta axis stator flux linkage
Figure BDA00033120030400000711
Figure BDA00033120030400000712
Figure BDA00033120030400000713
Wherein R is s Is a stator resistor;
Figure BDA00033120030400000714
is the current of the alpha axis and is,
Figure BDA00033120030400000715
is the beta axis current.
Step 3, linking the alpha-axis stator with a magnetic flux
Figure BDA00033120030400000716
And a beta axis stator flux linkage
Figure BDA00033120030400000717
After Park transformation, a d-axis stator flux linkage is obtained
Figure BDA00033120030400000718
And q-axis stator flux linkage
Figure BDA00033120030400000719
Then d-axis observation current
Figure BDA00033120030400000720
And q-axis observed current
Figure BDA00033120030400000721
Can be obtained from formula (4) and formula (5); obtaining d-axis observed current
Figure BDA00033120030400000722
And q-axis observed current
Figure BDA00033120030400000723
Then, a double current loop can be formed to control the motor;
Figure BDA0003312003040000081
Figure BDA0003312003040000082
wherein psi f Is stator flux linkage, L d Is d-axis inductance, L q Is a q-axis inductor;
step 4, eliminating d-axis inductance L through d-axis current response characteristics d The influence of the error on the Sliding Mode Observer (SMO) is specifically as follows:
step 4.1, current is set through d-axis
Figure BDA0003312003040000083
And d-axis observation current obtained from step 3
Figure BDA0003312003040000084
The d-axis inductance L can be obtained from the formula (6) d The deviation coefficient λ of (a);
Figure BDA0003312003040000085
wherein, K p Compensating the proportional gain for parameter errors, K i Compensating the integral gain for the parameter error;
step 4.2, d-axis inductance actual value L d_actual Obtained by formula (7);
L d_actual ＝(1+λ)L d (7)；
step 4.3, using the actual value L of the d-axis inductor d_actual The model of the sliding-mode observer is corrected in real time, namely d-axis inductance L can be eliminated d The impact of mismatch on rotor position estimation performance; the real-time correction specifically comprises the following steps:
step 4.3.1, obtaining the actual value L of the d-axis inductance d_actual Obtaining a sliding-mode observer model as shown in a formula (8);
Figure BDA0003312003040000091
wherein the content of the first and second substances,
Figure BDA0003312003040000092
and
Figure BDA0003312003040000093
for d-axis and q-axis voltages, L q Q-axis inductance, ω re Is the angular speed of the rotor, D is the differential operator,
Figure BDA0003312003040000094
Figure BDA0003312003040000095
the function sgn takes the value of
Figure BDA0003312003040000096
Figure BDA0003312003040000097
For sliding-mode observer gain, R s Is the stator resistance.
Step 4.3.2, the estimated angular speed of the rotor can be obtained by the PI module shown in FIG. 5 and low-pass filtering
Figure BDA0003312003040000098
As shown in formula (9);
Figure BDA0003312003040000099
wherein, K po To proportional gain, K io Is the integral gain;
estimating angular velocity for a rotor
Figure BDA00033120030400000910
Integral operation is carried out to obtain the estimated value of the rotor position
Figure BDA00033120030400000911
Thereby forming a rotating speed ring and controlling the motor.
When the internal permanent magnet synchronous motor traction system works in a weak magnetic (FW) area with optimized synchronous modulation, the following problems occur due to motor parameter mismatch and current loop bandwidth limitation: resulting in a large rotor position estimation error and reduced field weakening control performance of the position sensorless. Aiming at the built-in permanent magnet synchronous motor position sensorless weak magnetic control under SHEPWM, the invention provides a parameter error compensation method and a current observer to improve the position sensorless weak magnetic control performance of the SHEPWM.
The hardware structure of the system of the present invention is shown in fig. 2, and includes: the system comprises a rectification circuit, a filter circuit, a three-phase full-bridge inverter, an IPMSM (interior permanent magnet synchronous motor), an FPGA controller, an isolation driving circuit, a rotary transformer and a current acquisition circuit; the system adopts a rotary transformer to collect real position signals and compares the real position signals with an estimated position. Fig. 6 to 7 are waveform diagrams comparing the steady state performance and the dynamic performance of IPMSM under the control of the hardware system shown in fig. 2 and without the strategy and after SHE3 pulse optimization synchronous modulation using the strategy. FIG. 6 is a graph of steady state performance versus position estimation performance without and with this strategy: when the strategy is not used, the time-base wave current distortion is serious, and the estimation error of the rotor position reaches 0.4rad; after the strategy is used, parameter error correction and feedback current precision are improved. FIG. 7 is a graph of current performance versus position estimation performance versus waveform without and with this strategy: when the control strategy is not used, the peak amplitude of the fundamental current is increased from 3.5A to 7A during acceleration, and the position estimation error is obviously increased. After the control strategy is used, the current waveform is smoother, the average value of the rotor position estimation error is reduced, and the current waveform can be controlled within 0.3rad in the acceleration process.

Claims (5)

1. A position-sensor-free flux weakening control strategy for a permanent magnet synchronous motor of rail transit is characterized by being implemented according to the following steps:
step 1, to line voltage U dc Performing Fourier series expansion to obtain corresponding relation between modulation degree and fundamental voltage amplitude, and setting d-axis voltage according to the modulation degree
Figure FDA0003932256890000011
With a given voltage of q-axis
Figure FDA0003932256890000012
Corrected to obtain d-axis linear voltage
Figure FDA0003932256890000013
And q-axis linear voltage
Figure FDA0003932256890000014
Step 2, for three-phase current i A 、i B 、i C Sampling, clarke transform and PI regulation, and d-axis linear voltage regulation
Figure FDA0003932256890000015
And q-axis linear voltage
Figure FDA0003932256890000016
Carrying out inverse Park conversion to obtain alpha axis voltage
Figure FDA0003932256890000017
And beta axis voltage
Figure FDA0003932256890000018
Then, alpha-axis stator flux linkage is obtained through calculation
Figure FDA0003932256890000019
And a beta axis stator flux linkage
Figure FDA00039322568900000110
Step 3, the alpha-axis stator flux linkage
Figure FDA00039322568900000111
And a beta axis stator flux linkage
Figure FDA00039322568900000112
After Park transformation, a d-axis stator flux linkage is obtained
Figure FDA00039322568900000113
And q-axis stator flux linkage psi * q_new Then d-axis observation current is obtained through calculation
Figure FDA00039322568900000114
And q-axis observed current
Figure FDA00039322568900000115
Forming a double current loop;
step 4, eliminating d-axis inductance L through d-axis current response characteristics d The influence of the error on the sliding-mode observer can be eliminated, namely the d-axis inductance L d The effect of mismatch on rotor position estimation performance; the method specifically comprises the following steps:
step 4.1, current is set through d axis
Figure FDA00039322568900000116
And d-axis observation current obtained from step 3
Figure FDA00039322568900000117
The d-axis inductance L can be obtained from the formula (6) d The deviation coefficient λ of (a);
Figure FDA00039322568900000118
wherein, K p Compensating the proportional gain for parameter errors, K i Compensating the integral gain for the parameter error;
step 4.2, d-axis inductance actual value L d_actual Obtained by formula (7);
L d-actual ＝(1+λ)L d (7)；
step 4.3, using the actual value L of the d-axis inductance d_actual The model of the sliding-mode observer is corrected in real time, namely d-axis inductance L can be eliminated d The effect of mismatch on rotor position estimation performance.
2. The position-sensorless flux-weakening control strategy for the permanent magnet synchronous motor of the rail transit according to claim 1 is characterized in that in the step 1, the corresponding relation between the modulation degree M and the amplitude of the fundamental voltage is as shown in formula (1);
Figure FDA0003932256890000021
3. the position-sensorless flux-weakening control strategy for the permanent magnet synchronous motor of the rail transit according to claim 1 is characterized in that in the step 2, the position-sensorless flux-weakening control strategy specifically comprises the following steps: for three-phase current i A 、i B 、i C Sampling, clarke transforming to obtain d-axis current i d And q-axis current i q For d-axis current i d And q-axis current i q Performing PI regulation, and inputting a difference value between the PI regulation and the observed current into a voltage model as a compensation value so as to eliminate an integral drift effect; to d-axis linear voltage
Figure FDA0003932256890000022
And q-axis linear voltage
Figure FDA0003932256890000023
Carrying out inverse Park conversion to obtain alpha axis voltage
Figure FDA0003932256890000024
And beta axis voltage
Figure FDA0003932256890000025
Then, according to the formula (2) and the formula (3), the alpha-axis stator flux linkage is obtained
Figure FDA0003932256890000026
And a beta axis stator flux linkage
Figure FDA0003932256890000027
Figure FDA0003932256890000028
Figure FDA0003932256890000029
Wherein R is s Is a stator resistor;
Figure FDA00039322568900000210
is the current of the alpha axis and is,
Figure FDA00039322568900000211
is the beta axis current.
4. The position-sensorless flux-weakening control strategy for the permanent magnet synchronous motor of the rail transit as claimed in claim 1, wherein in the step 3, d-axis observation current
Figure FDA0003932256890000031
And q-axis observed current
Figure FDA0003932256890000032
Can be obtained from formula (4) and formula (5);
Figure FDA0003932256890000033
Figure FDA0003932256890000034
wherein psi f Is stator flux, L d Is d-axis inductance, L q Is the q-axis inductance.
5. The track traffic permanent magnet synchronous motor position sensorless flux weakening control strategy according to claim 1 is characterized in that in the step 4.3, the real-time correction specifically comprises:
step 4.3.1, obtaining the actual value L of the d-axis inductance d_actual Obtaining a sliding-mode observer model as shown in a formula (8);
Figure FDA0003932256890000035
wherein, ω is re Is the angular speed of the rotor, D is the differential operator,
Figure FDA0003932256890000036
Figure FDA0003932256890000037
the function sgn takes the value of
Figure FDA0003932256890000038
Figure FDA0003932256890000039
Gain of the sliding mode observer;
and 4.3.2, carrying out low-pass filtering through a PI (proportional integral) module to obtain the estimated angular speed of the rotor
Figure FDA0003932256890000041
As shown in formula (9);
Figure FDA0003932256890000042
wherein, K po To proportional gain, K io Is the integral gain;
estimating angular velocity for a rotor
Figure FDA0003932256890000043
Integral operation is carried out to obtain the estimated value of the rotor position
Figure FDA0003932256890000044
Thereby constituting a rotation speed loop.
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