CN114600377A

CN114600377A - Radio frequency transmitter

Info

Publication number: CN114600377A
Application number: CN201980101619.2A
Authority: CN
Inventors: 钱慧珍; 罗讯
Original assignee: Huawei Technologies Co Ltd
Current assignee: Huawei Technologies Co Ltd
Priority date: 2019-11-29
Filing date: 2019-11-29
Publication date: 2022-06-07
Anticipated expiration: 2039-11-29
Also published as: US20220286333A1; WO2021103001A1; CN114600377B; EP4054082A1; EP4054082A4

Abstract

The embodiment of the application provides a radio frequency transmitter which mainly comprises a radio frequency front end and a control circuit. The radio frequency front end comprises a current source set, a compensation circuit and a matching network. The compensation circuit may compensate for load impedance differences among subsets of the N current sources in the set of current sources. Therefore, the impedance mismatch of the current source set can be improved, the power loss of the current source set is reduced, and the efficiency of the radio frequency transmitter is improved.

Description

Radio frequency transmitter

Technical Field

The application relates to the technical field of wireless communication, in particular to a radio frequency transmitter.

Background

A radio frequency transmitter is a common communication element and may be used to transmit a Radio Frequency (RF) output signal, wherein the RF output signal is generated by a radio frequency front end of the radio frequency transmitter.

Currently, more and more radio frequency front ends integrate a set of current sources. The radio frequency transmitter may control the set of current sources to generate a set output signal, on the basis of which the RF output signal is derived.

However, due to the complex circuit structure of the current source set, when the radio frequency transmitter outputs an output signal in a millimeter wave band, a parasitic effect in the current source array is significant, which further causes the efficiency of the current source array to be low, and limits the improvement of the performance of the radio frequency transmitter. Therefore, the efficiency of the rf transmitter is still to be further improved.

Disclosure of Invention

In view of the above, embodiments of the present application provide a radio frequency transmitter, in which a compensation circuit is disposed in the radio frequency transmitter to improve efficiency of the radio frequency transmitter.

In a first aspect, an embodiment of the present application provides a radio frequency transmitter, which mainly includes a radio frequency front end and a control circuit. The radio frequency front end comprises a current source set, a compensation circuit and a matching network. The current source set comprises N current source subsets, N is an integer greater than 1, the current source subsets comprise at least one current source unit, output ends of the N current source subsets are connected in parallel through an output signal line, a first end of the output signal line is connected with the matching network, and a second end of the output signal line is connected with the compensation circuit; the compensation circuit may compensate for load impedance differences between the subset of N current sources; the control circuit can output a plurality of control signals, and the control signals correspond to the current source units in the current source set one by one; the current source unit in the current source set can output unit output signals under the control of corresponding control signals, and the set output signals of the current source set comprise unit output signals of the current source unit; the matching network may impedance match the aggregate output signal of the set of current sources.

Due to the complex structure of the current source set, when the radio frequency transmitter operates in the millimeter wave band, a parasitic effect will be more apparent in the current source set. The parasitic effect can also be understood as that parasitic capacitance and parasitic inductance exist between N current source subsets of the current source set, so that load impedances of different current source subsets are different, that is, impedance mismatch occurs in the current source set. This results in increased power consumption of the current source set, which in turn reduces the efficiency of the radio frequency transmitter. In the embodiment of the application, the compensation circuit is arranged in the radio frequency front end, and the compensation circuit compensates the load impedance difference among the N current source subsets, so that the impedance mismatch in the current source sets can be improved, and the efficiency of the radio frequency transmitter is improved.

For example, in the embodiment of the present application, the compensation circuit may include a first compensation inductor and a first compensation capacitor, one end of the first compensation inductor is connected to the first end of the output signal line, the other end of the first compensation inductor is connected to one end of the first compensation capacitor, and the other end of the first compensation capacitor is grounded. The first compensation inductance and the first compensation capacitance may form an LC resonant circuit, thereby adding two transmission zeros fz1 and fz2 in the load impedance curves of the respective current source subsets. Between the two transmission zeros fz1 and fz2, the load impedance of each subset of current sources increases with increasing operating frequency, and after reaching a maximum value, the load impedance of each subset of current sources decreases with increasing operating frequency. The inductance value of the first compensation inductor and the capacitance value of the first compensation capacitor are reasonably configured, so that the maximum value of the load impedance of each current source subset can be adjusted to be close to the value of the optimal load impedance Zopt between the two transmission zeros fz1 and fz 2. Therefore, the load impedance of each current source subset can be close to the same load impedance, so that the load impedance difference among the current source subsets is reduced, and the efficiency of the radio frequency transmitter is improved. Moreover, since the load impedances of the compensated current source subsets are close to the same load impedance, which is the optimal load impedance Zopt, the embodiment of the present application is also beneficial to improving the output power of the radio frequency transmitter.

In a possible implementation manner, the first compensation inductor is an inductor with an adjustable or non-adjustable inductance value, and/or the first compensation capacitor is a capacitor with an adjustable or non-adjustable capacitance value. Under the condition that at least one of the first compensation inductor and the first compensation capacitor is an adjustable element (an adjustable inductor or an adjustable capacitor), the values of the two transmission zeros fz1 and fz2 can be flexibly adjusted by adjusting the value of the adjustable element, thereby being beneficial to flexibly adjusting the working bandwidth of the radio frequency transmitter.

In one possible implementation, the radio frequency transmitter may include two sets of current sources, one end of the compensation circuit being connected to the second end of the output signal line of one of the sets of current sources, and the other end of the compensation circuit being connected to the second end of the output signal line of the other set of current sources.

For example, the two current source sets are a positive-phase current source set and a negative-phase current source set, where the positive-phase current source set includes a plurality of positive-phase current source units and the negative-phase current source set includes a plurality of negative-phase current source units; the control circuit is specifically used for outputting a plurality of digital branch signals, a plurality of positive phase driving signals and a plurality of negative phase driving signals, and the plurality of digital branch signals are respectively in one-to-one correspondence with the plurality of positive phase current source units and the plurality of negative phase current source units; the plurality of positive phase current source units are specifically used for outputting unit output signals under the control of corresponding positive phase driving signals and corresponding digital division signals; the negative phase current source units are specifically used for outputting unit output signals under the control of corresponding negative phase driving signals and corresponding digital division signals; and the matching network is specifically used for carrying out impedance matching on the set output signal of the positive phase current source set and the set output signal of the negative phase current source set.

For example, the compensation circuit may include a second compensation inductor, a second compensation capacitor, and a third compensation inductor, and the second compensation inductor, the second compensation capacitor, and the third compensation inductor are sequentially connected in series. In case of two current source sets, the compensation circuit may be connected in series between the two current source sets. Moreover, a potential 0 point exists between two ends of the second compensation capacitor, which can be equivalent to a virtual ground. Therefore, the compensation circuit can be equivalent to two LC resonance circuits, and can compensate load impedance difference between current source subsets in two current source sets respectively.

In a possible implementation manner, the second compensation inductor is an inductor with an adjustable or non-adjustable inductance value, and/or the second compensation capacitor is a capacitor with an adjustable or non-adjustable capacitance value, and/or the third compensation inductor is an inductor with an adjustable or non-adjustable inductance value. By adopting the implementation mode, the working bandwidth of the radio frequency transmitter can be improved, and specific analysis is not repeated.

In a possible implementation manner, in the current source set, any current source unit includes a first driving transistor and a second driving transistor, and the first driving transistor and the second driving transistor form a cascade cascode circuit; the control signal comprises a digital component signal and a driving signal, wherein the grid electrode of the first driving tube is used for receiving the digital component signal corresponding to the current source unit, and the drain electrode of the first driving tube is used for outputting a unit output signal of the current source unit; the grid electrode of the second driving tube is used for receiving a driving signal corresponding to the current source unit, and the source electrode of the second driving tube is grounded.

In one possible implementation manner, the control circuit comprises an encoder, a radio frequency signal source and a driving circuit, and the control signal comprises a digital sub-signal and a driving signal; the encoder may provide a plurality of digital division signals to the drive circuit and the set of current sources, respectively; the radio frequency signal source can provide a radio frequency input signal for the driving circuit; the drive circuit may generate a plurality of drive signals based on the radio frequency input signal and the plurality of digital division signals.

It should be appreciated that there are many possible types of radio frequency transmitters provided by embodiments of the present application, which may be digital quadrature transmitters or digital polar transmitters, for example. The control circuitry varies for different types of radio frequency transmitters.

Illustratively, the radio frequency transmitter provided by the embodiments of the present application is a digital quadrature transmitter. The encoder can receive the orthogonal baseband signal and convert the orthogonal baseband signal into a plurality of digital sub-signals; the radio frequency signal source can generate an orthogonal radio frequency signal CK_IAnd quadrature radio frequency signal CK_Q(ii) a The driving circuit can divide a plurality of digital signals into signals and orthogonal radio frequency signals CK_IAnd quadrature radio frequency signal CK_QInto a plurality of drive signals.

Specifically, the plurality of current source units of the current source set comprise a plurality of first current source units and a plurality of second current source units, and the plurality of first current source units and the plurality of second current source units are in one-to-one correspondence; the unit output signal of the first current source unit and the unit output signal of the second current source unit corresponding to the first current source are orthogonal signals.

For another example, the radio frequency transmitter provided in the embodiments of the present application is a digital polarization transmitter. The encoder can receive the baseband amplitude signal and convert the baseband amplitude signal into a plurality of digital sub-signals; the radio frequency signal source can generate a phase modulation signal; the drive circuit may convert the plurality of digital division signals and the phase modulation signal into a plurality of drive signals.

In a second aspect, the present application further provides a current source array, which can be used to assemble the radio frequency transmitter provided in any one of the first aspects. Exemplarily, the current source array includes F rows of current source cells, a first output signal line, a second output signal line, E first branch signal lines, and a compensation circuit, where F and E are integers greater than 1; the E first branch signal lines are arranged between the F rows of current source units, and the E first branch signal lines are parallel to the row arrangement direction of the F rows of current source units; one or more rows of current source units are arranged between any two adjacent first branch signal lines in the E first branch signal lines; k first branch signal lines in the E first branch signal lines are connected with the first output signal line, and E-K first branch signal lines are connected with the second output signal line, wherein K is an integer greater than or equal to 1; the first end of the first output signal line can output a first output signal of the first current source array, and the first end of the second output signal line can output a second output signal of the current source array; the second end of the first output signal line and the second end of the second output signal line are both connected with the compensation circuit; the compensation circuit may compensate for a load impedance difference between the plurality of rows of current source cells adjacent to the first branch signal line.

In one possible implementation manner, the first output signal line and the second output signal line are arranged perpendicular to the row arrangement direction of the current source units in the F rows, and the first output signal line and the second output signal line are arranged adjacently.

In a possible implementation manner, the first end of the first output signal line and the first end of the second output signal line are disposed at a side close to the current source unit in the first row of the current source units in the F rows, and the compensation circuit is disposed at a side close to the current source unit in the F rows.

In a possible implementation manner, the current source array further includes H rows of current source units and G second branch signal lines, where H and G are both integers greater than 1; the G second branch signal lines are arranged between the H rows of current source units and are parallel to the row arrangement direction of the H rows of current source units; in the G second branch signal lines, one or more rows of current source units are spaced between any two adjacent second branch signal lines; among the G second branch signal lines, L second branch signal lines are connected with the first output signal line, in addition, G-L second branch signal lines are connected with the second output signal line, and L is an integer greater than or equal to 1; and the compensation circuit is also used for compensating the load impedance difference between the current source units in the rows adjacent to the second branch signal line. The H rows of current source units are additionally arranged in the current source array, so that the number of the current source units in the current source array can be increased.

In one possible implementation, a first output signal line and a second output signal line are spaced between the H-row current source unit and the F-row current source unit. That is to say, the current source units in row F are disposed on the side of the first output signal line far away from the second output signal line, and the current source units in row H are disposed on the side of the second output signal line far away from the first output signal line. By adopting the layout mode, the wiring distances between the first output signal line and the second output signal line and the plurality of rows of current source units are reduced.

These and other aspects of the present application will be more readily apparent from the following description of the embodiments.

Drawings

Fig. 1 is a schematic structural diagram of a radio frequency transmitter according to an embodiment of the present application;

fig. 2 is a schematic circuit diagram of a current source unit according to an embodiment of the present disclosure;

fig. 3 is a schematic circuit diagram of a matching network according to an embodiment of the present disclosure;

fig. 4 is a schematic structural diagram of an rf transmitter according to an embodiment of the present application;

fig. 5 is a schematic diagram illustrating a corresponding relationship between a positive-phase current source unit and a negative-phase current source unit according to an embodiment of the present disclosure;

fig. 6 is a schematic circuit diagram of a matching network according to an embodiment of the present disclosure;

fig. 7 is a schematic structural diagram of a radio frequency signal source according to an embodiment of the present application;

fig. 8 is a schematic diagram illustrating a correspondence relationship between a first current source unit and a second current source unit according to an embodiment of the present disclosure;

fig. 9 is a schematic diagram illustrating a correspondence relationship between current source units in a digital quadrature transmitter with a differential rf front end according to an embodiment of the present application;

fig. 10 is a schematic structural diagram of a radio frequency signal source according to an embodiment of the present application;

FIG. 11 is a schematic diagram illustrating parasitic effects in a current source set according to an embodiment of the present disclosure;

fig. 12 is a schematic structural diagram of a compensation circuit according to an embodiment of the present disclosure;

fig. 13 is a schematic diagram of a current source array structure according to an embodiment of the present application;

fig. 14 is a schematic diagram of a current source array structure according to an embodiment of the present application;

fig. 15 is a schematic diagram of a current source array structure according to an embodiment of the present application;

fig. 16 is a schematic diagram of a current source array structure according to an embodiment of the present application;

fig. 17 is a schematic structural diagram of a current source array according to an embodiment of the present application.

Detailed Description

In order to make the objects, technical solutions and advantages of the present application more clear, the present application will be further described in detail with reference to the accompanying drawings. The particular methods of operation in the method embodiments may also be applied to apparatus embodiments or system embodiments. It is to be noted that "at least one" in the description of the present application means one or more, where a plurality means two or more. In view of this, the "plurality" may also be understood as "at least two" in the embodiments of the present invention. "and/or" describes the association relationship of the associated objects, meaning that there may be three relationships, e.g., a and/or B, which may mean: a exists alone, A and B exist simultaneously, and B exists alone. In addition, the character "/" generally indicates that the preceding and following related objects are in an "or" relationship, unless otherwise specified. In addition, it is to be understood that the terms first, second, etc. in the description of the present application are used for distinguishing between the descriptions and not necessarily for describing a sequential or chronological order.

The technical solutions in the embodiments of the present application will be clearly and completely described below with reference to the drawings in the embodiments of the present application.

With the increase of modern wireless communication rate, a radio frequency transmitter with high output power, high efficiency and high integration level is becoming an urgent need. For example, in the fifth generation (5)^thgeneration, 5G) communication, and in the field of terahertz (THz) imaging, the performance requirements for high output power, high efficiency and high integration of a radio frequency transmitter are increasingly pressing.

At present, more and more radio frequency transmitters are integrated with a current source set to improve the integration and transmission efficiency of the radio frequency transmitters. Fig. 1 is a schematic structural diagram of an rf transmitter 100 according to an embodiment of the present disclosure, where the rf transmitter may be a digital transmitter (digital transmitter). Rf transmitter 100 includes a control circuit 101 and an rf front end 102, where rf front end 102 includes a set of current sources 1021 and a matching circuit 1022.

Current source set 1021 includes N current source subsets (10211, 10212, … …, 1021N), N being an integer greater than 1, the current source subsets including at least one current source cell. As shown in fig. 1, the outputs of the subset of N current sources are connected in parallel by output signal line 104. The first terminal p1 of the output signal line 1023 is connected to the matching network 1024, and the output signal line 1023 can output the aggregate output signal of the current source set 1021, which includes N sub-signals respectively output by N current source subsets, to the matching network 1024 via the first terminal p 1. It can also be understood that N sub-signals respectively output by the N current source subsets are superimposed to form a set output signal O of the current source set 1021.

The control circuit 101 may output a plurality of control signals corresponding to the plurality of current source units in the current source set 1021 in a one-to-one manner. For example, the current source subset includes M current source units, and the current source set 1021 includes M × N current source units. The control circuit 101 may output M × N control signals, wherein<m,n>A control signal I_<m,n>And the m current source unit A in the n current source subset of the current source set 1021_<m,n>Correspondingly, M is taken from 1 to M and N is taken from 1 to N.

In one possible implementation, the control signal may include a digital division signal and a driving signal. That is, the control signal I_<m,n>Comprising a digital partial signal D_<m,n>And a driving signal S_<m,n>. Specifically, the control circuit 101 may output M × N digital partial signals and M × N drive signals. Wherein, the first<m,n>Digital partial signal D_<m,n>And the m current source unit A in the n current source subset of the current source set 1021_<m,n>Correspondingly, M is taken from 1 to M and N is taken from 1 to N. First, the<m,n>A drive signal S_<m,n>Also connected with the current source unit A_<m,n>And correspondingly. Current source unit A_<m,n>Can be in the digital division signal D_<m,n>And a drive signal S_<m,n>Under the control of (3), the output unit outputs a signal O_<m,n>。

It is noted that the number of current source cells in different current source subsets may also be different, in which case it can still be applied to the embodiments of the present application.

In general, one or more current source sets may be included in the rf front end 102. The rf front-end 102 may include at least a single-ended type and a differential type according to the number of current source sets. Specifically, the single-ended rf front end 102 includes only one current source set, and the differential rf front end 102 may include two current source sets, and the output signals of the two current source sets are opposite signals to each other. Next, two types of rf front-ends 102 are further explained by the following examples.

Single end type

The rf front end 102 in fig. 1 is a single-ended type, which includes only one current source set 1021. In general, the current source units in the current source set 1021 have the same circuit structure. As an example, the circuit structure of the current source unit < m, n > may be as shown in fig. 2. The current source unit comprises a first driving tube M1 and a second driving tube M2, and the first driving tube M1 and the second driving tube M2 form a cascade circuit.

Specifically, the source of the first driving transistor M1 is connected to the drain of the second driving transistor M2, and the source of the second driving transistor M2 is grounded. The gate of the first driving tube M1 is used for receiving the current source unit A_<m,n>Corresponding digital partial signal D_<m,n>The drain electrode of the first driving tube M1 is used for outputting a current source unit<m,n>Cell output signal O_<m,n>. The gate of the second driving tube M2 is used for receiving the current source unit A_<m,n>Corresponding drive signal S_<m,n>。

In the current source set 1021, the unit output signals of the respective current source units are superimposed on each other in the output signal line 1023, thereby constituting a set output signal O of the current source set 1021. In one possible implementation, as shown in fig. 1, the rf front end 102 further includes a matching network 1022. The matching network 1022 may impedance match the set output signal O of the set of current sources 1021.

For example, in the single-ended rf front end 102, the circuit structure of the matching network 1022 may be as shown in fig. 3. Specifically, matching network 1022 includes a capacitor Cp, an inductor L1, an inductor L2, and a capacitor Co. One end of the capacitor Cp is connected to one end of the inductor L1, and one end of the capacitor Cp connected to the inductor L1 may receive the set output signal O of the current source set 1021. The other terminal of the inductor L1 is used for receiving the supply voltage Vi, and the other terminal of the capacitor Cp is grounded. Inductor L1 and inductor L2 are magnetically coupled. One end of the inductor L2 is connected to one end of the capacitor Co, and the other end of the inductor L2 and the other end of the capacitor Co are grounded. The end of the capacitor Co connected to the inductor L2 may output the RF output signal RFout. The end of the inductor L1 connected to the capacitor Cp and the end of the inductor L2 connected to the capacitor Co are dotted terminals.

In the matching network 1022 shown in fig. 3, the capacitance values of the capacitor Cp and the capacitor Co, and the inductance values of the inductor L1 and the inductor L2 are configured according to the load impedance RL of the rf front end 102 and the optimal load impedance Zopt of the current source set 1021. The optimal load impedance Zopt is the load impedance when the efficiency, the output power and the like of the current source set 1021 are optimal. The optimal load impedance Zopt can be referred to in the prior art, and is not described in detail herein. That is, the matching network 1022 can match the load impedance RL of the RF front end 102 to the optimal load impedance Zopt of the current source set 1021, so as to facilitate reducing the power loss generated when the RF output signal RFout passes through the load circuit.

Of the differential type

Fig. 4 illustrates an application of the radio frequency front end 102 of differential type in a radio frequency transmitter. As shown in fig. 4, the rf front end 102 includes a current source set 1021 and a current source set 1025. Wherein the current source set 1025 comprises N current source subsets (10251, 10252, … …, 1025N). Generally, in the differential rf front end 102, the specific implementation of the current source set 1021 and the current source set 1025 can refer to the implementation of the current source set 1021 in the single-ended type, which is not described herein again.

It is noted that, in general, the current source set 1021 and the current source set 1025 have the same number of current source units. The difference is that the control circuit 101 provides a plurality of control signals for the current source units in the current source set 1021 and the current source units in the current source set 1025, respectively, such that the set output signal O of the current source set 1021 is⁺And the aggregate output signal O of the current source aggregate 1025^-Are opposite signals to each other. The current source set 1021 can also be referred to as a positive phase current source set, the current source units in the current source set 1021 can also be referred to as positive phase current source units, and the output signals of the positive phase current source units can also be referred to as positive phase unit output signals. The current source set 1025 may also be referred to as a negative phase current source set, the current source units in the current source set 1025 may also be referred to as negative phase current source units, and the output signals of the negative phase current source units may also be referred to as negative phase unit output signals.

Specifically, the control circuit 101 may output a plurality of digital division signals, a plurality of positive phase drive signals, and a plurality of negative phase drive signals. The digital sub-signals output by the control circuit 101 correspond to the positive-phase current source units in the current source set 1021 respectively, and the digital sub-signals also correspond to the negative-phase current source units in the current source set 1025 respectively. It should also be understood that any digital sub-signal output by the control circuit 101 corresponds to one positive-phase current source unit and one negative-phase current source unit, and the digital sub-signal can be used to control the corresponding positive-phase current source unit and the corresponding negative-phase current source unit. The positive phase driving signals output by the control circuit 101 correspond to the positive phase current source units in the current source set 1021, and the negative phase driving signals output by the control circuit 101 correspond to the negative phase current sources in the current source set 1025.

Illustratively, as shown in FIG. 5, a positive phase current source unit A⁺ _<m,n>The corresponding digital partial signal is D_<m,n>The corresponding positive phase driving signal is S⁺ _<m,n>That is, the positive-phase current source unit a⁺ _<m,n>Corresponding control signal I⁺ _<m,n>Comprising a digital partial signal D_<m,n>And a positive phase drive signal S⁺ _<m,n>. Normal phase current source unit A⁺ _<m,n>Can be divided into D in digital signals_<m,n>And the positive phase driving signal is S⁺ _<m,n>Under the control of (3), the output unit outputs a signal O⁺ _<m,n>. The unit output signals of the individual current sources in the set 1021 constitute the set output signal O of the set 1021 of current sources⁺。

As shown in fig. 5, the negative phase current source unit a^- _<m,n>The corresponding digital sub-signal is D_<m,n>Corresponding to a negative phase drive signal of S^- _<m,n>That is, the negative phase current source unit a^- _<m,n>Corresponding control signal I^- _<m,n>Comprising a digital partial signal D_<m,n>And the negative phase drive signal is S^- _<m,n>. Negative phase current source unit A^- _<m,n>Can be divided into D in digital signals_<m,n>And negative phase driving signal is S^- _<m,n>Under the control of (2), the output unit outputs a signal O^- _<m,n>. The cell output signals of the individual current sources in the set 1025 of current sources constitute the set output signal O of the set 1025 of current sources^-。

In one possible implementation, as shown in fig. 4, the matching network 1022 is connected to the first end p1 of the output signal line 1023 in the current source set 1021 and the first end q1 of the output signal line 1026 in the current source set 1025, respectively. Matching network 1022 may output signal O to the set of current sources 1021⁺And a set output signal O of the set 1025 of current sources^-And (6) performing impedance matching.

Exemplary, differential classIn the rf front end 102, the circuit structure of the matching network 1022 can be as shown in fig. 6. Specifically, the matching network 1022 includes a capacitor Cp, an inductor L1, an inductor L2, an inductor L3, an inductor L4, and a capacitor Co. One end of the capacitor Cp is connected to one end of the inductor L1, and one end of the capacitor Cp connected to the inductor L1 may receive the set output signal O of the current source set 1021⁺. The other end of the inductor L1 is connected to one end of the inductor L3. The other end of the inductor L1 may also receive a supply voltage Vi. The other terminal of the inductor L3 is connected to the other terminal of the capacitor Cp. The other terminal of the capacitance Cp may receive a set output signal O of the set 1025 of current sources^-. Inductor L1 is magnetically coupled to inductor L2, and inductor L3 is magnetically coupled to inductor L4. One end of the inductor L2 is connected to one end of the capacitor Co, and the other end of the inductor L2 is connected to one end of the inductor L4. The other end of the inductor L4 and the other end of the capacitor Co are grounded. The end of the capacitor Co connected to the inductor L2 may output the RF output signal RFout.

In the matching network 1022 shown in fig. 6, the capacitance values of the capacitor Cp and the capacitor Co, and the inductance values of the inductor L1, the inductor L2, the inductor L3, and the inductor L4 are configured according to the load impedance RL of the rf front end 102 and the optimal load impedance Zopt of the current source set 1021.

Next, the control circuit 101 is further exemplified by taking the single-ended rf front end 102 as an example. It is noted that, without being specifically illustrated, the following implementation of the control circuit 101 is equally applicable to the differential-type rf front-end 102.

The radio frequency transmitter provided in the embodiment of the present application may be a digital transmitter, that is, the control circuit 101 may receive a digital signal and generate the plurality of control signals according to the received digital signal. Compared with the traditional analog transmitter, the digital transmitter has the characteristics of high integration level, high efficiency and high power.

As previously mentioned, the control signal includes a digital divide signal and a drive signal. Illustratively, as shown in fig. 1, the control circuit 101 in the digital transmitter mainly includes an encoder 1011, a radio frequency signal source 1012, and a driving circuit 1013. Among them, the encoder 101 may provide a plurality of digital division signals to the driving circuit 1013 and the current source set 102, respectively. Specifically, the encoder 101 may directly transmit the plurality of digital partial signals to the current source set 1021, or may forward the plurality of received digital partial signals to the current source set 1021 by the driving current 1013, which is not limited in this embodiment of the present application.

The rf signal source 1012 may provide an rf input signal to the driver circuit 1013. The driving circuit 1013 may generate a plurality of driving signals according to the radio frequency input signal and the plurality of digital division signals.

Generally, the digital polar transmitter and the digital orthogonal transmitter are two common digital transmitters, and the digital polar transmitter and the digital orthogonal transmitter are respectively taken as an example for explanation.

Digital quadrature transmitter

In a digital quadrature transmitter, encoder 1011 may receive quadrature baseband signals, illustratively including baseband signal I and baseband signal Q, which may be orthogonal to each other, where baseband signal I may be denoted as I₁,……,I _BThe baseband signal Q can be expressed as Q₁,……,Q _B. B represents the number of bits of the baseband signal I and the baseband signal Q. The encoder 1011 may encode the quadrature baseband signal, and convert the quadrature baseband signal into the plurality of digital sub-signals according to the number of current source units in the current source set 1021.

The rf signal source 1012 may generate a quadrature rf signal CK_IAnd CK_Q. Illustratively, as shown in FIG. 7, the RF signal source 1012 includes a local oscillator signal source, a quadrature generator, and a symbol mapping circuit. The local oscillation signal source may generate a local oscillation signal LO. The orthogonal generator can generate mutually orthogonal local oscillation signals LO according to the local oscillation signals LO_IAnd LO_Q. The symbol mapping circuit may be based on a symbol signal (C) of the quadrature baseband signal_IAnd C_Q) The local oscillator signal LO_IAnd LO_QConversion to quadrature radio frequency signal CK_IAnd quadrature radio frequency signal CK_Q。

The driving circuit 1013 may divide a plurality of digital signals into a plurality of quadrature radio frequency signals CK_IAnd quadrature radio frequency signal CK_QConverted into the plurality of driving signals. Wherein, the orthogonal radio frequency signal CK_IFor generating a first drive signal S_IQuadrature radio frequency signal CK_QFor generating a second drive signal S_Q. Specifically, in the digital quadrature transmitter, the plurality of driving signals generated by the driving circuit 1013 include the plurality of first driving signals S_IAnd a plurality of second drive signals S_Q。

In the current source set 1021, a part of the current source units and the plurality of first driving signals S_IIn a one-to-one correspondence, this part of the current source units may also be referred to as first current source units. In the current source set 1021, another part of the current source units and the second driving signal S_QIn one-to-one correspondence, this part of the current source units may also be referred to as second current source units. There is a one-to-one correspondence between the plurality of first current source units and the plurality of second current source units in the current source set 1021.

Illustratively, as shown in FIG. 8, a first current source unit A_IAnd a second current source unit A_QCorrespondingly, the first current source unit A is controlled_IFirst drive signal S_IAnd controlling the second current source unit A_QSecond drive signal S_QAnd correspondingly. First current source unit A_IAnd a second current source unit A_QThe cell output signals of (1) are orthogonal to each other, and the cell output signals of both can be used to generate a sub-signal O of the output signal O of the current source set 1021. It can also be understood that, in the current source set 1021, the unit output signals of the plurality of first current source units and the output signals of the plurality of second current source units are superimposed at the output signal line 1023, thereby generating the output signal O of the current source set 1021.

It should be noted that if the rf front end 102 is of a differential type, the orthogonal rf signal CK generated by the symbol mapping circuit_IComprising a quadrature-phase radio-frequency signal CK_I ⁺And negative phase quadrature radio frequency signal CK_I ^-Quadrature radio frequency signal CK_QComprising a quadrature-phase radio-frequency signal CK_Q ⁺And negative phase quadrature radio frequency signal CK_Q ^-. Wherein, the normal phase quadrature radio frequency signal CK_I ⁺For generating a first drive signal S for a first current source unit in a set 1021 of current sources_I ⁺Negative phase quadrature radio frequency signal CK_I ^-For generating a first drive signal S for a first current source cell in the set 1025 of current sources_I ^-Quadrature-phase orthogonal RF signal CK_Q ⁺For generating a second drive signal S for a second current source unit in the set 1021 of current sources_Q ⁺Negative quadrature radio frequency signal CK_Q ^-For generating a second drive signal S for a second current source cell of the set 1025 of current sources_Q ^-。

Illustratively, as shown in fig. 9, when the digital quadrature transmitter includes a differential type rf front end 102, the current source set 1021 and the current source units in the current source set 1025 correspond to the driving signals. Specifically, the current source unit a_I ⁺For any one of the first positive phase current source units in the current source set 1021, current source unit A_Q ⁺In the current source set 1021, and the current source unit A_I ⁺And a corresponding second non-inverting current source unit. Current source unit A_I ^-Is a current source unit A in the current source set 1025_I ⁺Corresponding first negative phase current source unit, current source unit A_Q ^-Is a current source unit A in the current source set 1025_I ^-And a corresponding second negative phase current source unit. Meanwhile, the current source unit A_I ^-And current source unit A_Q ⁺There is also a correspondence between them.

In FIG. 9, digital signal D_IAnd a first driving signalNumber S_I ⁺For controlling the current source unit A_I ⁺Digital signal D_QAnd a second drive signal S_I ⁺For controlling the current source unit A_Q ⁺Current source unit A_I ⁺And a current source unit A_Q ⁺The cell output signals of (1) constitute a set output signal O of a current source set 1021⁺Is divided into signals o⁺。

Digital signal D_IAnd a first drive signal S_I ^-For controlling the current source unit A_I ^-Digital signal D_QAnd a second drive signal S_I ^-For controlling the current source unit A_Q ^-Current source unit A_I ^-And a current source unit A_Q ^-Constitutes the aggregate output signal O of the current source aggregate 1025^-Is divided into signals o^-. Wherein the partial signal o⁺And the partial signal o^-Are opposite signals.

Digital polarization transmitter

With continued reference to fig. 1, in a digital polar transmitter, an encoder 1011 may receive a baseband amplitude signal a, which may be denoted as a₁,……,a _B. The encoder 1011 may encode the baseband amplitude signal a, and convert the baseband amplitude signal a into the plurality of digital sub-signals according to the number of current source units in the current source set 1021.

The radio frequency signal source 1012 may generate a phase modulated signal PM. Illustratively, as shown in FIG. 10, the RF signal source 1012 includes a local oscillator signal source, a quadrature generator, and a phase modulator. The local oscillation signal source may generate a local oscillation signal LO. The orthogonal generator can perform phase modulation on the local oscillation signal to obtain mutually orthogonal local oscillation signal LO_IAnd LO_Q. The phase modulator may convert the local oscillator signal LO according to the baseband phase signal phi corresponding to the baseband amplitude signal a_IAnd LO_QConverted into a phase modulated signal PM. The baseband amplitude signal a and the baseband phase signal phi are both obtained from the baseband signal input to the rf transmitter 100.

The driving circuit 1013 may convert the plurality of digital division signals provided by the encoder 1011 and the phase modulation signal PM provided by the rf signal source 1012 into the plurality of driving signals.

The basic architecture of rf transmitter 100 is described above. However, the digital orthogonal transmitter, the digital polarization transmitter, or other radio frequency transmitter implemented based on current source set may have a problem of low efficiency when operating in millimeter wave band.

In particular, the complex interconnections in current source set 1021 may cause significant parasitic effects when rf transmitter 100 is operating in the millimeter-wave band, thereby reducing the efficiency of rf transmitter 100. As shown in fig. 11, the subset of current sources (10211, 10212, … …, 1021N) is connected in parallel to the output signal line 1023. Due to parasitic effects, parasitic capacitances and parasitic inductances between adjacent subsets of current sources may occur. Illustratively, a parasitic inductance L is connected in series between current source subset 10211 and current source subset 10212_k1And a parasitic capacitor C connected in parallel_k1. Similarly, a parasitic inductor L is connected in series between current source subset 10212 and current source subset 10213 (not shown in the figure)_k2And a parasitic capacitor C connected in parallel_k2。

Due to the existence of parasitic inductance and parasitic capacitance, load impedance is different between different subsets of current sources in the current source set 1021, that is, impedance mismatch occurs in the current source set 1021. The impedance mismatch increases the power loss of current source set 1021, which in turn results in a decrease in the efficiency of radio frequency transmitter 100.

In view of this, as shown in fig. 1, the rf transmitter 100 according to the embodiment of the present application further includes a compensation circuit 1024, and the compensation circuit 1024 is connected to the second end p2 of the output signal line. The compensation circuit 1024 may compensate for differences in load impedance between subsets of current sources in the set of current sources 1021. Accordingly, providing compensation circuit 1024 in rf transmitter 100 facilitates reducing a difference in load impedance between subsets of current sources, thereby facilitating increasing an efficiency of rf transmitter 100.

Next, the compensation circuit 1024 will be further described by taking a single-ended rf front end and a differential rf front end as examples.

Single end type

As shown in fig. 11, in the single-ended rf front end 102, the compensation circuit 1022 includes a compensation inductor L_s1And a compensation capacitor C_s1By compensating for inductance L_s1And a compensation capacitor C_s1And (4) grounding. In particular, the compensation inductance L_s1Is connected to the second terminal p2 of the output signal line 1023 to compensate the inductance L_s1Another terminal of (1) and a compensation capacitor C_s1Is connected to a compensation capacitor C_s1And the other end of the same is grounded. Specifically, the ground in the present application refers to a ground potential of an ac signal, such as a ground potential of a dc signal or a dc power supply potential for a single-ended circuit, or an intermediate potential of a differential signal for a differential circuit.

Compensation inductance L_s1And a compensation capacitor C_s1An LC resonant circuit may be constructed so that two transmission zeroes fz1 and fz2 are added to the load impedance curve of each subset of current sources. Between the two transmission zeros fz1 and fz2, the load impedance of each subset of current sources increases with increasing operating frequency, and after reaching the maximum value, the load impedance of each subset of current sources decreases with increasing operating frequency.

Reasonably configured compensation inductor L_s1Inductance value and compensation capacitor C_s1The load impedance of each subset of current sources may be maximized between the two transmission zeros fz1 and fz2 to a value close to the ideal load impedance Zopt, which corresponds to an operating frequency within the operating frequency band of the rf transmitter 100, such that the load impedance variance of each subset of current sources is reduced.

Moreover, it is reasonableConfiguring a compensation inductance L_s1Inductance value and compensation capacitor C_s1The capacitance value of (2) can also adjust the load impedance of each current source subset to a value close to the optimal load impedance Zopt in the working frequency band. It can be seen that, with the compensation circuit 1022 provided in this embodiment of the present application, the load impedances of the respective subsets of current sources may be close to the same load impedance, which is beneficial to reducing the load impedance difference between the respective subsets of current sources and improving the efficiency of the radio frequency transmitter 100. Moreover, since the load impedances of the compensated current source subsets are close to the same load impedance, which is the optimal load impedance Zopt, the compensation circuit 1022 provided in the embodiment of the present application is also beneficial to increasing the output power of the rf transmitter 100.

In the embodiment of the present application, the compensation inductance L_s1The inductor can be an inductor with adjustable inductance value or an inductor with nonadjustable inductance value. Compensation capacitor C_s1The capacitance value of the capacitor can be adjustable, and the capacitance value of the capacitor can be nonadjustable. In the compensation inductance L_s1And a compensation capacitor C_s1In this case, when at least one of the transmission zeros is an adjustable element (an adjustable inductor or an adjustable capacitor), the values of the transmission zeros fz1 and fz2 can be flexibly adjusted by adjusting the value of the adjustable element, so that the operating bandwidth of the radio frequency transmitter 100 can be flexibly adjusted. For example, in the case where fz1 is less than fz2, increasing the value of fz1 may be beneficial to increase the operating bandwidth of rf transmitter 100, thereby improving the performance of rf transmitter 100 in a higher frequency range.

Of the differential type

The differential type rf front end 102 includes two current source sets — current source set 1021 (positive phase current source set) and current source set 1025 (negative phase current source set).

In one possible implementation, two compensation circuits may be included in the rf front end, one of which is connected to the second segment p2 of the output signal line 1023 to compensate for the difference in load impedance between the N subsets of current sources in the current source set 1021. Another compensation circuit is connected to the second terminal q2 of the output signal line 1026 for compensating for differences in load impedance between the N subsets of current sources in the set 1025 of current sources. The specific structures of the two compensation circuits can refer to the compensation circuit 1022 shown in fig. 11, which is not described in detail.

In another possible implementation, as shown in fig. 4, one end of the compensation circuit 1022 is connected to the second segment p2 of the output signal line 1023, and the other end of the compensation circuit 1022 is connected to the second end q2 of the output signal line 1026. The compensation circuit 1022 can compensate for the difference in load impedance between N subsets of current sources in the current source set 1021 and can also compensate for the difference in load impedance between N subsets of current sources in the current source set 1025.

Illustratively, as shown in FIG. 12, the compensation circuit 1022 includes a compensation inductance L_s2And a compensation capacitor C_s2And a compensation inductance L_s3And compensating for the inductance L_s2And a compensation capacitor C_s2And a compensation inductance L_s3Are connected in series in sequence. In particular, the compensation inductance L_s2Is connected to the second terminal p2 of the output signal line 1023 to compensate the inductance L_s2Another terminal of (1) and a compensation capacitor C_s2Is connected at one end. Compensation capacitor C_s2Another end of (1) and a compensation inductance L_s3Is connected to the compensation inductance L_s3And the other end thereof is connected to the second end q2 of the output signal line 1026.

It can be understood that the compensation capacitor C_s2Are opposite in polarity, thereby compensating the capacitance C_s2Has a potential 0 point between its two ends, it can also be understood that the compensation capacitor C is_s2There is a virtual ground between the two ends of the capacitor. Therefore, the compensation circuit 1022 can simultaneously perform compensation effects on the current source set 1021 and the current source set 1025, similar to the compensation circuit 1022 in fig. 11.

In the embodiment of the present application, the compensation inductance L_s2The inductance value of the inductor can be adjustable, and the inductance value of the inductor can be nonadjustable. Compensation capacitor C_s2The capacitance value of the capacitor can be adjustable, and the capacitance value of the capacitor can be nonadjustable. Compensation inductance L _s3The inductance value of the inductor can be adjustable, and the inductance value of the inductor can be nonadjustable. In the compensation inductance L_s2And a compensation capacitor C_s2And a compensation inductance L_s3In this case, when at least one of the transmission zeros is an adjustable element (an adjustable inductor or an adjustable capacitor), the values of the transmission zeros fz1 and fz2 can be flexibly adjusted by adjusting the value of the adjustable element, so that the operating bandwidth of the radio frequency transmitter 100 can be flexibly adjusted.

In summary, the embodiments of the present application provide the compensation circuit 1022 in the rf transmitter 100 to compensate for the difference of the load impedances between the N current source subsets in the current source set, so as to improve the efficiency of the rf transmitter 100.

Based on the same technical concept, the embodiments of the present application further provide a current source array, which can be used to implement any one of the radio frequency transmitters provided in the above embodiments. For example, the current source array can be applied to a digital polarization transmitter and a digital quadrature transmitter, and can be used for realizing a single-ended radio frequency front end and a differential radio frequency front end. Next, the above scenarios will be explained.

Digital polarization transmitter

Single-ended radio frequency front end

As shown in fig. 13, the current source array includes F rows of current source cells, an output signal line 1302, a compensation circuit 1301, and E branch signal lines 1305. In a possible implementation manner, a substrate 1303 may be further included, and the substrate 1303 may be configured to carry the F rows of current source units, the output signal line 1302, the compensation circuit 1301, and the E branch signal lines 1305.

A blank square in the current source array represents a current source cell. When the rf front end of the digital polar transmitter is of single-ended type, the compensation circuit 1301 includes a compensation inductor L_S1And a compensation capacitor C_S1And compensating for the inductance L_S1Is connected to the first end of the output signal line 1302, compensating the inductance L_S1Another terminal of (1) and a compensation capacitor C_S1Is connected to one end of, compensatesCapacitor C_S1And the other end of the same is grounded.

Referring to fig. 13 and fig. 1, the current source units in the row F in fig. 13 may be equivalent to the current source units in the current source set 1201 in fig. 1, and the output signal line 1302 in fig. 13 may be equivalent to the output signal line 1203 in fig. 1.

As shown in fig. 13, E branch signal lines 1305 are disposed between the current source units in the F rows, and all of the E branch signal lines are parallel to the row arrangement direction of the current source units. In addition, one or more rows of current source cells are spaced between any adjacent branch signal lines 1305. The branch signal line 1305 may transmit the cell output signal of the current source cell adjacent thereto to the output signal line 1302. The first end p1 of the output signal line 1302 can output the aggregate output signal O, the second end p2 of the output signal line 1302 and the compensation inductor L_S1And (4) connecting.

Radio frequency front end of differential type

As shown in fig. 14, the current source array further includes an output signal line 1304. In the current source unit in row F, K1305 branched signal lines are connected to the first output signal line, and E-K branched signal lines 1305 are connected to the output signal line 1304, where K is an integer greater than or equal to 1. In general, in the structure shown in fig. 14, K is F/2, and F is an even number.

Here, the current source unit connected to the output signal line 1302 through the branch signal line 1305 can be understood as a positive-phase current source unit a⁺. The current source unit connected to the output signal line 1304 through the branch signal line 1305 can be understood as a negative phase current source unit a^-。

In general, the positive phase current source unit A⁺A row of current source units formed to be connected with the negative phase current source unit A^-Are alternately arranged. When the rf front end of the digital polar transmitter is of the differential type, the compensation circuit 1301 may include a compensation inductor L_S2And a compensation capacitor C_S2And a compensation inductance L_S3And compensating for the inductance L_S2And a compensation capacitor C_S2And a compensation inductance L_S3Are connected in series in sequence.

As shown in fig. 14, in the embodiment of the present application, the inductance L is compensated_S2A compensation inductor L may be disposed adjacent to the output signal line 1302_S3A compensation capacitor C may be disposed adjacent to the output signal line 1304_S2Can be arranged on the compensation inductor L_S2And a compensation inductance L_S3In between. The layout manner shown in fig. 14 is advantageous for reducing the wiring length.

Referring to fig. 14 and fig. 4, in the current source array of fig. 14, the rows of current source units connected to the output signal line 1302 may be equivalent to the current source set 1201 in fig. 4, wherein the current source units are positive phase current source units a⁺. The rows of current source cells connected to the output signal line 1304 may correspond to the current source assembly 1205 in fig. 4, in which the current source cells are negative phase current source cells a^-。

The output signal line 1302 corresponds to the output signal line 1203 in fig. 4, and can receive the positive phase current source unit a⁺Thereby outputting a set output signal O of the set of current sources 1201⁺. The output signal line 1304 is equivalent to the output signal line 1206 in fig. 4, and can receive the negative phase current source unit a^-Thereby outputting a set output signal O of the set of current sources 1205^-。

As shown in fig. 14, the output signal line 1302 and the output signal line 1304 are arranged perpendicular to the row arrangement direction of the current source cells in row F, and the output signal line 1302 and the output signal line 1304 are arranged adjacent to each other. With this layout, it is advantageous to reduce the wiring length of the branch signal line 1305.

As shown in fig. 14, the first end p1 of the output signal line 1302 and the first end p1 of the output signal line 1304 are disposed on a side close to the current source cells in the first row of the current source cells in the F rows, and the compensation circuit 1301 is disposed on a side close to the current source cells in the F-th row of the current source cells in the F rows. That is, F rows of current source cells are spaced between the first terminal p1 and the second terminal p2 of the output signal line 1302. The output signal line 1304 works the same. With this layout, it is advantageous to reduce the wiring length of the

output signal lines

1302 and 1304.

Digital quadrature transmitter

Single-ended radio frequency front end

As shown in fig. 15, the current source array includes a plurality of first current source units a_IAnd a plurality of second current source units A_QFor any row of current source units, the first current source unit A_IAnd a second current source unit A_QAre alternately arranged in sequence. The E branch signal lines 1305 in the current source array are connected to the output signal line 1302. The current source array shown in fig. 15 is similar to that shown in fig. 13 and will not be described again.

Radio frequency front end of orthogonal type

The current source array shown in fig. 16 is similar in structure to the current source array shown in fig. 14, except that the positive phase current source unit a is provided for any one row⁺And further comprises first current source units A which are alternately arranged in sequence_I ⁺And a second current source unit A_Q ⁺. For any row of negative phase current source units A^-And comprises first current source units A alternately arranged in sequence_I ^-And a second current source unit A_Q ^-。

In a possible implementation manner, another H rows of current source units and G branch signal lines 1306 may be further included in the current source array, where H and G are integers greater than 1, so as to increase the number of current source units. Taking a digital quadrature transmitter whose rf front end is of a differential type as an example, a current source array suitable for the digital quadrature transmitter may be as shown in fig. 17.

G branch signal lines 1306 are disposed between the H rows of current source units, and the G branch signal lines 1306 are parallel to the row arrangement direction of the H rows of current source units; one or more rows of current source units are spaced between any two adjacent branched signal lines 1306 in the G branched signal lines 1306; of the G branched signal lines 1306, L branched signal lines 1306 are connected to the output signal line 1302, and G-L second branched signal lines are connected to the output signal line 1304, where L is an integer equal to or greater than 1. In this case, the compensation circuit 1301, too, can compensate for the load impedance difference between the plural rows of current source units adjacent to the branch signal line 1306.

As shown in fig. 17, the output signal line 1302 and the output signal line 1304 may be disposed between the current source cells of the F row and the current source cells of the H row to reduce the wiring length. It is also understood that the current source units in row F are disposed on the side of the output signal line 1302 away from the output signal line 1304, and the current source units in row H are disposed on the side of the output signal line 1304 away from the output signal line 1302.

While preferred embodiments of the present invention have been described, additional variations and modifications in those embodiments may occur to those skilled in the art once they learn of the basic inventive concepts. Therefore, it is intended that the appended claims be interpreted as including preferred embodiments and all such alterations and modifications as fall within the scope of the invention.

It will be apparent to those skilled in the art that various modifications and variations can be made in the embodiments of the present invention without departing from the spirit or scope of the embodiments of the invention. Thus, if such modifications and variations of the embodiments of the present invention fall within the scope of the claims of the present invention and their equivalents, the present invention is also intended to encompass such modifications and variations.

Claims

The radio frequency transmitter is characterized by comprising a radio frequency front end and a control circuit, wherein the radio frequency front end comprises a current source set, a compensation circuit and a matching network;

the current source set comprises N current source subsets, N is an integer greater than 1, each current source subset comprises at least one current source unit, output ends of the N current source subsets are connected in parallel through an output signal line, a first end of the output signal line is connected with the matching network, and a second end of the output signal line is connected with the compensation circuit;

the compensation circuit is used for compensating load impedance differences among the N current source subsets;

the control circuit is used for outputting a plurality of control signals, and the control signals correspond to a plurality of current source units in the current source set one by one;

a plurality of current source cells in the current source set for outputting cell output signals under control of corresponding control signals, the set output signals of the current source set including cell output signals of the plurality of current source cells;

the matching network is used for carrying out impedance matching on the set output signals of the current source set.
The rf transmitter of claim 1, wherein the compensation circuit includes a first compensation inductor and a first compensation capacitor, one end of the first compensation inductor is connected to the first end of the output signal line, the other end of the first compensation inductor is connected to one end of the first compensation capacitor, and the other end of the first compensation capacitor is grounded.
The rf transmitter of claim 2, wherein the first compensation inductor is an inductor with an adjustable or non-adjustable inductance value, and/or wherein the first compensation capacitor is a capacitor with an adjustable or non-adjustable capacitance value.
The rf transmitter of claim 1, comprising two of said sets of current sources, one end of said compensation circuit being connected to a second end of an output signal line of one of said sets of current sources, the other end of said compensation circuit being connected to a second end of an output signal line of the other of said sets of current sources.
The radio frequency transmitter of claim 4, wherein the compensation circuit includes a second compensation inductor, a second compensation capacitor, and a third compensation inductor, the second compensation capacitor, and the third compensation inductor being serially connected in sequence.
The radio frequency transmitter of claim 4 or 5, wherein the second compensation inductor is an inductor with an adjustable or non-adjustable inductance value, and/or the second compensation capacitor is a capacitor with an adjustable or non-adjustable capacitance value, and/or the third compensation inductor is an inductor with an adjustable or non-adjustable inductance value.
The radio frequency transmitter of any one of claims 4 to 6, wherein the two current source sets are a positive phase current source set and a negative phase current source set, wherein the positive phase current source set comprises a plurality of positive phase current source units and the negative phase current source set comprises a plurality of negative phase current source units;

the plurality of normal-phase current source units are specifically used for outputting normal-phase unit output signals under the control of corresponding control signals;

the negative phase current source units are specifically used for outputting negative phase unit output signals under the control of corresponding control signals;

the matching network is specifically configured to perform impedance matching on the aggregate output signal of the positive-phase current source aggregate and the aggregate output signal of the negative-phase current source aggregate.
The radio frequency transmitter of any one of claims 1 to 7, wherein in the set of current sources, a current source unit comprises a first driving transistor and a second driving transistor, and the first driving transistor and the second driving transistor form a cascade cascode circuit;

the control signal comprises a digital sub signal and a driving signal;

the grid electrode of the first driving tube is used for receiving the digital sub signal corresponding to the current source unit, and the drain electrode of the first driving tube is used for outputting the unit output signal of the current source unit;

the grid electrode of the second driving tube is used for receiving the driving signal corresponding to the current source unit, and the source electrode of the second driving tube is grounded.
The radio frequency transmitter of any one of claims 1 to 8, wherein the control circuit comprises an encoder, a radio frequency signal source, and a drive circuit, the control signal comprising a digital divide signal and a drive signal;

the encoder is used for respectively providing a plurality of digital division signals to the driving circuit and the current source set;

the radio frequency signal source is used for providing a radio frequency input signal to the driving circuit;

the driving circuit is used for generating a plurality of driving signals according to the radio frequency input signal and the plurality of digital division signals.
The radio frequency transmitter of claim 9, wherein the encoder is specifically configured to receive a quadrature baseband signal, convert the quadrature baseband signal to the plurality of digital component signals;

the radio frequency signal source is specifically used for generating an orthogonal radio frequency signal CK_IAnd quadrature radio frequency signal CK_Q；

The driving circuit is specifically configured to divide the plurality of digital sub-signals and the orthogonal radio frequency signal CK_IAnd CK_QInto the plurality of drive signals.
The radio frequency transmitter of any one of claims 1 to 10, wherein the radio frequency transmitter is a quadrature transmitter;

the plurality of current source units of the current source set comprise a plurality of first current source units and a plurality of second current source units, and the plurality of first current source units and the plurality of second current source units are in one-to-one correspondence;

the unit output signal of the first current source unit and the unit output signal of the second current source unit corresponding to the first current source unit are orthogonal signals.
The radio frequency transmitter of claim 9, wherein the encoder is specifically configured to receive a baseband amplitude signal, convert the baseband amplitude signal to the plurality of digital sub-signals;

the radio frequency signal source is specifically used for generating a phase modulation signal;

the driving circuit is specifically configured to convert the plurality of digital division signals and the phase modulation signal into the plurality of driving signals.
A current source array is characterized by comprising F rows of current source units, a first output signal line, a second output signal line, E first branch signal lines and a compensation circuit, wherein F and E are integers which are more than 1;

the E first branch signal lines are arranged between the current source units in the F rows and are parallel to the row arrangement direction of the current source units in the F rows,

one or more rows of current source units are arranged between any two adjacent first branch signal lines in the E first branch signal lines;

among the E first branch signal lines, K first branch signal lines are connected with the first output signal line, and in addition, E-K first branch signal lines are connected with the second output signal line, wherein K is an integer greater than or equal to 1;

the first end of the first output signal line is used for outputting a first output signal of the current source array, and the first end of the second output signal line is used for outputting a second output signal of the current source array;

the second end of the first output signal line and the second end of the second output signal line are both connected with the compensation circuit;

the compensation circuit is used for compensating load impedance difference between a plurality of rows of current source units adjacent to the first branch signal line.
The current source array according to claim 13, wherein the first output signal line and the second output signal line are arranged perpendicular to a row arrangement direction of the F rows of current source cells, and the first output signal line and the second output signal line are arranged adjacently.
The current source array of claim 14, wherein the first end of the first output signal line and the first end of the second output signal line are disposed at a side close to a first current source cell of the F current source cells, and the compensation circuit is disposed at a side close to an fth current source cell of the F current source cells.
The current source array according to claim 14 or 15, wherein the current source array further comprises H rows of current source units and G second branch signal lines, H and G each being an integer greater than 1;

the G second branch signal lines are arranged between the H-row current source units, and the G second branch signal lines are parallel to the row arrangement direction of the H-row current source units;

in the G second branch signal lines, one or more rows of current source units are spaced between any two adjacent second branch signal lines;

among the G second branch signal lines, L second branch signal lines are connected with the first output signal line, in addition, G-L second branch signal lines are connected with the second output signal line, and L is an integer greater than or equal to 1;

the compensation circuit is further used for compensating load impedance differences among the current source units in the rows adjacent to the second branch signal line.
The current source array of claim 16, wherein the first output signal line and the second output signal line are spaced between the H row of current source cells and the F row of current source cells.